JPS5949738B2 - Receiver with adaptive digital modem - Google Patents

Description

DETAILED DESCRIPTION OF THE DISCLOSURE A receiver includes means for sampling and non-interferentially demodulating a received signal to provide a digital code to each of orthogonal first and second channels. include.

The equalization circuitry equalizes the incoming signal and the phase modification circuitry modifies the phase of the equalized signal. In response to the transmission of the prologue in the first channel, a quotient is formed by dividing the signal in the first channel by the signal in a second channel that is orthogonal to the first channel. A fast learning network derives the five phase errors by determining the arctangent of . A large phase jump is made to prevent the arctangent from becoming larger than a reference number. The timing error of the sampling means is derived by determining the arctangent of the quotient formed by dividing the current digital symbol in a channel by the preceding digital symbol in that channel.

In response to an introductory impulse, the introductory samples in the impulse response are stored while the phase error is calculated from the main samples. The phase of the introduction sample is then modified. Equalization coefficients are derived from samples of the phase corrected impulse response. Using this apparatus in accordance with the preferred method of the invention, the receiver is adapted to receive normal data in less than 30 milliseconds. BACKGROUND OF THE INVENTION Field of the Invention This invention relates generally to digital data modems, and more particularly to modems that have the ability to rapidly adapt to the characteristics of a transmission channel.

Description of the Prior Art Multipoint communication networks typically include a master station that selectively communicates with one of a plurality of slave stations.

The number of slave stations may range from 5 to 12 for a particular device, for example.
Varies between stations. The transmitter and receiver of the master station are coupled to the respective transmitters and receivers of the slave stations by telephone lines. In this type of communication network, the transmitter of the master station and the receiver of the slave station are always coupled.

The master station queries each of the slave stations in turn to determine whether there is data to be transmitted. If a particular slave station has data to send to the master station, that station responds to the inquiry signal by sending an acknowledge signal followed by the data to be transmitted. Before the master station's receiver can accurately receive data from one of the slave stations, it must adapt its characteristics to the characteristics of the telephone line. The characteristics of these telephone lines are primarily due to the varying distances from the master station to each slave station; ? F so τH1゜↓
The changing characteristics of telephone lines may be sufficiently tracked using the Shizuku;ko+=;:= method.

However, when the receiver is first connected, its characteristics may differ significantly from those of the telephone line. Under these starting conditions, the relatively slow tracking methods of the prior art took as long as two seconds to adapt the receiver to the telephone line. Since this adaptation time is multiplied by the number of slaves for a single interrogation of the network, this significantly reduced the data yield of prior art systems. To facilitate quick adaptation of the receiver to the telephone line,
Various attempts have been made to provide special start-up procedures.

Typically, a particular permutation of data is transmitted by a transmitter to provide a signal with known characteristics to a receiver. Timing and phase error signals were generated in response to this particular permutation. These error signals were highly dependent on the accuracy of the automatic gain control in the receiver. As a result, there was a high probability of incorrect timing and phase learning. Although these adaptation techniques in the prior art have been effective to some extent, it has always been desirable to reduce receiver adaptation time.

Any such reduction in θ time is multiplied by the number of stations in the multipoint communication network, significantly reducing the adaptation time associated with a single interrogation of the network. It would also be desirable to provide a signal whose phase, timing, and equalization are independent of the signal magnitude and adjust to current conditions. SUMMARY OF THE INVENTION The modem of the present invention has the property of adapting not only the phase and timing characteristics but also the equalization characteristics to a particular communication channel.

This adaptation is accomplished within 30 milliseconds after sending the interrogation signal. This very short time is a great advantage during interrogation of a multipoint communication network, since as many as 20 slave stations can be interrogated in a period of less than one second.

A further advantage is the specific timing and phase error signal induced independent of the amplitude of the received signal.
This independent relationship is desirable because the adaptation is less dependent on the accuracy of the automatic gain control. As a result, the probability of incorrect tamming and learning is reduced. j
This quick learning technique is particularly suited for use with transmitters and receivers that both have first and second channels.

According to the invention, a preamble of data is sent to the first channel, while nothing is sent to the second channel.
This prologue includes a carrier wave, a modulated tone and a pair of impulses. At the receiver, the carrier waves in the preamble are used to detect transmissions and adjust automatic gain control. During the modulated tone, the phase error is calculated as a function of the arctangent of the quotient formed by dividing the signal in the second channel by the signal in the first channel. To ensure that this quotient does not approach infinity, fast learning circuitry can determine early on whether the phase error is greater than 45°. If so, a large 60° phase jump is applied so that the quotient is not greater than the reference number. Also, in response to a modulated tone, the timing is adjusted as a function of the arctangent of the quotient formed by dividing the current signal in one channel by the immediately preceding symbol in that channel. Ru. A corresponding phase jump is calculated from the timing jump. To make final adjustments to the timing, a first impulse is transmitted to provide signal components over a spectrum of significant frequencies.

Following this final timing correction, a second impulse is sent to facilitate equalization adaptation. A final phase correction is desirable before this equalization adjustment is made. This phase correction is calculated from the main samples of the impulse response, which corrects the phase of the introduction samples of the impulse response by storing the introduction samples and making subsequent phase corrections of these samples. used for Equalization adjustments are then made by calculating equalization coefficients from the impulse response samples.

With these phase, timing and equalization adjustments associated with the apparatus and method of the present invention, the receiver is ready for successful data reception. These and other features and advantages of the invention will become more apparent from the description of the preferred embodiments along with the accompanying drawings. DESCRIPTION OF THE PREFERRED EMBODIMENTS A multipoint communication network is shown in FIG.
Generally indicated by

The communication network 11 includes a main station 13 and a plurality of slave stations 15, 1
7, and 19, which transmit data via telephone lines 21 and 23. Main station 13 typically includes an office machine 25 and a modem characterized by a transmitter 27 and a receiver 29. The office machine 25 is connected to the telephone line 21 via a transmitter 27 and to the telephone line 23 via a receiver 29.
connected to. In this type of communication network 11, each slave station 15, 17 and 19 includes an office machine 31 and a modem characterized by a transmitter 33 and a receiver 35.

Office machine 3 in each of slave stations 15, 17, and 19
1 is connected to the telephone line 23 via an associated transmitter 33;
It is connected to the telephone line 21 via an associated receiver 35. Although only three slaves 15, 17 and 19 are shown in FIG. 1, it will be appreciated that communications network 11 can include any number of slaves. Telephone lines 21 and 23 may be those designated by the Bell System as unconditional telephone channels. These channels are leased telephone lines and include a pair of wires for each telephone line 21 and 23. The procedure for transmitting data in such a communication network 11 is such that the master station 13 selectively sends each of the slave stations 15, 17 and 19 to determine whether there is any data to be transmitted. It is started when a query is made to .

If there is no data to send to office machine 31 in slave stations 15, 17 and 19, it typically responds to the inquiry by sending a disapproval 7 signal. In the American standard code for information exchange, such a signal is coded as NAK. Although this communication is very short, the receiver 29 of the main station 13 will still need to connect the telephone line 29 until it can accurately receive this signal or any data.
must be adapted to the characteristics of The characteristics of these telephone channels 23 are different for each of the slave stations 15, 17 and 19. Probably. In the past, each adaptation of the receiver 29 to the characteristics of the telephone line 23 took a considerable amount of time, for example 2 seconds.

In a network containing as many as 20 slave stations, the total adaptation time was on the order of 740 seconds. No data was transmitted during this time, and as a result, the production value of prior art systems was significantly reduced. A particular advantageous feature of the modem of the present invention is its ability to adapt to the characteristics of the telephone line within a short period of 30 milliseconds.

This feature is based on the receiver 29 of the main station 13 and the slave stations 15, 17.
and 19 are of particular advantage. Since the receiver 29 must communicate with each of the transmitters 33 individually, it must be individually adapted to each telephone line associated with each of the transmitters 33. These quick learning characteristics
The transmitter 27 of the master station 13 and the receivers 35 of the slave stations 1}5, 17 and 19 are less advantageous.

Since these transmitters and receivers are always on-line, they only need to adapt to each other the first time they are switched on. The transmitter 27 is the receiver 3
Communicate with all 5 at the same time. Therefore, each receiver 35
will adapt to the associated telephone line until it is switched off.
Maintain its adaptation. The modem of the present invention has a "quick learning" capability that allows the modem to rapidly adapt to the characteristics of the telephone line 23. FIG. 2 shows the transmitter 33 in one of the slave stations 15, 17 and 19 in more detail.

The office machine 31 supplies a plurality of data words here indicated by the symbol d. Here, the subscript indicates the th data word. Following the associated office machine 31, the transmitter 33 includes an encoder 37, which has a speed of 4800 baud per second (bps).
It is adapted to receive digitized data d, from the office machine 31 at a specific rate such as . In the encoder 37, the incoming data is randomized and encoded separately to provide phase matched or DI, code and phase 90° different DQ codes in the I channel or Q channel. Data symbols DIJ and DQ represent one of a plurality of data levels, the number of which depends on the speed of operation.

For 4800 bPS operation, one of three levels (eg ±1.7
32,0), and one of five levels (eg, ±2, ±1, O) in the other I and Q channels.

Digital symbols are typically generated at a sample rate of 1600 samples per second, so that the time interval between adjacent symbols is 1/1600th of a second. This interval is usually called the baud interval, and the frequency with which the codes occur is called the baud rate. At a baud rate of 1600 codes per second, the modem output is 1600 codes per second times 3 bits per code, or 4800 bits per second. When correctly encoded, the input data symbol passes through an inhibitor 39 (described in detail below) and a pair of digital low-pass filters 41 and 4 in the I and Q channels.
3 can be conveyed to each.

These filters 41 and 43, in conjunction with similar filters in receiver 29, are tuned to give the system ideal impulse characteristics. After being properly filtered, I
The signals in the and Q channels are passed to multipliers 42 and 44, respectively, where they are multiplied by digital quantities from sine/cosine ROM 45 at a carrier frequency such as 1600 Hertz.

These modulated signals are combined in an adder 47, converted to analog form in a digital to analog converter 49, and smoothed by an analog low pass filter 51. In its analog form, the signal is conveyed to telephone line 23. Transmitter 33 also includes a starting permutation generator 55, which is controlled by a timer 57.

Timer 57 is typically clocked by the baud rate. In response to the inquiry signal from the transmitter 27,
Inhibitor 39 prevents data from encoder 37 from passing through digital low pass filters 41 and 43.

At the same time, the starting permutation generator 55 conveys data in a known order to the I channel. No data is sent during the Q channel. As explained in detail below, the generator
The channels may also be connected to carry data in a known order. In this case, no data is sent during the I channel. The known order of data in the I channel is used by receiver 29 to facilitate adapting it to telephone line 23.

Both the inhibitor 39 and the starting sequence generator 55 are enabled in response to an interrogation signal by a timer 57 operating via an ENABLE gate 59.

In the first state, the enabling gate 59 closes the circuit;
In the second state it operates as a switch providing an open circuit. When the starting permutation is complete, timer 57
changes the state of enable gate 59 to inhibit generator 55 and permit transmission of data from encoder 37 to each of the I and Q channels. The signal transmitted over telephone line 23 is distorted more or less depending on the quality of telephone line 23.

For example, telephone line 23 may move the entire data spectrum. This is typically referred to as a frequency offset. Poor telephone line 23 quality typically causes phase jitter, such that the phase of the carrier of the received signal is relative to the phase of the carrier of the transmitted signal.
It changes in a sinusoidal manner. There may also be a difference in the initial state between the phase of the carrier wave of the transmitter 33 and the phase of the carrier wave of the receiver 29. This is commonly referred to as phase offset. Telephone line 23 typically exhibits symmetric as well as asymmetric delay and attenuation distortions.

This distortion is based on the processing that telephone line 23 imparts to each particular frequency in the spectrum with respect to the frequency of the carrier wave. For example, certain frequencies in the spectrum experience greater delays than other frequencies in the spectrum. Similarly, telephone line 23 attenuates some frequencies more than others. Of course, it is desirable that receiver 29 be able to compensate for all of these undesirable characteristics of telephone line 23 in order to minimize the difference between the received and transmitted signals. Furthermore, it is desirable that receiver 29 be able to adapt to telephone line 23 in as short a period of time as possible in order to minimize the time used for operations other than data transmission. Receiver 29 is shown in more detail in FIG. It will be done.

This receiver 29 has the property of adapting to telephone lines such as lines 2 and 3 in 30 milliseconds or less to minimize delays before data is accurately received. Receiver 29 typically receives an incoming analog signal from telephone line 23 and passes this signal through analog filter 63 and automatic gain control 65. Filter 6
3 selects the desired band and automatic gain control 65 provides the desired signal level. An analog-to-digital converter 67, which may include a sampler, samples and digitizes the incoming analog signal at a rate, such as 6400 times per second, corresponding to a signed multiple or the baud rate of the transmitter 33. It is set up to do so. Converter 67 is controlled in a conventional manner by a timing signal, characterized by a stream of clock pulses.

Converter 67 responds to each pulse by taking a single sample of the analog signal and representing the amplitude of that sample as a digital number. Pulses in a timing signal are separated by a timing interval,
It can be adjusted to change the sampling rate of converter 67. It will be seen that in this embodiment, throughout the remainder of the receiver 29, the signal is in digital form.

The digital signal from converter 67 is sent to a pair of multipliers 6
9 and 7], the sine/cosine ROM7
multiplied separately by the amount from 3.

In this way, the signal is non-interferentially demodulated to the receiver 2.
It is separated into 9 I channels and 9 Q channels. To select the desired baseband from the demodulated signal, the signals are passed from multipliers 69 and 71 to a pair of digital low pass filters 75 and 77 in the 1 and Q channels, respectively.

It was previously mentioned that filters 75 and 77 in the receiver are tuned together with filters 41 and 43 in transmitter 33 to provide the ideal impulse response. In the preferred embodiment, filters 41, 43 and filters 75, 77 are arranged to provide raised cosine signaling. This type of signaling is generally preferred over partial response signaling because it provides a faster response time. Partial response signaling provides a narrow data spectrum, which may be advantageous for transmitting data over communication lines with narrow bandwidth. Raised cosine signaling, on the other hand, provides a wider data spectrum and faster response time, which is particularly desirable for providing a "quick learning" characteristic to the modem. Filters 75 and 77
The signals at the outputs of are generally designated as XI and XQ in the I and Q channels, respectively. Following filters 75 and 77, the two baseband signals in the I and Q channels are passed to equalization network 79.
This circuitry 79 corrects for symmetrical as well as asymmetrical delay and attenuation distortions produced by telephone line 23.
At the output of equalization network 79, the equalized signals in the I and Q channels are generally designated as Y and YQ''. The equalized signals Y and YQ'' are then passed through a phase correction network. 81, which compensates for frequency offset, phase offset, and phase jitter generated by telephone line 23. At the output of phase modification network 81, the equalized and phase modified signals in the I and Q channels are generally designated as YI and YQ. Following phase correction, the signals in the I and Q channels are detected in detection circuitry 83.

At the output of detection circuitry 83, the detected signals are generally designated as DI and DQ. The codes forming the DI and circle signals are expressed as follows with respect to data d. From the output of detection circuitry 83, signals DI and DQ are
It is passed along with signals YI and YQ from network 81 to error calculator 87 .

In the preferred embodiment, error calculator 87 provides an error signal to adjust the sampling rate of converter 67, equalization of circuitry 79, and phase correction of circuitry 81 to current conditions. By providing a single error calculator 87 that depends only on the incoming data, the receiver 2
It is particularly advantageous for receiver 29 that the timing, equalization, and phase of 9 are modified to compensate for imperfections in the incoming signal. Upon modification of these characteristics, the detected data from the detection circuitry 83 is passed to a decoder 89 where the signals are separately decoded, restoring the randomization and processing of the main station 13. This will be communicated to machine 25. Error calculator 87 provides error signals EI and EQ to the I and Q channels, respectively. These signals are expressed as follows. where YI and YQ are the equalized and phase corrected signals in the I and Q channels, respectively.

DI and country are detected signals and 7? o is the value 1 of the ideal signal impulse response.

is an estimate adjusted to the current state of .

Equalization control circuitry 91 receives error signals EI and EQ to provide equalization error signals that control equalization circuitry 79.
in response to the detected signal DI and the sensor.

These equalization error signals from the control circuitry 91 are in accordance with the quantity.

It is also communicated to the decision threshold control 93 which generates. This quantity. is used to set the threshold of the detection circuitry 83 and the error terms EI and E are determined in the manner described above.
It is also used to calculate Q. Timing control circuitry 94 is made responsive to the EI and EQ signals from error calculator 87 to control the timing of sampler 67.

Timing control circuitry 94 may be of a type responsive to specific signals in equalization circuitry 79. This timing control circuitry is adaptable for use with the raised cosine signaling method preferred in the modem of the present invention. Error calculator 87 also provides a signal EPLL derived from error terms EI and EQ.

This signal EPLL is conveyed to phase control circuitry 95, which provides a signal to phase modification circuitry 81. receiver 29
The device for rapidly adapting the telephone line 23 to the telephone line 23 is primarily controlled by the rapid learning circuitry 97. This circuit network 97 is
It is shown in block form in FIG. 3 and in more detail in FIG. Quick learning circuitry 97 is adapted to receive signals YI and YQ from phase modification circuitry 81 .
These signals YI and YQ each include a plurality of digital symbols generated as a result of sampling by converter 67. For example, the YI signal includes the current symbol YIo occurring in the current baud interval and the symbol YI-1 occurring in the previous baud interval. Similarly, the YQ signal is the current symbol YQo occurring in the current baud interval.
and the code YQ. generated in the previous baud interval. ．． Contains 1. Prior to transmitting data onto telephone line 23, a transmitter, such as transmitter 33, sends a special quick learning prologue to a receiver, such as receiver 29.

This introductory section may be similar to that shown in FIG. 5 and designated by reference numeral 99. The prologue 99 for the preferred embodiment of the invention has five periods each equal to ten baud intervals. These periods are times T shown in FIG. This is clarified by T1. First period T.

- During T1, an unmodulated carrier wave 101 is transmitted. This carrier wave may be the same as the carrier wave for data transmission,
In the preferred embodiment it has a frequency of 1600 Hertz. During the second period T1-T4, the carrier wave is modulated with a particular frequency to generate a modulated tone 103. The particular frequency may be one quarter of the carrier frequency, which in the preferred embodiment is 400.
It's Hertz. At the end of the third period T4-T5, a first impulse 105 is transmitted.

Similarly, at the end of the fourth period T5T6, a second impulse 107 is transmitted. After a delay associated with the fifth period T6-T7, normal data transmission can begin. To
The time from T7 corresponds to the operations performed in receiver 29 according to the flow chart shown in FIG.

First period T. - During T1, the receiver 29 detects the carrier wave and causes the automatic gain control to adjust the received signal to a predetermined value. This operation is illustrated by block 109 in the flow chart of FIG. After a period, such as 10 baud, the signal should be sufficiently adjusted so that the receiver 29 is better adapted to the timing, phase and equalization characteristics of the telephone line 23. During the second period T1-T4, the modulated tone 103 (
5) is transmitted to the receiver 29.

In response to this modulated tone 103, in FIG. As shown by block 110, a significant phase jump occurs between times T1 and T2. As shown by block 112, a second phase jump occurs between times T2 and T3. From time T3 to T4, both timing and phase jumps occur, as indicated by block 114. Third period T4-T
At the end of 5, an impulse 105 is transmitted to the receiver 29
The timing and phase of block 1 shown in FIG.
16, jump together.

A phase modification is also performed at the end of the fourth period T5-T6 in response to the second impulse 107. This phase modification is illustrated by block 118 in FIG. The equalization adjustment represented by block 120 is also made in the fourth period after the phase correction associated with block 118. After a suitable delay associated with a fifth period T6-T7, normal operation of receiver 29 begins, as indicated by block 122 in FIG. To explain the second period T1-T4 in more detail, the phase adjustment in the quadrature modulation method is
and one of the Q channels. It was found that this could be accomplished most quickly by transmitting a tone and then measuring the phase error.

A particularly accurate and easy-to-perform phase error measurement is
It has been discovered that this can be done by taking the arc tangent of the quantity formed by dividing the signal in one channel by the signal in the other channel.
For example, if the quick learning prologue 99 is sent to the Q channel, the phase error is measured by the following equation. Here YIo is the code occurring in the I channel at the output of phase modification network 81 during the current baud interval.

YQo is the code generated in the Q channel at the output of phase modification network 81 during the current baud interval.

Assuming that the prologue 99 is transmitted in the Q channel and there is no phase error, the sign YIo will be equal to O.

Thus the quotient YIo/YQo will be equal to O, and the arctangent of that quotient will also be O. Under this condition, the quantity Δφ is equal to O, indicating no phase error. It has been found that the arcsine of the quotient YIo/YQo gives similar results.

Thus, if there is no phase error and signal YIo is equal to O, then the arcsine of its quotient will also be O. However, in certain embodiments it may be more desirable to use an arctangent function. This is because this function goes from 0 to 1 in the range of 0° and 45°. On the other hand, the sine function ranges from 0° to 45°, with a magnitude of 0 to slightly 0. It only becomes 707. Thus, the arctangent function provides greater accuracy for more typical low value phase errors. On the other hand, there may be difficulties in obtaining arctangent functions for angles between 45° and 90°.

In this range, the arctangent function goes from a value of 1 to infinity. Since it is difficult to maintain accuracy over such a wide range of values, it is desirable in the first place to ensure that the phase error is less than 45°.
This means that the magnitude of YIo is the reference level l of the receiver. This is achieved in the preferred embodiment, giving a constant phase jump of about 60 degrees when greater than or equal to. This is more easily understood with reference to the 400 hertz demodulated tone shown in FIG. If no timing errors existed, this 400 hertz tone would be on line 11.
It will be sampled at a time corresponding to 1. If the timing error is a maximum of 1/2 baud, then a sample is taken at a time corresponding to line 113. This maximum timing error, which corresponds to 1/2 baud in the preferred embodiment, produces a sign, one of the alternating signals having a magnitude of 2.451o. Without timing errors, the sample would have a value of 1.7321o. These dimensions are indicated in FIG. 8 by a pair of circles 115 and 117, respectively. 1 yen
A pair of vectors 119 and 1 related to 15 and 117
21 are each drawn for YIo components equal to 1o.

The angular difference between the Q-axis and these vectors 119 and 121 is the phase error Δφ to be corrected. If, corresponding to vector 119, there is no timing error and the YIo component is 1. If it is smaller, the maximum phase error is found to be 35.3°. In comparison, if, corresponding to vector 121, the timing error is maximum and the YIo component is l. If smaller, the maximum phase error is 24. It is Ding. In either case, if the YIo component is l. If smaller, the phase error (regardless of timing error) will be less than 45°. Thus, the arc-tangent function is, if YIo is 1. It is particularly desirable to determine the magnitude of the phase error if the magnitude of the YIo component is less than 1.

If larger, the reference phase of the receiver 29 will have a phase error of 4
It is jumped by 60° to make it smaller than 5°. For example, if the YQo component is equal to O, which indicates a 90° phase error, a 60° jump will reduce the phase error to 30° regardless of the evening timing error. YIo is l. In the other extreme case, a 60 degree jump of zero timing error corresponding to vector 119 would result in a phase error of 24.7 degrees. Vector 12
A 60 degree jump with a maximum timing error corresponding to 1 would result in a phase error of 35.9 degrees. These values are tabulated in FIG. YIo is l. It can be seen that the phase error is smaller than 45°. If YIo is l. In this case, by jumping the reference phase by 60°, the phase error is made smaller than 45°. Under these circumstances, an arctangent function for determining the phase error according to Equation 1 is particularly desirable. The 60° phase jump associated with block 110 of FIG. 6 is determined according to the following equation:

△φ=-60°SGN (YIoYQo) (Formula 2) 60
゜ The direction of the jump is determined by the amount △φ determined according to Equation 2.
According to the sign.

When the sign of Δφ is negative, the referenced phase is adjusted clockwise in FIG. When the sign is positive, the referenced phase is adjusted counterclockwise in FIG. The signal Δφ calculated according to Equation 2 is guided by a fast learning network shown in detail in FIG.

This quick learning network 97 each has a pair of conductors 12
3 and 125, receive signals YIo and YQo.
These signals are both multiplied in a multiplier 127,
A comparator 129 is provided with the resulting product. Comparator 12
9 conveys the sign of this product to the multiplier 131, and this multiplier 1
31 also receives the element -60°. Multiplier 131
At the output of the signal △φ=-60°SGN(YIo
YQo) is supplied to conductor 133. Vector component Y
If Io is greater than or equal to lo, this Δφ signal on conductor 133 is used to jump the reference phase. Y
The magnitude of Io is determined by transmitting the YIo signal on the conductor 123 to the magnitude extractor 135 and passing the resulting signal to the comparator 137.
Amount 1 in.

Determined by comparing with The output of the comparator 137 has a YIo component having a magnitude of at least l. , it provides a signal on conductor 139. Conductor 1
This signal on 39 is used to transfer the Δφ signal on conductor 133 to phase control network 95. Since the phase error is guaranteed to be less than 45°, the phase correction described in Equation 1 is more easily induced and determines the accurate phase error.

This calculation is done according to the structure shown in FIG. In this structure, network 97 includes a switch 141 having a first position, a second position and a third position corresponding to terminals 143, 144 and 145. The second switch 147 corresponds to a pair of terminals 149 and 151.
It has a first and second position. These switches 141
and 147 in their first position, conductor 12
3 passes the signal YIo through the switch 141 to the multiplier 153.
It is connected to terminal 143 so as to transmit the signal. conductor 125
passes the signal YQo through the switch 147 to the inverse R
Connected to terminal 149 to communicate to OM 155.
The signal 1/YQo at the output of ROM 155 also supplies certain elements to multiplier 153. The resulting product YIo/Y
Qo is communicated to ARCTANROM 157 to provide the quantity ARCTANYIo/YQo. By transmitting this signal through the inverter 159, the signal Δφ=-ARCTAN(YIo/YQo
). This signal on conductor 161 is communicated to phase control network 95, as shown in FIG. In phase control network 95, the reference phase angle is adjusted by Δφ degrees, which corresponds to the phase angle of the received signal. In the method described above, both large and fine phase jumps of the magnitude shown in blocks 110 and 112 of FIG. 6 are performed using the modulated tone 103 shown in FIG. Once the phase of receiver 29 is accurately adjusted, the timing is also adjusted using modulated tone 103 according to the following equation:

Here, YQo is the signal YQ-1 in the Q channel in the current baud interval, and the magnitude of the signal YQo in the Q channel in the previous baud interval is greater than or equal to the magnitude of YQ-1.

Calculating the timing error according to Equation 3 is more easily understood with reference to the timing diagram shown in FIG.

In this timing diagram, the sample times are divided into equally spaced vertical lines 163, 165.
, 167 and 169. these lines 16
Each adjacent pair of 3-169 defines a baud interval. Signal 171 is a signal that matches the frequency and phase of the 400 Hz modulation signal and is shown to show the accurate timing by reconverter 67.

If the timing is accurate, samples taken at times corresponding to lines 163-169 will be equal in magnitude. Signal 173 is also shown;
It is 1/4 baud ahead of 1. A sample of signal 173 taken at a time corresponding to line 165 is
It will be noted that the size is larger than the sample taken at the time corresponding to . Thus, the condition that the magnitude of YQo is greater than or equal to the magnitude of YQ-1 is satisfied, and the timing error according to Equation 3 is equal to line 16.
Signal 175 is shown lagging 1/4 baud behind signal 171, as determined by the baud interval defined by 3 and 165.

A sample of signal 175 taken at a time corresponding to line 165 will have a smaller magnitude than the previous signal taken at a time corresponding to line 163. Under these conditions, calculating the timing error according to Equation 3 is less desirable. This is because the tangent function will be larger than the number of criteria. However, timing errors occur on lines 165 and 167.
is easily and accurately determined in the bow interval defined by . It will be noted that the sample taken at the time corresponding to line 167 has a larger magnitude than the sample taken at the time corresponding to line 165. Thus, the condition YQo≧YQ-1 is satisfied. As previously noted, sampling a properly timed signal, such as signal 171, provides equal magnitude samples YQo and YQ-1.

Thus, the quotient in Equation 3 has a nominal magnitude, and the arctangent of this quotient represents an angle of 45°. 9 in equation 3
Divided by the amount 0°, this is O. yielding a quotient of 5, and when this quantity is added to the constant -0.5 in Equation 3, it is shown that the timing error ΔT becomes O. The timing of receiver 29 is 0.25 as shown at 173.
Leading the baud signal 171, the tangent of YQ-1/YQo is equal to 22.5°.

Dividing this amount by 90° gives a quotient of 0.25. When this quotient is added to the constant -0.5, a timing error of -0.25 is indicated. This error instructs receiver 29 to retard its timing to correct the timing error. Receiver 29 receives O. If it is 25 baud behind, receiver 29 is instructed to retard its timing by 0.75 baud.

This is equivalent to an advance of 0.25 baud for a 400 hertz modulated tone 103. Equation 3 is implemented according to the configuration of circuitry 97, as shown in FIG.
Conductor 125 carries signal YQo to delay network 177 whose output is connected to terminal 145 of switch 141. Then, set switch 141 to the third position, and switch 141 to the third position.
With 47 in the first position, ARCTANYQ-1/YQ
A signal o is supplied to the output section of the ROM 157. This signal is passed to multiplier 179 along with two signals having a magnitude equal to 1/90. The resulting product is passed to the positive input terminal of summation network 181. This network receives the signal 0.5 at its negative input terminal. The resulting signal ΔT, shown in Equation 3 above, is provided on conductor 188 at the output of summing network 181. A signal having characteristics corresponding to the condition that the magnitude of YQ0 is equal to or greater than the magnitude of YQ-1 is connected to a third switch having first and second positions corresponding to terminals 185 and 187, respectively. j183.

Amount 1. A signal equal to is maintained at terminal 187. Conductor 1
25 conveys the signal YQo to a magnitude extractor 189, which provides the signal 1YQo1 to the positive input terminal of a comparator 191. This signal is also passed through a delay network 193 to provide a signal YQ-1 at terminal 18ζ5, which in turn is passed through switch 183 in the first position to the negative input terminal of comparator 191. It will be done. On conductor 195 at the output of comparator 191, signal 1YQol≧IYQ
-1I can be used in timing control circuitry 94 (FIG. 3) to activate a timing error signal on conductor 188. Once the timing is adjusted,
It is well known that the phase must also be modified to some extent in response to timing deviations.

The phase jump of this corresponding carrier wave is derived from the following equation. Phase error of reference carrier wave Δφ=-θ (△T)
(Equation 4) where ΔT is the change in timing by hundredths of a baud, derived from Equation (3), and θ is the number of reference carrier angles per baud.

For the specific example given, the carrier is 1600Hz
Since , 1 Hz per baud, and θ equals 360°, the phase jump of the reference carrier corresponding to the timing jump in this particular example is calculated as follows.

△φ=-360△T (Equation 5) A signal corresponding to Equation 5 is added to multiplier 1 by multiplying signal △T on conductor 188 together with a signal having a magnitude corresponding to -360°.
97.

The resulting product provides the signal Δφ of Equation 5 on conductor 199. This signal on conductor 199 is passed to phase control circuitry 95 to adjust the timing and phase of the receiver reference to the current state. Third period T4-T5
During this period, a first impulse 105 is transmitted into the prologue 99 .

In general, impulses such as impulse 105 are particularly useful for making fine timing and phase corrections. The carrier modulated tone 103 provides a typical double sideband signal, which represents information only in the higher and lower sideband frequencies, whereas the impulses represent frequencies over the entire frequency spectrum. have. The resulting impulses cause the adaptation of the receiver 29 to make more accurate channel measurements. A typical response of receiver 29 to an impulse, such as first impulse 105, is shown in FIG. 11 and designated by reference numeral 196.

Telephone line impulse response 96 by converter 67
The sampling of is indicated by a plurality of equally spaced vertical lines 198. These lines 198 are also designated by the associated symbol YQ. These signs Y
Q is a main code YQo, secondary codes YQ-1, YQ-2, and YQ-3 (generally called introduction codes), and YQ1
, YQ2 and YQ3 (generally called tracking codes). It has been found that the timing error ΔT is calculated using these sign values according to either of the following two equations: Equations 6 and 7 both give good approximations of the timing error for the impulse response across various channels.

Typical values for YQ1 and YQo are 12th for various timing errors between 0.25 and -0.25.
Shown in the figure.

For each timing error, the value of ΔT calculated according to equations 6 and 7 is also tabulated in FIG. It can be seen that Equation 6 probably gives the most accurate approximation of the timing error ΔT. On the other hand, Equation 7 gives a better approximation of the timing error ΔT and is easier to derive. Formula 7
ΔT is induced by placing switches 141 and 147 in their respective second positions.

The product YQ1/YQo at the output of multiplier 153 is passed to multiplier 201 with element -0.67. The resulting product provides the timing error signal ΔT described in Equation 7. This signal is routed on conductor 203 to converter 67.
can be conveyed to. When the first impulse 105 is received, it may also be desirable to correct any accumulated phase errors due to the frequency representation.

This error correction is accomplished in a manner similar to that discussed with reference to the method step indicated by block 112 of FIG. A signal equal to that expressed by Equation 1 is thus induced and transmitted on conductor 161 to phase control network 95. As previously mentioned, phase corrections are also conveniently performed based on timing jumps made according to equations 6 or 7.

For the preferred embodiment, this corresponding phase jump has a magnitude equal to Δφ=-360ΔT. If necessary, the phase jump associated with the new phase measurement and the phase jump associated with the timing jump are calculated according to the following equations. A signal representing Equation 8 is derived by passing the signal ΔT on conductor 203 to multiplier 205 with an element -360°.

This product, together with the signal Δφ on the conductor 161, is obtained by the summator 20
7 can be conveyed. The resulting signal, given by Equation 8 above, is conveyed on conductor 209 to phase control network 95. By modifying the phase and timing according to block 116 of FIG.
07, proceed to calculate the equalization coefficients. In order to correctly calculate the equalization coefficient from an impulse, the impulse must have periods of zero transmitted energy both before and after the impulse with sufficient guard time.
During this zero energy interval, the receiver 29 is not phase locked, so a phase error occurs due to frequency changes in the channel. If the protection period is 10 baud, the frequency change will be 10 hertz and the baud time will be 1
/1600sec=6250 microseconds. Thus, for example, a phase error of 2.25 degrees per baud would cumulatively give a total phase error of 22.5 degrees. Such a phase error causes an error of as much as 38% in the calculation of the equalization coefficient. If there is a phase error, it is the main code Y
It will affect not only Io, but also the slave codes, including the introduced samples YQ-1, YQ-2 and YQ-3.

All these signs are important in calculating the equalization coefficients.
However, the detection and calculation of the phase error according to Equation 1 is performed after the introduction pulse has passed. Therefore, the phase correction at that time is based on the main code and the sub tracking codes YQ1, YQ.
2. It will only affect YQ3. In the preferred embodiment of the invention, this problem is solved by storing previously received lead-in samples until the impulse's main sample YQo occurs.

By measuring the overall phase error according to Equation 1,
The receiver phase reference is adjusted to the current state in the manner described above. If this phase modification occurs during the current time interval, only the main and tracked samples will be affected by this phase modification. To correct the phase of the introduced sample, the accumulated 4
The values YIj and YQ are replaced by the corresponding quantities YI↑ and YQ'j' calculated according to the following equations.

Eight ? ? ? ▲-I 1A? ^――
◆A Here, j is greater than or equal to -3 and less than or equal to 0.

Following phase correction of both the introduction and tracking samples,
These signals, collectively referred to as Y↑ and YO, are normalized as related to the amplitude change according to the following equation:

Here, , is greater than or equal to -4 and less than or equal to -1.

The phase correction of the introduced samples is performed in the equalization coefficient calculator 211 shown in block diagram form in the quick learn 2 network 97 of FIG.

The dirt is shown in detail in FIG. quick learning circuit 97
, conductors 123 and 125 (b signals YIj and Y
Q, is transmitted to the calculator 211. In more detail, these signals are routed through the summation network 2 as shown in FIG.
]3 and 2]5. Circuit networks 213 and 215
The signals from are stored in storage registers 217 and 219, respectively. The sign YIJ from register 217 is sequentially transmitted to a pair of multipliers 3221 and 223. Similarly, the YQ3 code is passed to a pair of multipliers 225 and 227. A signal equal to sine Δφ, provided by the sine/cosine ROM, is also passed to multipliers 223 and 225. Similarly, 4 signals equal to cosine △φ are sent to the multiplier 2
21 and 227 with additional elements. Multiplier 223
and 227 are added in adder 229;
The signal YO3 of Equation 9 is supplied to the conductor 231. Multiplier 22
The product supplied by 1 and 225 is the differential adder 233
The signal Y in Equation 9: ↑. , the conductor 235
supply to. The signals on conductors 231 and 235 can be fed back to provide additional inputs to respective summation networks 215 and 213. When the mourning phases of both the introduction and tracking samples are corrected, Y7t) and Y0,
Both signals are passed to multipliers 237 and 239, respectively. In these multipliers 237 and 239, the signals are divided into elements 1/YO. Multiplied by o. The resulting product Y year 3 and Y<? 3 are stored in a pair of storage registers 241 and 243, respectively. After the introduction and tracking samples are phase corrected and normalized according to block 118 of FIG. 6, equalization coefficients are calculated.

The outputs of the LPEs 75 and 77 of the I and Q channels become the inputs of the equalization network 79 (FIG. 3),
J and XQ are expressed by the following equations. Here, DIj
and DQ are the distorted data symbols at the jth baud interval.

DI,-1 and DQ-1 are the distorted data symbols in the baud interval immediately before the jth baud.

hI-1 and hQ-1 are the values at the sample rate of the impulse response of the raised cosine characteristic of the distortion of the in-phase channel and the channel with a different phase of 90° in the introduction section, respectively, multiplied by the demodulation carrier, and further calculated for phase correction. is multiplied by the sine and cosine components of Δφ.

In equalization network 79, a transversal equalizer (not shown) is provided with a plurality of taps with variable coefficients CIj and CQj and multiplies by the code XI and XQ.

In the preferred embodiment, the transversal equalizer in equalization network 79 has ten taps and a corresponding number of equalization coefficients. For purposes of this discussion, these transversal equalizers have three taps CL1, CI-2, and CI-3 in the I channel and three taps CQ-1, CQ-3 in the Q channel. 2 and CQ-
Let's assume we only have 3. Let us further assume that the center tap CIo in the I channel is equal to 1.0 and that the center tap CQo in the Q channel is equal to O. Under these assumptions, the signal at the output of equalization network 79 can be expressed as: Equation 11 is Equation 12
The data and errors are correlated, and the coefficients for the intersymbol terms are solved to have a magnitude of 0 to minimize the effects of intersymbol interference. Following this procedure for the assumed number of taps, the coefficients CIJ and CQ
is adjusted to the following calculated value to equalize the XI and XQ signals. ~1)-b' - The above equation shows the main intersymbol interference hI-1 and hQ-
Applies when 1 is supplied.

A more general solution considering the introduction and 90° channel distortions hIK and hQK, respectively, is given by the following general solution. Here, hIK=YO, {all k\0} where hQ, = Y books {all k:O} These formulas are used in the equalization coefficient calculator 211 shown in FIG.
satisfied by a portion of

The phase-corrected and normalized signal 1/YO-1 is transferred to conductor 2.
47 is transmitted to the inverter 245. This signal is transmitted to multiplier 249 and multiplier 25
It is squared by 1. Its output is connected to the positive input terminal of summation network 253. The phase corrected, Cali-normalized signal Y+-1 is also passed to multiplier 249 and squared in multiplier 255, the output of which is connected to the negative input terminal of summation network 253.

The product from multiplier 249 is multiplied by a quantity of 2 in multiplier 257 and its output is connected to the positive input terminal of summation network 259 . The quantity Y+-2 from register 241 is passed to the positive input terminal of summation network 259. Similarly, the quantity Y6-2 from the register 243 is
is transmitted to the negative terminal of Signal Y Imaichi, is register 2
41 directly to the register 261, and the equalization coefficient CQ
-1 is supplied.

The sign 1/YO-1 on conductor 247 is conveyed to register 263 to provide equalization coefficient CI-1. Satellite circuit network 25
The output of 3 is connected to the register 263, and the output of equalization coefficient CI
-2 is supplied. Similarly, the output of summation network 259 is connected to register 261 to provide equalization coefficient CQ-2.
~ The output of summation network 259 is also transmitted to multiplier 265 and multiplier 267.

The sign Y now -1 provides an additional element to the multiplier 267;
The resulting product is passed to the negative terminal of summation network 269. This circuit network 269 also receives the symbol Y0-3 at its negative power terminal. The outputs of summation network 253 and conductor 247 are connected to multiplier 27 and the resulting product is communicated to the positive input terminal of summation network 269. This circuit network 26
The output of 9 is transmitted to the register 263 and is equalized by the equalization coefficient CI-3.
supply. The signal on conductor 247 is conveyed to multiplier 265 and the resulting product is conveyed to the positive input terminal of summation network 273.

This network 273 also receives at its positive input the sign Y+-3.The output of the circuit 253 is combined with the sign Y+-1 in the multiplier 225 and the product of the result is added to the summation circuit. to the positive input terminal of network 273. The output of network 273 is connected to register 261 and provides equalization coefficient CQ-3. As described above, Equation 14 is satisfied, providing equalization coefficients to registers 261 and 263 that are passed to equalization network 79 on respective conductors 277 and 279.

These coefficients are combined with signals X1 and XQ as shown in Equation 12 to provide equalized signals Y and YQ''. With this adjustment, the receiver 29 becomes ready to receive data.

It will be appreciated that the rapid learning circuitry 97 and method of the present invention is particularly advantageous because it adapts the receiver 29 to the characteristics of the telephone line 23 in a minimum time of less than 30 milliseconds. This very short time is particularly advantageous during interrogation of multipoint communication systems, since as many as 20 slave stations can be interrogated in a period of less than one second.

This significantly increases the data yield of the system.
Although the invention has been disclosed with respect to particular method steps and apparatus for accomplishing the method, the method may include other steps and the apparatus may be embodied in other ways. This will be clear to those skilled in the art.

The gist of the present invention is as follows.

1. the first and second channels and have properties that adapt to the characteristics of the transmission line into which the modulated signal is transmitted and from which an incoming signal having analog characteristics and a first specific phase is received; A receiver characterized in that it includes the following steps: first means for sampling an incoming signal to provide a sampled signal having digital characteristics and a first specific phase; second means for demodulating the sampled signal for providing a first component of the demodulated signal on one channel and a second component of the demodulated signal on a second channel; receiving; the receiver has adjustable characteristics for providing a second specific phase and a phase error that depends on the difference between the first specific phase of the incoming signal and the second specific phase of the receiver. Phase correction means.

third means for dividing the first component of the demodulated signal by the second component of the demodulated signal to form a quotient; dependent on an angle whose trigonometric function is substantially equal to said quotient; and a second particular phase of the receiver having a magnitude substantially equal to the first particular phase of the incoming signal; (phase modification means responsive to said particular signal) to provide a phase error having a magnitude of zero. 1, in which the particular signal has the property of being angle-dependent, the sine function of the angle being substantially equal to said quotient {
．． 2, wherein the particular signal has angle-dependent properties whose angular tangent function is substantially equal to said quotient; i. In the receiver described in 1, the modulated signal sent to the transmission line is
a first component associated with a first channel of the receiver and a second component associated with a second channel of the receiver, the first component of the modulated signal having a magnitude of substantially zero; and said quotient is obtained by dividing the signal in the first channel of the receiver by the signal in the second channel of the receiver.
a third means formed into the first and second channels and into which the modulated signal is fed and into which an incoming analog signal having first timing characteristics is received; A receiver characterized by adapting to the characteristics of the transmission line, comprising the steps of: providing a sampled signal with digital characteristics and adapting the first timing characteristics of the incoming signal to the sampling means; means for sampling the incoming analog signal according to the second timing characteristic to provide the receiver with a timing error that depends on the difference between the plurality of first digital symbols; and a plurality of second digital codes in a second channel, the first and second digital codes both including a current code and a preceding code. means for demodulating the sampled signal to provide a quotient; means for dividing the preceding symbol in the first channel by the current symbol in the first channel to produce a quotient; and means for providing a particular signal having an angle-dependent characteristic whose trigonometric function of the angle is substantially equal to said quotient; 6. sampling means responsive to the magnitude of said particular signal to provide a timing characteristic of said second and to provide a substantially zero magnitude to the receiver timing error; 6. 7. The receiver described in item 5, in which the trigonometric function is a tangent function. 5
, wherein the modulated signal transmitted to the transmission line has a first component associated with a first channel of the receiver and a second component associated with a second channel of the receiver. 8. a first component of the modulated signal associated with the first channel of the receiver has a magnitude substantially equal to zero; 5, wherein the sampling means has the characteristic of sampling the incoming analog signal once during each of a plurality of baud intervals, each of which has a period equal to a particular angle. , further comprising: means for providing a signal representative of a second quotient formed by dividing said particular angle by said particular angle; and said sampling means: Responsive to the signal representative of the second quotient to adjust the conversion means by an amount dependent on the magnitude of the second quotient so as to provide a substantially zero magnitude to the timing means of the receiver.

9. A receiver having the property of being responsive to impulses transmitted to the receiver over a transmission line in order to minimize the effects of interchannel and intersymbol interference during the transmission of successive data symbols; means for sampling the impulses to provide a series of digital samples; providing a series of digital symbols on in-phase channels of the receiver and a series of digital symbols on 90° out-of-phase channels of the receiver; means for demodulating the digital samples to provide a digital signal; equalization means responsive to a plurality of variable coefficients for combining the digital code with the variable coefficients so as to adjust the relative magnitude of the digital code; digital codes in at least one of said in-phase and 90° out-of-phase channels in order to induce at least one specific signal having the property of adapting the coefficients of the equalization means to the current state. means for responding; and means for equalizing:
10. responsive to said particular signal for varying the coefficients of the equalization means in such a way as to minimize the effects of inter-channel and inter-symbol interference; 9, in which the series of data samples in the channel that are in phase and the series of digital samples that are 90° out of phase are each composed of a master sample and a plurality of slave samples that are ahead of the master sample in time. The means for including and inducing the sample and a plurality of secondary samples that are behind the main sample in time is responsive to the secondary samples that are ahead of the main sample in time to induce the specific signal.

11.9, in which the equalization network consists of the main coefficients associated with the in-phase channels and the phase 9
the impulses are transmitted to the receiver on one of the in-phase channels and the 90° out-of-phase channels; The dominant coefficient in one of the phase channels has a fixed magnitude of substantially zero, the in-phase channel and the 90° out-of-phase channel? The principal coefficients in the other have a fixed magnitude of substantially -.

12. A receiver having in-phase channels and 90° out-of-phase channels, adapted to receive data signals having phase errors as well as intersymbol and interchannel interference, and including: main sample, main sample, means for sampling the received signal to provide a digital signal including a plurality of lead-in samples leading the samples and a plurality of tracking samples trailing the main sample; in-phase channels and 90° out-of-phase channels; A second means for demodulating the digital signal to provide a demodulated signal having a sign associated with the main sample and the slave sample in the channel; a third having variable characteristics for phase modifying the demodulated signal in the in-phase channel and the 90° out-of-phase channel to provide a phase modified signal having a sign associated with the sample and slave sample; means for equalizing the phase corrected signals in the in-phase channel and the 90° out-of-phase channel to provide an equalized signal having a sign related to the main sample and the slave sample; A fourth means having the characteristics,
and fifth means responsive to the phase modified signal if the received signal has impulse characteristics for changing the characteristics of the fourth means to minimize intersymbol and interchannel interference. 13. 12, wherein the third means comprises: responsive to at least one of the signs of the demodulated signal relative to the main samples of the digital signal to calculate a phase error; seventh means for storing the code of the demodulated signal associated with the introduced samples of the digital signal; and a seventh means for modifying the phase of the code stored by the seventh means. eighth means 14 responsive to the phase error calculated by the means;
．． 12, wherein the fourth means comprises: for providing a normalized signal having a sign associated with the primary and secondary samples; for normalizing the phase corrected signal; seventh means responsive to the sign in the phase modified signal associated with the main samples of the digital signal; Eighth means with variable characteristics for equalizing normalized signals in channels with 90° phase difference; 15. 13, wherein the sixth means is configured to calculate a phase error in one of the in-phase channels and the 90° out-of-phase channels associated with the main samples of the digital signal. formed by dividing the sign of the demodulated signal by the sign of the demodulated signal in the other of the in-phase and 90° out-of-phase channels associated with the main samples of the digital signal. which responds to the arctangent of the quotient

[Brief explanation of the drawing]

FIG. 1 is a block diagram of a point-to-point communication network including a master station communicating via telephone lines with a plurality of slave stations, each including a transmitter having the features of the present invention. FIG. 2 shows a block diagram of one of the transmitters shown in FIG.
This is a diagram. FIG. 3 is a block diagram of one of the receivers shown in FIG. 1, including rapid learning circuitry to quickly adapt the receiver's timing, phasing, and equalization to the characteristics of the telephone line. I'm here. Figure 4 shows
4 is a block diagram of the rapid learning network shown in FIG. 3; FIG. FIG. 5 is a rapid learning prologue for use with the rapid learning network of the present invention. FIG. 6 is a side chart illustrating the preferred method of the present invention. Figure 7 shows
3 illustrates the modulated tones used with the initial phase and timing corrections of the preferred method; FIG. 8 is a phase diagram showing the effect of a 60° phase jump in response to a phase error greater than 45°. FIG. 9 is a table showing the maximum phase error when a 60 degree jump is made in response to a phase error greater than 45 degrees. FIG. 10 is a block diagram illustrating the effects of timing modification in accordance with the preferred method of the present invention. FIG. 11 is an impulse response showing large and small samples generated by the converter in the receiver. FIG. 12 is a table showing timing errors calculated in accordance with the preferred method of the present invention. FIG. 13 shows an equalization factor calculator for adjusting equalization factors in accordance with the preferred method of the present invention.
11... Communication circuit network between other points, 13... Main station, 15
, 17, 19... slave station, 21, 23... telephone line, 25... office machine, 27... transmitter, 29...
Receiver, 33... Transmitter, 35... Receiver, 37.
... Encoder, 39... Inhibitor 41, 43...
·Digital. Low pass. Filter, 45...Sign/
Cosine ROM147...Adder 49...Converter, 51...Analog. Low pass. filter, 55
...Start permutation generator, 57...Timer 59.
...Enabling gate, 63...Analog. filter, 6
5...Brain gain control, 67...Converter, 69,7
1... Multiplier, 73... Sine/cosine ROM,
75, 77...low pass. Filter, 79... Equalization circuit network, 81... Phase correction circuit network, 83... Detection circuit network, 87... Error calculator, 89... Decoder, 9
1... Equalization control circuit network, 93... Decision threshold control, 9
4... Timing control circuitry, 95... Phase control network, 97... Rapid learning circuitry, 99... Rapid learning preamble section, 101... Unmodulated carrier wave, 103
...Tone, 105...Impulse, 107...
Impulse,]09...Pulse gain control, 1]0...Large phase jump, 112...Small phase jump, 114...Timing and phase jump, 11
6... Timing and phase jump, 118... Phase correction, 120... Equalization adjustment, 123, 125...
Conductor, 127... Multiplier, ] 29... Comparator, 13
1... Multiplier, 133... Conductor, 135... Size extractor, 137... Comparator, 139... Conductor, 1
41...Switch, 147...Switch, 153.
... Multiplier, 155 ... Impurse ROM1 157
...ARCTANR0M, 159...Inverter,
161... Conductor, 179... Multiplier, 181...
Summation circuit network, 183... Switch, 189... Magnitude extractor, 191... Comparator, 193... Delay circuit network, 197... Multiplier, 201... Multiplier, 205
... Multiplier, 207 ... Multiplier, 211 ... Calculator, 213, 215 ... Summation circuit network, 217 ... Register, 221, 223 ... Multiplier, 225, 22
7... Multiplier, 229... Adder 233... Differential adder, 237, 239... Multiplier, 241, 24
3...Storage register, 245...Invar 249
. . . Multiplier, 251 . . . Multiplier, 253 . . . Sum seeker circuit network, 257 .
267... Multiplier, 269... Sum seeker network, 27
1...Multiplier, 273...Summing circuit network.

Claims (1)

[Claims] 1 a sampling means 67 for receiving an incoming input analog signal and providing a sampling signal, and a sampling means 67 for providing a first component of the demodulated signal on a first channel and a second component of the demodulated signal on a second channel. for supplying the ingredients of
Demodulation means 69, 71, 7 for demodulating the sampled signal
3, and calculation means 1 for dividing the first component of the demodulated signal by the second component of the demodulated signal to form a quotient.
53, 155, and an inverse trigonometric function calculating means 15 for supplying a specific signal having characteristics depending on the value of the inverse trigonometric function of the arctangent or arcsine of the quotient.
7, and phase correction means 81, 95 responsive to a specific signal provided by the inverse trigonometric function calculation means 157 to provide a phase error for the incoming input analog signal.
and the modulated signal sent to the transmission line is
a first component associated with a first channel of the receiver and a second component associated with a second channel of the receiver, the first component of the modulated signal having a magnitude of substantially zero. and the quotient is calculated by dividing the signal in the first channel of the receiver by the signal in the second channel of the receiver.
Characteristics of the first and second channels formed in the calculation means 153, 155 and the transmission path through which the modulated signal is transmitted and the incoming signal is received; A receiver having an adaptive digital modem, characterized in that the receiver is adapted to a digital modem.