JPS6129233B2 - - Google Patents

Description

[Detailed description of the invention]

When driving an induction motor using a frequency converter, the present invention enables operation at high rotational speed even if there is a limit to the output frequency of the frequency converter, and also suppresses fluctuations in the motor's main magnetic flux due to changes in load current. The present invention relates to a control device for an induction motor. Generally, when an AC motor is driven by a frequency converter, the rotational speed N of the motor depends on the output frequency of the converter. Usually, the rotational speed N is operated at a synchronous speed given by the following equation or less. N=120/p (1) where p: number of poles On the other hand, the output frequency of a converter usually has an upper limit depending on its characteristics. For example, in a cycloconverter that inputs a constant frequency voltage and converts it into a variable frequency sinusoidal voltage, the upper limit of its output frequency is limited to about 1/3 of the power supply input frequency or less. Therefore, the rotational speed is also limited according to the above-mentioned relationship, and if the load device requires even higher rotation speed, it will not be possible to meet that request. One way to solve this problem is to use a motor with a polyphase primary winding and a secondary winding, so that the current flowing through both windings is in opposite phase to each other, so that the speed is twice the synchronous speed. It has been known to make driving possible. However, this method basically has the characteristic that the main magnetic flux of the motor fluctuates as the load current changes, and the shunting characteristic (motor magnetic flux (motor voltage) does not change according to the load current). It cannot be applied to rolling mill drives that require the following characteristics. Therefore, in order to solve this problem, the present applicant first proposed Japanese Patent Application No. 1122/1983. The electric motor used for this has multiphase windings on each of the stator and rotor sides, but because the stator side is easier to cool than the rotor side, the windings on the stator side If the number, that is, the amount of magnetomotive force is made larger than that on the rotor side, the electric motor can be made smaller. The present invention has been made in response to the above-mentioned problems, and its purpose is to reduce the amount of magnetomotive force used to drive a rolling mill when there is a difference in the amount of magnetomotive force between the windings on the stator side and the rotor side. It is an object of the present invention to provide a control device for an induction motor that can obtain shunt characteristics suitable for the following applications. The present invention is characterized by a motor having primary and secondary windings having different amounts of magnetomotive force, the windings being connected in opposite phase to each other, and a variable frequency current being supplied to the motor. In a motor control device constituted by a frequency converter, a current control circuit for controlling the magnitude and phase of the output current of the converter, etc., the magnetomotive force vector generated by the primary winding and the secondary winding, Also, when the phase angle between the composite vector and the primary magnetomotive force vector is α, and the phase angle between the composite vector and the secondary magnetomotive force vector is β, the primary
The purpose is to control the phase of the output current of the converter in accordance with the sum of both phase angles α and β such that the magnitude of the composite vector does not change even if the magnitude of the secondary magnetomotive force changes. FIG. 1 shows a configuration diagram of an embodiment of the present invention. 1
is a cycloconverter (hereinafter referred to as CYC) that inputs AC from a commercial AC power supply (3φ AC) and outputs variable frequency three-phase AC, and consists of thyristor pure bridge circuits U _P , U _N , which are connected in antiparallel. V
It consists of three sets: _P , V _N and W _P , W _N. 2 is the 3-phase primary winding U to W (stator side) and the secondary winding
An induction motor with U'~W' (rotor side), 2
The secondary winding is connected in series with the primary winding via a slip ring (not shown). 3 is a speed command circuit;
4 is a speed detector that detects the rotational speed of the electric motor 2;
5 is a speed deviation amplification device that amplifies the speed command signal from the speed command circuit 3 and the output signal of the speed detector 4; 6 is the output signal of the speed deviation amplification device 5 and the magnetic flux command from the magnetic flux command circuit 7; an arithmetic circuit that outputs a current value command signal and a current phase command signal based on 8;
1 is a position detector that outputs a three-phase sine wave position signal whose phase corresponds to the rotation angle of the rotating shaft of the electric motor 2; 9 is a phase shifter that shifts the phase of the position signal according to a current phase command signal; 10 is a current value command A multiplier that multiplies the signal by the output signal of phase shifter 9 and outputs a current command signal (sine wave signal) for controlling the output current (U phase) of CYC1 _; A _current detector 12 detects the output current of the current detector 11; 12 a current deviation amplification device that compares and amplifies the current command signal and the output signal of the current detector 11; 13 a current deviation amplification device 12;
U phase ( _UP , U _N ) of CYC1 according to the output signal of
14 is a gate output circuit that supplies a gate signal to U _P or U _N depending on the direction of the output current of U _P or U _N . Although only the control circuit for the U phase of CYC is shown in the figure, there are control circuits similar to those 9 to 14 described above for other phases as well. The description of them will be omitted. FIG. 2 shows a detailed configuration diagram of the arithmetic circuit 6, with the arithmetic circuit shown within the broken line. 15 is a multiplier for squaring the output signal of the speed deviation amplification device 5; 16 is a multiplier for squaring the current value command signal I _P (to be described later); and 17 is a multiplier for squaring the output signal of the speed deviation amplifier 5; 18 is a multiplier that squares the output signal of the multiplier 17; 19 is a subtracter that subtracts the output signal of the multiplier 15 from the multiplier 16; 20 is a multiplier having an amplification rate equal to the magnetomotive force ratio; This is a subtracter that subtracts the output signal of the multiplier 15 from the multiplier 18. 21 and 22 are subtracters 19,
A square root circuit that takes out the square root of the output signal of 20;
23 is an adder that adds the output signals of both square root circuits; 24 is an adder that compares the output signal of adder 23 with the magnetic flux command, and outputs a current value command signal I _P that provides a relationship such that the former signal matches the magnetic flux command. 25 is an inverse sine circuit that outputs a signal having an inverse sine function relationship with respect to the value obtained by dividing the output signal of the speed deviation amplifier 5 by the current value command signal I _P ;
Similarly, 26 is an inverse sine circuit that outputs a signal having an inverse sine function relationship with respect to the value obtained by dividing the output signal of the speed deviation amplifier 5 by the output signal of the amplifier 17, and 27 is an inverse sine circuit of both inverse sine circuits. This is an adder that adds output signals. Next, the operating principle of this embodiment will be explained using the vector diagram shown in FIG. Figure 3 shows the primary winding magnetomotive force F ₁ and the secondary winding magnetomotive force when three-phase sinusoidal currents are passed through each of the primary and secondary windings so that they are in opposite phases. The relationship between magnetic force F ₂ is shown. Each magnetomotive force becomes a circular magnetic field that rotates at a constant speed because each winding is excited by a three-phase sinusoidal current. Now, suppose that at time t=0, F ₁ is at position O in the figure and F ₂ is at position O' in the figure. At that time,
Rotor (secondary side) due to electromagnetic force due to F ₁ and F ₂
receives torque and rotates clockwise. Next, if we look at the state after t seconds after time has elapsed,
As shown, F ₁ is advanced by ω _H t (ω _H : excitation angular frequency) in electrical angle. On the other hand, the rotor advances by ω _r t in electrical angle. Further, F ₂ advances by ω _r t−ω _H t since the secondary side is excited in the opposite phase. By the way, by setting ω _H to 1/2 after ω _r , F ₂ advances by ω _r t−ω _H t=ω _H t (2). This relationship can be realized by using the position detector 8 whose output signal angular frequency is 1/2ω _r . As described above, the phase relationship between F ₁ and F ₂ remains unchanged from time t=0, torque is generated continuously, and the rotor continues to rotate. Looking at the relationship between ω _H and rotational angular velocity ω _r /p (p: number of pole pairs), the rotational angular velocity is 2·ω _H /p. That is, even if the output frequency of the frequency converter is constant, the rotation speed can be doubled compared to the conventional one. On the other hand, the main magnetic flux φ of the electric motor is given by the following when the angle formed by F ₁ and F ₂ is α+β, as shown in the vector diagram of FIG. φα=F ₁ cosα+F ₂ cosβ (3) In addition, in FIG. 4, F ₁ >F ₂ . Here, when considering the case where the phase angles α and β are fixed, φ has a characteristic proportional to F ₁ and F ₂ , that is, the primary current I ₁ . This has a direct winding characteristic as in the conventional case, and cannot be applied to rolling mill drives, etc. However, if α and β are controlled according to I ₁ ,
It is also clear from equation (3) that φ can be kept constant regardless of I ₁ . The relationship that satisfies this condition is that in the triangle shown in Figure 4, the size of the two sides F ₁ and F ₂ (F ₁ and
Even when F ₂ is a constant ratio) changes, the other side φ (F ₁ +F ₂ ) is kept constant. The torque generated by the electric motor is T (=
Based on the torque command signal, _{a signal proportional to F 1 cosα} and F ₂ cosα is calculated, and a _linear signal is generated so that the sum is controlled to a predetermined value. Control the current I ₁ (F ₁ , F ₂ ) and simultaneously
The phase changes α and β are calculated based on the I ₁ command signal and the torque command signal to control the current phase (phase difference between F ₁ and F ₂ ). The above is the principle of the present invention. Next, the operation of the above embodiment will be explained. A signal for commanding the torque of the electric motor (hereinafter referred to as torque command signal τ _P ) is taken out from the speed deviation amplifier 5. The multiplier 15 takes out a signal proportional to the square of the signal τ _P , and calculates the difference between it and the signal proportional to the square of the current value command signal from the multiplier 16. Then, a signal proportional to F ₁ cosα is taken out from the square root circuit 21. On the other hand, the amplifier 17 extracts a signal of a predetermined ratio (k times) to the signal I _P . The ratio k
is matched to the ratio of F ₂ and F ₁ . In the multiplier 18, the subtracter 20, and the square root circuit 22, a signal proportional to F ₂ cosβ is extracted in the same manner as described above. The adder 23 adds signals proportional to F ₁ cosα and F ₂ cosβ, and extracts a signal proportional to φ shown in FIG. 4. The magnetic flux deviation amplification device 24 amplifies the deviation between this signal and the magnetic flux command and outputs a current value command signal I _P . The signal I _P is input to the multiplier 16 as described above.
Since a negative feedback loop is formed in which the signal is fed back to the amplifier 17, the output signal of the adder 23 always matches the magnetic flux command. On the other hand, the inverse sine circuits 25 and 26 perform the calculation shown in the following equation to generate a signal α proportional to the phase angles α and β.
_P and β _P are extracted.

[Table] Here, τ _P : The magnitude of the torque command signal I _P : The magnitude of the current value command signal Then, the current phase command signal which is the sum of the signals α _P and β _P is taken out from the adder 27. Next, the overall operation will be explained using FIG. The current of the motor is controlled in proportion to the signal I _P according to the operation of a current control circuit comprised of multipliers 10 and below. Further, its frequency is controlled according to the signal from the position detector 8. Note that the angular frequency ω _H of this signal is 1/2 of the rotating electrical frequency ω _r
is set to . Further, the phase of the current is controlled by a phase shift circuit 9 according to a current phase command signal. In that case, if _the reference position detection signal is phase-shifted by _θ , the phase difference between F ₁ and F ₂ will change by 2θ, so the phase shift circuit 9 The position detection signal is expressed as (α+
The phase is shifted by β)/2. As a result of controlling the magnitude and phase of the current as described above, the magnitude of the torque is controlled while the main magnetic flux amount φ remains constant. At this time, the magnitude of the induced voltage in the primary winding and the secondary winding is kept constant regardless of changes in current. However, since the phase difference between each induced voltage changes to some extent depending on the current, the motor voltage, which is the vector sum of both induced voltages, exhibits characteristics that change slightly depending on the current. If it is necessary to control the motor voltage to a more uniform voltage, a signal related to the current can be added to the magnetic flux command from the magnetic flux command circuit 7 to eliminate fluctuations in the motor voltage. Therefore, according to the present invention, in the above-described motor control device capable of operating at a rotational speed twice the synchronous speed determined by the output frequency of the frequency converter, the primary winding and the secondary winding of the motor are Even when there is a difference in the magnetomotive force of the windings, it is possible to provide a motor control device that can obtain shunting characteristics and is suitable for driving a rolling mill, etc. Note that the present invention is not limited to using a CYC as a frequency converter, and similar effects can be obtained using other types of converters. Further, the number of phases of the winding of the motor is not limited to three, but may be other multiphase.

[Brief explanation of the drawing]

Fig. 1 is a block diagram of a motor control device showing an embodiment of the present invention, Fig. 2 is a detailed circuit block diagram of the circuit elements shown in Fig. 1, and Figs. 3 and 4 explain the operation of this embodiment. This is a diagram for 1...Cycloconverter, 2...Electric motor, 3
... Speed command circuit, 4 ... Speed detector, 5 ... Speed deviation amplifier, 6 ... Arithmetic circuit, 7 ... Magnetic flux command circuit, 8 ... Position detector, 9 ... Phase shifter, 10 …
... Multiplier, 11 ... Current detector, 12 ... Current deviation amplifier, 13 ... Automatic pulse phase shifter, 14 ...
Gate output circuit.

Claims (1)

[Claims] 1. An induction motor in which a polyphase primary winding and a secondary winding are connected in opposite phases, a position detector that detects the rotational position of the induction motor, and a variable frequency alternating current connected to both windings of the induction motor. one frequency converter that commonly supplies current; a speed control circuit that outputs a torque command signal according to the deviation between the speed command signal and the speed feedback signal;
a current calculation circuit that outputs a current command signal and a current phase command signal according to the magnetic flux command signal that determines the torque command signal and the terminal voltage of the induction motor; and a phase shifter that shifts the phase according to the frequency converter, and the phase of the output current of the frequency converter is based on the phase shift position signal obtained from the phase shifter, and the frequency is adjusted so that the magnitude is proportional to the current command signal. converter control means for controlling the converter, and the current calculation circuit calculates a phase angle between a composite vector of magnetomotive force vectors generated by the primary winding and the secondary winding and the primary magnetomotive force vector by α and When the phase angle between the composite vector and the secondary magnetomotive force vector is β, the magnitudes of the primary and secondary magnetomotive forces are both in phase so that the magnitude of the composite vector does not change even if the torque command signal changes. A control device for an induction motor, characterized in that a current phase is determined according to the sum of angles α and β.