JPS625489B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to a gain control device using transistors, and particularly aims to provide a device having a configuration suitable for equipment operating at low voltage.

Generally speaking, it is desirable for a gain control device to have no change in the DC operating point of the amplification transistor due to the gain control operation, and to be able to maintain the amplification characteristics of the device linearly regardless of its advantages. As long as there is a sufficient dynamic range, a relatively large change in the DC operating point can be tolerated. However, when the circuit power supply voltage is as low as a few volts, such as in battery-powered equipment, changes in the DC operating point not only significantly limit the gain control range but also greatly change the frequency characteristics, which impairs the performance of the equipment. This may cause deterioration. As an example of a conventional circuit related to the present invention,
This is described in Publication No. 15355 "Variable Gain Control Circuit".

This is done by transistors 1 and 2 as shown in Figure 1.
It consists of a differential amplifier circuit consisting of a constant current source 3 and a variable impedance circuit consisting of diodes 5 and 6 and a constant current source 4. These are driven by a signal current source 7, and the constant current sources 3 and 4 The gain is controlled by changing the current ratio between the In order to prevent the output DC level of the circuit from changing due to gain control, the current value of the current source 3 is fixed and only the current source 4 is made variable.

Current transmission efficiency of this circuit η ₁ (input signal current i
The ratio of output signal current i _p to _s is calculated as follows: η ₁ = i p /i, where the current amplification factors of transistors 1 and 2 are h _e and the current values of constant current sources 3 and 4 are I ₁ and I ₅ , _respectively . _s = 1/(1/h _e +I ₅ /I ₁ )
...(1), and under the condition of 1/h _e <<I ₅ /I ₁ , it can be expressed as η ₁ I ₁ /I ₅ ...(2). Therefore, in order to obtain a desired gain control range, a corresponding current change must be applied to the constant current source 4. By the way, the normal signal current source 7 is composed of a resistor 12, a signal voltage source 11, and a DC voltage source 10 as shown in FIG. 2, and by setting the resistance value of the resistor 12 sufficiently large compared to the load side impedance. Although it is practical, if the gain is controlled by changing the current source 4, the current change amount Δ of the current source 4
The voltage change caused by the product of I and the resistance value of the resistor 12 shown in FIG. 2 results in a change in the base bias of the transistors 1 and 2 shown in FIG. For example, if the resistance value of the resistor 12 is several kilohms, a current change of 1 mA results in a base bias fluctuation of several volts, so it cannot be used when the power supply voltage _Vcc is as low as several volts. Also, large variations in base bias are caused by transistors 1 and 2 in Figure 1.
This results in a large change in the collector-base reverse bias of the transistor, which changes the transistor junction capacitance and causes a change in frequency characteristics.

As mentioned above, in a gain control device, it is desirable to keep the frequency characteristics constant, and also to prevent the operating point of the amplification transistor from changing due to the gain control operation in order to facilitate DC coupling with the next stage. It is rare.

The present invention aims to eliminate the above-mentioned conventional drawbacks by providing a gain control device in which the change in the operating point of an amplifying transistor is small, and will be described in detail below with reference to the drawings.

FIG. 3 shows a basic configuration diagram of the present invention, and parts having the same functions as those in FIG. 1 showing the conventional example are given the same reference numerals. 1, 2 in Figure 3
3 is a transistor whose base electrode is a differential input terminal and whose emitter electrodes are commonly connected to form a differential amplifier circuit; 3 is a constant current source having a current value I ₁ connected to the emitter connection point of the transistors 1 and 2; 7 is a signal current source connected to the differential amplifier circuit input terminal, 8 and 9 are load resistors, and 13 and 14 are respective emitter electrodes connected to the differential amplifier circuit input terminal (bases of transistors 1 and 2). A transistor 15 is connected to the base electrodes of 13 and 14, and a transistor 15 is connected to the base electrodes of 13 and 14, and the transistor 15 is connected to the base electrodes of 13 and 14, and has its collector electrodes commonly connected and its base electrodes commonly connected to form a variable impedance circuit between the input terminals. A constant current source with a current value to be controlled I ₂ , 1
Reference numerals 6 and 17 are impedance circuits connected between the differential input terminals, both of which have an impedance value of _Z2 .

In this configuration, the gain is controlled by changing the current value of the constant current source 15, thereby controlling the gain of the transistor 1.
This can be achieved by changing the impedance between the collectors 3 and 14 and by changing the shunt ratio between the signal current i _s and the input signal current of the differential amplifier circuit. The impedance circuits 16 and 17 control the variation in the current amplification factor _{h e} of the differential amplifier stage transistors 13 and 14 when the gain control device is at maximum gain (when I ₂ = 0 and the impedance of the variable impedance circuit is large enough to be considered infinite). This is to reduce the variation in maximum gain due to These matters will be explained in detail next.

First, if the collector-emitter saturation voltage of a junction transistor is V _CE(sat) , then Therefore, the impedance Z ₃ in the saturation region is becomes. where k: Boltzmann constant, q: electronic charge, T: absolute temperature, α _N : forward common base current amplification factor, α _i : reverse common base current amplification factor, r
_e : emitter bulk resistance, r _sc : collector bulk resistance, I _c : collector current, I _B : base current, I
_E : Emitter current. In equation (4), the bulk resistance components of the differential values r _e and r _sc of the second and third terms in equation (3) are omitted because they are much smaller than the first term component in equation (4). Also, α _N /(1−α _N ) is the emitter grounded current amplification factor h _e , so equation (4) is This is the CE-to-E impedance of one transistor in the variable impedance circuit.

On the other hand, the input impedance Z ₁ of transistor 1 of the differential amplifier circuit is Z ₁ = r _bb1 +k _T /q h _e /I _E1 k _T /q
h _e /I _E1 ...(6). Here r _bb1 : Base spreading resistance, I _E1 :
This is the emitter current of transistor 1. By the way, in FIG. 3, since differential input signals with opposite positive and negative polarities are supplied to the input terminals, the emitter connection point of transistors 1 and 2, the collector connection point of transistors 13 and 14, and the connection point of impedance circuits 16 and 17 are virtual. Since it can be regarded as a grounding point and the circuit is symmetrical with respect to the virtual grounding point, the explanation will proceed based on only one side.

The signal current transmission efficiency η, which is the ratio of the output signal current to the input signal current of the gain control device according to the present invention
₂ , if the input signal current is i _s and the output signal current of transistor 1 is i _p , then i _p is Z ₃ and Z ₁ in equations (5) and (6).
and h of the input signal shunted by Z ₂ in Figure 3.
Because it is _e times , and substituting equations (5) and (6), we get becomes. Furthermore, since the collector of the transistor 13 is open or connected to the power supply voltage _Vcc line with high resistance, the collector current _Ic in equation (8) is almost zero, and can be simplified as follows.

Furthermore, h _e ≫1/(1−α _i ), and I _E1 = I ₁ /
2, Since I _B = I ₂ /2, finally η ₂ = 1/{1/h _e +2V _T /I ₁ 1/Z ₂ +I
₂ /I ₁ (1/1-α _i )}... (10). Here, V _T =k _T /q.

The second term 2V _T /I ₁ ·1/Z ₂ in equation (10) is an effective term for reducing variations in maximum gain, which will be described later, and is set as a constant C ₁ . Here, 1/h _e +C ₁ ≪I ₂ /I ₁ (1
/1-α _i ), equation (10) becomes as follows.

η ₂ 1/I ₂ /I ₁ (1/1-α _i )=I ₁ /I ₂ k
...(11) Here, k=(1/1-α _i ), and k is larger than 1 and usually has a value of about 3. Therefore, equation (11) is replaced by equation (2) η, which indicates the transmission efficiency η ₁ in the conventional example.
Comparing ₁ = I ₁ /I ₅ , I ₁ is a constant current, I ₂ , I ₅
It can be seen that in the present invention, a control current of 1/k is sufficient to obtain the same transmission efficiency when is used as the control current for controlling the gain. This means that the base bias fluctuations of transistors 1 and 2 in FIG. 3 are 1/k compared to that in the conventional example shown in FIG. This greatly improves the stability of characteristics.

By the way, gain control devices are often used built into AGC (automatic gain control) circuits, and in that case, if there is a large variation in the maximum gain of the gain control device, the gain control range of the AGC circuit may vary. This causes variations in In other words, the minimum input signal level that can be controlled to the same output level as an AGC circuit will vary greatly. The second term in equation (10) is used to reduce this variation, and now in equation (10), when I ₂ /I ₁ = 0, the maximum signal transmission efficiency η ₂ (max) is η _{2( nax)} =1/{1/h _e +2V _T /I ₁ 1/Z ₂
}...(12) becomes. Here, if we set 1/h _e <<2V _T /I ₁ 1/Z ₂ , η _2(nax) becomes η _2(nax) I ₁ /2V _T ·Z ₂ =C ₁ (13). Usually h _e = 50 to 300, so C ₁ is 0.1
It takes a value of ~1. Now let us temporarily set C ₁ = 0.1 and h
When _e varies between 50 and 300,
The variation in the maximum gain η _{2 (nax)} in equation (13) is about 1 dB, and when there is no C ₁ term (C ₁ = 0)
Compared to the variation in the maximum gain η _{2 (nax)} of about 16 dB, it can be seen that the variation in the maximum gain is significantly improved by the C ₁ term.

FIG. 4 shows an embodiment in which the impedance circuits 16 and 17 in the basic circuit of FIG. 3 are fixed resistors 16a and 17a having a resistance value R ₂ , and the other components are the same as in FIG. 3. In the embodiment shown in FIG. 4, the transmission efficiency η ₂ is the same as equation (10), η ₂ =1/{1/h _e +2V _T /I ₁ R ₂ +I ₂ /I
₁ (1/1-α _i )}... (14) The gain control at this time is performed by making the current value I ₁ of the constant current source 3 a constant current and controlling the current value I ₂ of the constant current source 15. The maximum transmission efficiency η _2(nax) at this time is η _2(nax) = 1/{1/h _e +2V _T /I ₁ R ₂ }...
…(15) becomes. In the above equation, V _T =k _T /q, which is a constant value of 26 mV at room temperature (T = 300〓). on the other hand
I ₁ R ₂ is the product of current and resistance, and in normal integrated circuit technology, the directions of variations in current and voltage cancel each other out, so an absolute accuracy of about 2 to 5% can be obtained. Therefore, 2V _T /I ₁ in equation (15)
Since the value of R ₁ =C ₁ can be set with relatively high precision, variations in the maximum transmission efficiency η _{2 (nax)} can be made small in the circuit of the embodiment shown in FIG.

FIG. 5 shows an embodiment in which the impedance circuits 16 and 17 of the basic circuit shown in FIG. 3 are constructed with diodes 16b and 17b and a constant current source 18 with a current value _I5 , and in this case, the impedance _Z2 becomes the diode operating resistance Z ₂ = 2V _T /I ₅ ... (16). Substituting this into equation (10), we get η ₂ =1/{1/h _e +I ₅ /I ₁ +I ₂ /I ₁ (
1/1−α _i )}... (17) is obtained. Here, if the ratio of I ₅ and I ₁ is selected so that 1/h _e ≪I ₅ /I ₁ = C ₁ (constant), the maximum transmission efficiency η _2
(nax) variations can be suppressed. Furthermore, since the term _C1 does not include a temperature component as in equation (15), the temperature fluctuation is smaller than in the embodiment shown in FIG. Gain control at this time is based on the current value I ₁ of constant current source 3 and constant current source 15.
This is done by changing the ratio of the current value I ₂ of I ₁ ,
Either or both of _I2 may be changed.
However, when changing _I1 , the current _I5 of constant current source 18
It is also necessary to take into account that I ₅ /I ₁ is constant by changing the following. This can be solved by using common base transistors with an emitter area ratio of I ₅ /I ₁ when configuring the constant current sources 3 and 18.

FIG. 6 shows the base current of the transistors 13 and 14 in the basic configuration of FIG.
In this example, if the current values of the current sources 19 and 20 are I ₃ , I ₄ and 1, and I ₃ = I ₄ , then the transmission efficiency η ₂ is calculated as η ₂ = 1 from equation (10). /{1/h _e +2V _T /I ₁ Z ₂ +2I ₃ /
I ₁ (1/1-α _i )}... (18) It is expressed as follows.

As described above, the present invention uses the CE-to-E saturation impedance of at least two pairs of transistors as a variable impedance circuit, and controls the gain by controlling the ratio to the input impedance of the differential amplifier stage. Since the control current for varying the variable impedance circuit is a fraction of that of the conventional example, it is possible to reduce the variation in the base bias of the differential amplifier stage, which increases the circuit dynamics range and improves the frequency characteristics. Great improvement in terms of stabilization. Further, the present invention has significant effects such as less variation in maximum gain.

In addition, in the embodiment, the transistors 13 and 14 are
Although NPN transistors are used, it goes without saying that the objects and effects of the present invention can also be achieved with PNP transistors.

[Brief explanation of the drawing]

Fig. 1 is a block diagram of a conventional gain control device, Fig. 2 is an equivalent circuit diagram of a current source, Fig. 3 is a basic block diagram of the present invention, and Figs. 4, 5, and 6 are respectively in accordance with the present invention. FIG. 1, 2...transistor, 3...constant current source, 7
... Signal current source, 13, 14 ... Transistor,
15... constant current source, 16, 17... impedance circuit, 16a, 17a... fixed resistance, 16b,
17b...Diode, 18... Constant current source, 1
9, 20... Current source.

Claims (1)

[Scope of Claims] 1. A differential amplifier circuit comprising at least a first and second transistor pair, each having a base electrode as a differential input terminal and an emitter electrode connected in common; At least a third,
a variable impedance circuit including a fourth transistor; and a second impedance circuit connected between the input terminals and having an impedance value proportional to the input impedance value of the differential amplifier circuit; A dynamic signal current source is connected, a common connection point of the emitters of the first and second transistors is connected to the first constant current source, and base electrodes of the third and fourth transistors constituting the variable impedance circuit are connected to these transistors. A second constant current source for controlling the base current of the differential amplifier is connected, and either one or both of the first and second constant current sources is controlled to change the gain of the differential amplifier. Control device. 2. The gain control device according to claim 1, wherein the base electrodes of the transistor pair constituting the variable impedance circuit are commonly connected, and the connection point is connected to a second constant current source. 3. The second constant current source is composed of third and fourth constant current sources, and the base electrodes of the third and fourth transistors forming the variable impedance circuit are connected to the third and fourth constant current sources, respectively. 2. The gain control device according to claim 1, wherein the gain is controlled by the current ratio of the third and fourth constant current sources and the first constant current source. 4 As the second impedance circuit, first and second impedance circuits each have an anode electrode connected to each of the input terminals and a cathode electrode connected in common.
2. The gain control device according to claim 1, comprising a diode, and a fifth constant current source having a current value at a fixed ratio with respect to the first constant current source is connected to the cathode electrode connection point. 5. The first constant current source is a fixed current source, and the second impedance circuit includes a fixed resistor having a resistance value at least larger than the emitter resistance of the first and second transistors of the differential amplifier circuit. The gain control device according to item 1.