JPS626425B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to a switching device with a high switching frequency.

In recent years, switching power supplies have been widely used as DC power supplies with low voltage and high current specifications for semiconductor devices, especially ICs, due to their small size, light weight, and high efficiency.

There are many types of switching power supplies, and the main ones are simple power converters (Johns circuit, Morgan circuit, etc.), high voltage generation circuits for television receivers, or inverter circuits (Royer circuit, McRabbettsford circuit, etc.). There is. However, since all of these devices directly switch the voltage of the input DC power supply, they can be used at low switching frequencies of around 10 kHz, but at around 100 kHz, leakage inductance of power transfer transformers and switching elements (transistors, etc.) Due to the stray capacitance of the thyristor, a large transient response is superimposed on the output, making it difficult to design a product that meets specifications. In particular, losses occur when the switching element is turned on or off, which causes a decrease in power transmission efficiency. Furthermore, since the above-mentioned transient response has an extremely high frequency component, it is emitted from the power supply as noise and has an adverse effect on other equipment.

This invention was made in view of the above points, and when a switching element becomes conductive, there is no voltage across the switching element, and when a switching element is disconnected, there is no current flowing through the switching element. In addition, by selecting the values of passive elements such as transformers and resonant capacitors, and the conduction time and switching period of switching elements, we have developed a high-frequency switching device that prevents unnecessary transient responses and improves power transfer efficiency. It provides:

The present invention will be explained in detail below with reference to Examples.

FIG. 1 is a circuit diagram showing an embodiment of the present invention. Reference numeral 1 denotes an input DC power source, and a switching element 3 is connected to both ends of the input DC power source via a primary inductance L of a power transfer transformer 2. For example, a transistor is used as the switching element 3, and a switching pulse is applied from a pulse generator 5 to its control terminal 3a (corresponding to the base in the case of a transistor) via a resistor 4.
A resonance capacitor 6 and a damper diode 7 are connected in parallel to both ends of the switching element 3. Here, the resonance capacitor 6 has a capacitance sufficiently larger than the stray capacitance of the switching element 3. The transformer 2 equivalently has a primary inductance L, a value L _e obtained by converting the sum of the secondary inductance and the leakage inductance of the transformer itself to the primary side (hereinafter referred to as additional inductance), and a transformation ratio of 1:-n. The voltage generated on the secondary side of the ideal transformer TR is converted into DC by the rectifier diode 8 and the smoothing capacitor 9, and is output to the output terminal 10.
is taken out as a DC output to the load 11.

Next, the operation will be explained with reference to the waveform diagram in FIG. Now, when a switching pulse as shown in FIG. 2a is applied from the pulse generator 5 to the control terminal 3a of the switching element 3, the switching element 3
becomes conductive, but since the inductance L is connected in series, t=t _S
A current i _C flows which increases linearly between ~T _p . During the next period from t= _tp to _tpo , the switching element 3 is forcibly turned off, so the current i _C suddenly drops to zero. At this time, as shown in Figure 2c, t=
The current i _L flowing through the primary inductance L of the transformer 2 between t _S and t _po flows into the resonant capacitor 6 due to inertia even when i _C becomes zero.
As a result, i _L is t=
It changes with a cosine curve between _tpo and _te . On the other hand, a current i _L of inductance L is applied across both ends of the resonance capacitor 6 (both ends of the switching element 3).
At the same time as begins to flow due to inertia, a voltage u _C begins to occur, and this voltage u _C changes in a sine curve as shown by the solid line in Figure 2d. This voltage u _C is t=
After reaching zero at _te , it tries to turn back further in the negative direction, but because of the presence of the damper diode 7, it remains fixed at zero. That is, the current i _L flowing through the inductance L at t=t _e still has a finite value, but its direction is negative, so from now on, as shown in Figure 2e, the damper current flows through the damper diode 7. i _D , and therefore the current u _C is kept at zero between t=t _e and _{t p} when the damper current i _D is flowing. When the switching element 3 becomes conductive again by the next switching pulse, it returns to the initial state, and the same operation is repeated, and power is transmitted from the output terminal 10 to the load 11. During this time, the current i _S flowing through the rectifier diode 8
As shown in FIG. 2 f, the voltage u _C is the voltage of the input DC power supply 1 (input voltage) E _i and the converted value E _p of the output voltage E ₀ appearing at the output terminal 10 to the primary side of the transformer 2. ′ (=nE ₀ ), the time t when the sum E _i +E _p ′ is exceeded
The flow starts from =t _f , and after u _c falls below E _i +E _p ' again, the flow continues until t = t _p due to the inertia of the inductance L, and then ends.

Here, the waveform shown by the solid line in FIG. 2 corresponds to the conduction time T _po of the switching element 3 and the switching period T
This is a case where the inductances L, L _e , the capacitance C of the resonance capacitor 6, E _i , E _p ', etc. are appropriately selected. If these selections are incorrect _, the voltage u _C across the switching element 3 will respond as shown by the broken line in _FIG . is u _R
, and this residual voltage u _R is rapidly short-circuited by the conduction of the switching element 3, resulting in a sharp negative spike SB.

On the other hand, according to the present invention, the above-mentioned T _po ,
By appropriately selecting T, L, L _e , C, E _i , E _p ', etc., the time t=t _po when the switching element 3 is disconnected as shown by solid lines in FIG. 2 b and d can be determined. It is possible to prevent i _C and u _C from overlapping each other, and due to the existence of t _e such that t _e −t _S <T, even at the time point t=t _e when the switching element 3 becomes conductive, i _C and u _C do not overlap. Since they can be prevented from overlapping, it is possible to eliminate the occurrence of unnecessary transient responses such as the spike pulses described above.

Hereinafter, T _po , T, L, L _e , C, according to this invention,
The selection conditions for E _i , E _p ′, etc. will be explained. Third
The figure shows the relationship between L _e /L and E _p '/E _i so that i _C and u _C do not overlap when the switching element 3 is disconnected and conductive.
By selecting these within the range shown by hatching, the conditions are satisfied. That is, L _e /L
When E _p '/E _i is plotted on the vertical axis and E p '/E i is plotted on the horizontal axis, the point P at (E _p '/E _i , L _e /L) = (0.4, 0.8),
Point Q at (0.7, 0.8), Point A at (1.0, 0.7), (1.3,
Point R at (0.44), point S at (1.54, 0.1), (10.0,
Point T at (0.1), point U at (10.0, 10.0), (0.4,
10.0) The relationship between L _e /L and E _p '/E _i is selected so that it falls within the range surrounded by the closed polygonal line connecting point O in 10.0).

On the other hand, L and C are selected as follows.
In the configuration shown in Fig. 1, if the currents i L and i _Le flowing through the primary inductance _L and additional inductance L _e of the transformer 2, and the voltage u _C across the resonance capacitor 6 are taken as state variables, the following state equation is obtained. is obtained.

X〓AX＋B……(1) However Waveform analysis of this state equation is performed using the Rungekuttergil method, L _e /L is used as a parameter, and the square root of the ratio of L and C, that is, the characteristic impedance Z _p =
FIG. 4 shows an investigation of changes in the output power P _put using √ as a variable. In that case E _p ′/E _i =1.15
Furthermore, Z _p was varied from 20 to 70 Ω in four cases where L _e /L was 0.5, 0.6, 0.7, and 0.9. Here, the curve shown by the broken line when L _e /L is out of the hatched range in Figure 3, L _e /L = 0.5, is:
In FIG. 2, at the time t=t _p , u _C remains, which is a state in which no mode is applied. Also L _e /L
20≦Z _p ≦40 shown by the solid line in the curve when =0.6
The range of is the state where the mode rides, and is indicated by the dashed line.
The range of 40<Z _p is a state in which no mode is applied.
Furthermore, all the curves when L _e /L < 0.7 are in the mode, and the output power P _put is also 180W to 50W.
The range of degrees changes smoothly. From the above results, it was found that in the above example, the value of Z _p should be selected within the range of 20 to 80 Ω.

As is clear from FIG. 4, the selection range of the characteristic impedance Z _p is determined by E _p '/E _i , L _e /L, the magnitude of the required output power P _put , and the range in which the mode rides. Therefore, when selecting Z _p , first select so that each value of E _p '/E _i and L _e /L falls within the shaded area in Figure 3, and also select the required output power P _put . After determining the value, Z _p is selected so that the mode is on.

Next, the output power P _put is a function of T _po /T,
FIG. 5 shows this relationship with E _p '=100, 140, and 150 _V as parameters for E _i =130 V and Z _p =40 Ω. From now on, T _po /T
It can be seen that the change in output power P _put when changed from 0.15 to 0.45 increases almost linearly from 35 to 270W.

Although the circuit configuration itself shown in FIG. 1 is the same as the high voltage generating circuit of a television receiver, there are the following differences between this high voltage generating circuit and the present invention. The first point is that the selection of E _p ' is extremely different. In other words, in the high voltage generation circuit of a television receiver, E _p ' is E _i
In contrast, in the present invention, E _p ' is set to be approximately 1/2E _i to 2E _i . The second point is that the selection of L and L _e in FIG. 1 is significantly different. That is, in the case of a high voltage generation circuit, L _e is set to be several percent or less of L, but in this invention, as shown in FIG. 3, L _e /L
is a function of E _p ′/E _i , L _e is 10% or more of L,
For example, if we take the case of point A where E _p ′/E _i =1,
70% or more. The third point is that in the case of a high voltage generation circuit, the stray capacitance of the secondary winding connected in parallel to the output terminal of the transformer, in conjunction with the additional inductance L _e , has a significant negative effect on the waveform of the flyback pulse and the ringing waveform during the scanning period. On the other hand, in this invention, the switching frequency of the switching element is high and the number of turns of the transformer secondary winding is small, so this stray capacitance is very small at about 1 to 5 PF, so the existence of this stray capacitance is The point is that it is not a problem at all.

As described in detail above, according to the present invention, the switching element can be used in a state close to an ideal switch without switching loss, thereby improving power transfer efficiency. This effect is clear from FIG. In other words, Figure 6a shows the ASO (Area Safety) when the DC voltage is simply chopped.
The curve is rectangular as a whole because voltage and current remain during the transition period when the switching element becomes conductive or disconnected. In this case, the area of the rectangle becomes a loss. On the other hand, the ASO curve in this invention is as shown in FIG. 6b. In other words, when the switching element is disconnected, the voltage across it has become zero, while when it is conductive, some voltage remains, but the trajectory length until it reaches zero is much shorter than in Figure 6a. I can see that Therefore, the loss in the switching element becomes extremely small.

Furthermore, according to the present invention, as described above, when the switching element becomes conductive, there is almost no voltage across it, and when it becomes disconnected, no current flows, so that the switching element shown in FIG. As is clear from the voltage and current waveforms at , no unnecessary transient response occurs. This is very effective as a noise countermeasure.

Note that the present invention is not limited to the configuration shown in FIG. 1; for example, the same applies to a configuration in which a tap is equivalently provided in the middle of the inductance L _e and one end of the inductance is connected to the tap. It is possible to implement

[Brief explanation of the drawing]

Fig. 1 is a circuit configuration diagram showing one embodiment of this invention, Fig. 2 is a waveform diagram of each part to explain its operation, and Figs. 3 to 5 explain selection conditions for numerical values of each part in this invention. FIGS. 6a and 6b are graphs showing the ASO curve of the switching element, and FIG. 7 is a diagram showing the voltage and current waveforms of the switching element in this invention. 1... Input DC power supply, 2... Power transmission transformer, 3
...Switching element, 6...Resonance capacitor, 7
...damper diode, 8...rectifier diode, 9
...smoothing capacitor.

Claims (1)

[Claims] 1. A switching element is connected to the input DC power supply via the primary winding of the transformer, and a resonance capacitor is connected so as to resonate in parallel with the primary winding of the transformer, so that the switching element is connected to the input DC power supply through the primary winding of the transformer. In a device that rectifies and smoothes voltage to obtain DC output, the value converted to the transformer primary side of the sum of the inductance on the secondary side of the transformer and the leakage inductance of the transformer itself.
The ratio of Le to the inductance L on the primary side of the transformer
E _p '/E when Le/L is plotted on the vertical axis and the ratio E _p '/E _i between the converted value E _p ' of the output voltage to the transformer primary side and the input voltage E _i is plotted on the horizontal axis. _i , Le/L) is (0.4, 0.8) at point P, (0.7, 0.8) at point Q, (1.0,
Point A at (0.7), Point R at (1.3, 0.44), (1.54, 0.1)
Le/L and E so that they fall within the range surrounded by the closed line connecting point S of , point T of (10.0, 0.1), point U of (10.0, 10.0), and point O of (0.4, 10.0). A high frequency switching device characterized in that the relationship between _p '/E _i is selected.