JPS6341249B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION (1) Technical Field of the Invention The present invention relates to a three-tap transversal type amplitude equalizer in the IF (intermediate frequency) band that equalizes amplitude distortion due to microwave fading.

(2) Prior art and problems Conventionally, space diversity and frequency diversity have been used to equalize amplitude distortion due to fading of microwave signals, but they have not been used to equalize the amplitude of IF signals converted from microwave signals. An IF band amplitude equalizer is used.

Figure 1 shows a one-dimensional three-dimensional diagram, which is a typical example of an amplitude equalizer.
1 is a block diagram showing a tap transversal type cosine equalizer (hereinafter simply referred to as a transversal equalizer); FIG. In the figure, 1 is an input terminal;
2, 4, 5, 6, and 8 are hybrids, 3 and 7 are delay lines with a delay time τ, 9 is an output terminal, and R is a terminating resistor.

In Figure 1, the IF signal IN of the microwave signal
is input from input terminal 1. The output signal SI from the hybrid (hereinafter also referred to as HYB) 5 in the center of the three taps is used as the main signal, and its amplitude is set to 1. Ignoring transmission losses due to cables and the like inside the transversal equalizer, the amplitudes of the output signals S2 and S3 of the left and right hybrids 2 and 8 of the hybrid 5 are equal, where S1/S2=S3/S1=γ.
Signal S2 is a leading wave of τ time with respect to S1, signal S3
becomes a delayed wave of τ time with respect to S1. Main signal S
1 is expressed as e ^j 〓 ^t , the leading wave S2 is expressed as γe ^j 〓 ^(t+ 〓 ⁾ , and the delayed wave S3 is expressed as γe ^j 〓 ^(t- 〓 ⁾ .

In the above circuit configuration, the output signal S of HYB2
2=γe ^j 〓 ^(t+ 〓 ⁾ and HYB8 output signal S3=γe ^j 〓 ^(t- 〓 ⁾
are combined in HYB4, and the combined signal is combined with the main wave e ^j 〓 ^t in HYB6, and the transfer function T(ω) at the output terminal 9 in this case is T(ω) = e ^j 〓 ^t + μe ^j 〓 ^(t+ 〓 ⁾ ＋γe ^j 〓 ^(t- 〓 ⁾ =e ^j 〓 ^t
(1+γe ^j 〓〓〓〓+γe ^-j 〓〓) = e ^j 〓 ^t (1+2γcosωτ) ...(1) where the inside of parentheses shows the amplitude characteristic of the transversal equalizer expressed as A(ω)=1+2γcosωt .

Since the above amplitude characteristic A(ω) does not include the j term which is the phase peak, the above transversal equalizer does not include delay distortion.

The above-mentioned equalizer equalizes inherent amplitude distortion, and has the drawback that it cannot equalize amplitude distortion that changes with changes in fading.

In order to eliminate the above drawbacks, the applicant has made an idea. The idea behind this is as follows. In other words, in a 3-tap transversal equalizer that equalizes the primary amplitude distortion and secondary amplitude distortion of a microwave IF signal that has undergone amplitude distortion due to fading,
comprising a first variable phase shifter and a second variable phase shifter controlled by a control voltage that equalizes the primary amplitude distortion, and controlled by a control voltage that equalizes the secondary amplitude distortion. The composition circuit includes an O-π phase shifter and combines the outputs of the first variable phase shifter and the second variable phase shifter. is combined with the main signal from the central tap of the 3-tap transversal equalizer to form an IF signal output, and the IF signal output is divided into predetermined frequency bands by a plurality of bandpass waveformers and each detected. the first and second variable The phase shifter is composed of a 90° hybrid, a sub-variable phase shifter, and a hybrid, respectively.
The sub variable phase shifter in each of the first and second variable phase shifters is controlled by the control voltage that equalizes the primary amplitude distortion, and the sub variable phase shifter in each of the first and second variable phase shifters is controlled by the control voltage that equalizes the secondary amplitude distortion. controlling a 0-π phaser,
The first variable phase shifter and the second variable phase shifter are controlled by the 0-π phase shifter.
By controlling the amplitude of the composite signal with the variable phase shifter,
The idea is to equalize the amplitude distortion of the IF signal (the details of this idea will be explained later in section (6)).

The problem with the above idea is that the first and second
The problem lies in how to realize the configuration of the variable phase shifter, especially the configuration of each sub-variable phase shifter therein. This is because the configuration should be as practical, inexpensive, easy to control, and simple.

(4) Object of the Invention In view of the above-mentioned problems, an object of the present invention is to provide an amplitude equalizer that is more practical, inexpensive, easy to control, and has a simple configuration.

(5) Structure of the Invention In order to achieve the above object, the present invention is characterized in that the sub variable phase shifter is configured by a variable attenuator.

(6) Embodiments of the invention Before explaining embodiments of the present invention, the above-mentioned idea,
That is, the configuration that is the premise of the present invention will be explained in detail.

FIG. 2 is a block diagram showing the basic configuration forming the premise of the present invention. In this figure, the same components as in FIG. 1 are designated with the same reference numbers and symbols. In Fig. 2, an IF signal IN that has undergone amplitude distortion due to fading is input to input terminal 1, and this signal IN is input to HYB2 as in Fig. 1, and delay line 3 → HYB5 → delay line HYB after 7
8, HYB5 outputs e ^j 〓 ^t as the main signal, HYB2 outputs the advanced wave e ^j 〓 ^(t+ 〓 ⁾ ,
A delayed wave e ^j 〓 ^(t- 〓 ⁾ is output from HYB8.

These advanced waves e ^j 〓 ^(t+ 〓 ⁾ and delayed waves e ^j 〓 ^(t- 〓 ⁾ are input to the first and second variable phase shifters 10 and 11, respectively, whose phases change by 360 degrees depending on the control voltage, and are inputted to a predetermined phase shifter 10 and 11, respectively, whose phases change by 360 degrees depending on the control voltage. set to phase. the first and second variable phase shifters 10,
The outputs of 11 are combined in HYB 12, and the combined signal is sent to variable attenuator 1 controlled by a control voltage.
3, the predetermined amplitude γ (amplitude ratio with the main signal) is set.

The output of the variable resistance attenuator 13 is input to the HYB14 and is combined with the main wave e ^j 〓 ^t output from the HYB5,
In this way, the IF signal IN having amplitude distortion is equalized to a predetermined waveform and output from the output terminal 9 (OUT).

In the above circuit system, if each of the leading wave and delayed wave signals is simultaneously rotated by the control voltage at the angles of +θ and -θ using the respective variable phase shifters 10 and 11, the transfer function T(ω) at this time is becomes as follows.

T(ω)=e ^j 〓 ^t +γ{e ^j( 〓 ^t- 〓〓 ^- 〓 ⁾ ＋e ^j( 〓 ^t+ 〓〓 ⁺
〓 ⁾ }=e ^j 〓 ^t [1+γ{ ^-j( 〓〓 ⁺ 〓 ⁾ ＋e ^j( 〓〓 ⁺ 〓 ⁾ ｝]=e ^j 〓 ^t { ¹⁺² 〓 ^cos( 〓〓 ⁺ 〓 ⁾ }……(
2), and its amplitude characteristic A(ω) is A(ω)=1+2γcos(ωτ+θ).

In the above, the first and second variable phase shifters 10, 1
1 equalizes the primary amplitude distortion, and the variable resistance attenuator 13
The secondary amplitude distortion is equalized. That is, the first and second variable phase shifters 10 and 11 set an arbitrary phase θ for equalizing amplitude distortion, and the variable resistance attenuator 13 sets a predetermined amplitude γ. From this, the first-order amplitude distortion generated in the band [ω1, ω2] by changing θ and γ,
Secondary amplitude distortion can be equalized. FIG. 3 is a block diagram for explaining the configurations of the first variable phase shifter 10 and the second variable phase shifter 11 shown in FIG. 2. Since the first and second variable phase shifters 10 and 11 are the same, the first variable phase shifter 10 will be described.

The output signal S2 from the HYB2 is input to the 90 degree HYB16 of the first variable phase shifter 10, and a signal S2-0 having the same phase as the S2 signal has a phase difference of 90 degrees from the S2 signal.
The signal S2-90 is output.

The S2-0 signal is input to the first balanced mixer 17, and the S2-90 signal is input to the second balanced mixer 18.

On the other hand, the control voltage is shaped into a sine wave control voltage by a sine wave shaping circuit 19 to control the second balanced mixer 18, and is also input to a cosine wave shaping circuit 20 to control the balanced mixer 17. The outputs of these first and second balanced mixers 17 and 18 are
By combining in HYB21, the phase can be made variable 360 degrees.

Here, if a predetermined value is given to the control voltage, the phase is set to a predetermined value.

The above first variable phase shifter 10 has a 90 degree HYB and a first variable phase shifter 10.
It consists of seven or more circuit elements such as a second balanced mixer, a sine wave shaping circuit, a cosine wave shaping circuit, and HYB. In this case, by inverting the polarity of the output of the 180 degree variable phase shifter using a 0-π phase shifter,
A phase shifter equivalent to a 360 degree variable phase shifter can be realized.

FIG. 4 is a block diagram showing an amplitude equalizer that has been simplified in view of the explanation of FIG. 3 and is a direct premise of the present invention. In this figure, the same components as in FIG. 2 are designated with the same reference numbers or symbols. In Figure 4, the IF signal with amplitude distortion due to fading is sent to HYB2 and HYB5.
2, S1, and S3 signals.

Signal S2 is 90°HYB2 of the first variable phase shifter 22.
3', and is branched into signal S2-90, which has a phase difference of 90 degrees with the signal S2 and the in-phase component S2-0. Signal S2
-90 is controlled to a predetermined amplitude by a sub variable phase shifter 24 controlled by a control voltage. As a result, the output of the sub variable phase shifter 24 and the S2-0 signal are input to the HYB 25, and by appropriately changing the control voltage, the phase of the signal output from the HYB 25 can be made variable.

On the other hand, 90°HYB23, sub variable phase shifter 26,
The second variable phase shifter 28, which is composed of HYB27 and has the same configuration as the first variable phase shifter 22, also controls the sub variable phase shifter 26 with the same control voltage.
The phase of the signal S3 can be rotated using the same principle as the variable phase shifter 22.

The respective outputs of the first and second variable phase shifters 22 and 28 are combined in the HYB 12, and the combined signal is sent to a 0-π phase shifter 29 controlled by the control voltage, with an amplitude corresponding to the control voltage. control and polarity are given, making the phase and amplitude γ of the composite signal variable. The signal whose phase is varied is the main signal S1.
The signals are synthesized in the HYB 14, and the synthesized IF signal output is output (OUT) from the output terminal 9.

With the above circuit configuration, the output signals from each HYB 2 and 5 are S1=e ^j 〓 ^t , S2=e ^j( 〓 ^t- 〓〓 ^- 〓 ⁾ , S3=e ^{j (} 〓 ^t- 〓〓 ^- 〓 ⁾ , and if γ is the amplitude (main wave and amplitude ratio) set by the O-π phase shifter 29, the transfer function T(ω) in this circuit system is T( ω)=e ^j 〓 ^t +γ{e ^j( 〓 ^t- 〓〓 ^- 〓 ⁾ ＋e ^j( 〓 ^t+ 〓〓 ⁺
〓 ⁾ } =e _j 〓 _t [1+γ{e ^-j( 〓〓 ⁺ 〓 ⁾ +e ^j( 〓〓 ⁺ 〓 ⁾ }], therefore, the amplitude characteristic A(ω) is A(ω) = 1+
2γcos(ωτ+θ). When the control phase in the 0-π phase shifter 29 is inverted from 0 to π depending on the value of the control voltage, the amplitude characteristic becomes A(ω)=1-2γcos
(ωτ+θ). From this, if we change θ corresponding to the value of the control voltage and and observe it in an arbitrary frequency band, we can obtain a control signal with a first-order slope that equalizes the first-order amplitude distortion, and a second-order slope that equalizes the second-order amplitude distortion. The slope control signal equalizes the primary and secondary amplitude distortions of the IF signal caused by fading.

Now, going into the main discussion, the subject of the present invention is how to realize the sub variable phase shifters 24 and 26 shown in FIG. 4, respectively.

FIG. 5A is a block diagram showing a first example of variable phase shifters 22, 28, and FIG. 5B is a vector diagram of the signals appearing in FIG. 5A. Note that since the variable phase shifters 22 and 28 have the same configuration, only the variable phase shifter 28 will be described. This first example is the most legitimate choice from the term "variable phase shifter," and the balanced mixer 31 realizes a sub variable phase shifter. Note that 90° HYB23 and HYB27 are already shown in FIG.
These 90゜HYB23, balance mixer 31 and
When the signals from the HYB 27 are represented by B and B, respectively, these signal vectors are as shown in FIG. 5B.

However, balanced mixers are generally expensive, and since they handle positive and negative voltages as control voltages, there has been a demand for a simpler configuration. Therefore, we decided to change the idea of a "variable phase shifter" to a variable attenuator. This is the gist of the invention. This is based on the fact that even with a variable attenuator, the phase can be made variable.

FIG. 6A is a block diagram showing a second example of variable phase shifters 22, 28, and FIG. 6B is a vector diagram of the signals appearing in FIG. 6A. Since the variable phase shifters 22 and 28 have the same configuration, the variable phase shifter 28 (the second example is shown as 28') will be described. Note that 90° HYB23 and HYB27 are already shown in FIG. These 90゜HYB
23, the balanced mixer 31, and the HYB 27 are respectively expressed as (0° phase shift signal), and these signal vectors are as shown in FIG. 6B. In other words, even if only the amplitude of the signal from 90° HYB23 (90° phase shifted signal) is made variable without changing the phase, the phase of the signal from HYB27 will ultimately be variable depending on the amplitude. Become.
This is based on the fact that the signal is generated by a vector combination of signals. With this focus,
The sub-variable phase shifter is composed of a variable attenuator 32.

By the way, what is clear from the vector diagram in FIG. 6B is that the phase θ can only change within the range of 0-π/2. This point is shown in Figure 5B.
This is a disadvantage since θ can vary in the range 0-π. However, from a practical point of view, if the delay line τ is appropriately selected, it is extremely unlikely that θ will be changed from 0 to π, and it is sufficient if it can be changed from 0 to π/2. As a result, the above-mentioned disadvantages are sufficiently offset by the advantages of the variable attenuator, which are inexpensive, simple in construction, and easy to control. Simple configuration means that it can be configured with a known π-type connection of PIN diodes, and easy control means that a voltage of both positive and negative polarities is not required as a control voltage.

FIG. 7 is a block diagram illustrating an amplitude equalizer according to the present invention, in which the variable attenuators of FIG. 6A are introduced as reference numerals 32 and 32', and the amplitude equalizer 40
to do.

FIG. 8 is a block diagram showing an example in which the amplitude equalizer of FIG. 7 is configured as an automatic amplitude equalizer. In this figure, the same components as in FIG. 7 are designated with the same reference numbers and symbols. 9A and 9B are waveform diagrams used to explain the operation of FIG. 8. In FIG. 8, when the IF signal output OUT outputted from the HYB 14 has an amplitude distortion waveform as shown by the solid curve in FIG. 9A, the frequency band Δf ₁ of the front stage of the IF signal output, It is divided into each frequency band using a band pass filter (BPF) suitable for each of the frequency band Δf ₂ and the frequency band Δf ₃ of the latter part. That is, the output of HYB14 passes through AGC-AMP51 and becomes △f ₁ at BPF52.
Then, the BPF 53 divides the signal into Δf ₂ and the BPF 34 divides it into Δf ₃ .

△f ₁ of the divided IF signal band is a detector (hereinafter referred to as
DET) is detected as a △f ₁ ' signal at 55,
Δf ₂ in the IF signal band is detected by the DET 56 as a Δf ₂ ' signal, and Δf ₃ in the IF signal band is detected by the DET 57 as a Δf ₃ ' signal.

The detected signals △f ₁ ′ and △f ₃ ′ are sent to the differential amplifier 5.
8 to form a difference signal component of Δf ₁ ′−Δf ₃ ′=e ₁ . The difference component signal e ¹ indicates the first-order amplitude distortion of the fading amplitude distortion, and corresponds to the control voltage of the first-order slope control signal at the time of amplitude equalization. The control voltage is applied to the first and second variable phase shifters 22 and 2.
8 to automatically equalize the primary amplitude distortion.

Also, the detected signals △f ₁ ′ and △f ₃ ′ are
It is input to the differential amplifier 59 via _R1 (△
An output signal of f ₁ ′+△f ₃ ′)×1/2−△f ₂ ′=e ₂ is formed. The signal e ₂ indicates secondary amplitude distortion due to fading, and corresponds to the control voltage of the secondary slope control signal at the time of amplitude equalization. The control voltage controls the 0-π phase shifter 29 to automatically equalize the secondary amplitude distortion. In this way, the amplitude distortion shown by the solid line in FIG. 9A is shaped into a predetermined equalized waveform (output out in FIG. 8) shown in FIG. 9B.

As described above, it is possible to detect the amplitude distortion of the IF signal spectrum and automatically equalize the primary and secondary amplitude distortions at the same time using the detected signal.

(7) Effects of the Invention As described above in detail, according to the present invention, it is possible to obtain a circuit that realizes the circuit configuration of FIG. 4 at low cost, with a simple configuration, and with easy control.

[Brief explanation of the drawing]

Figure 1 shows a one-dimensional three-dimensional diagram, which is a typical example of an amplitude equalizer.
FIG. 2 is a block diagram showing the basic configuration of the present invention, and FIG. 3 is a block diagram showing the tap transversal type cosine equalizer. FIG. 4 is a block diagram for explaining the configuration of the variable phase shifter 11, and FIG. 5A is a block diagram showing an amplitude equalizer that is simplified in view of the explanation in FIG. 3 and is a direct premise of the present invention. 5B is a vector diagram of the signal appearing in FIG. 5A, and FIG. 6A is a block diagram showing a second example of the variable phase shifters 22, 28. 6B is a vector diagram of the signal appearing in FIG. 6A, FIG. 7 is a block diagram showing the amplitude equalizer according to the present invention, and FIG. 8 is a vector diagram of the signal appearing in FIG. 6A.
A block diagram showing an example in which the amplitude equalizer shown in the figure is configured as an automatic amplitude equalizer, and FIGS. 9A and 9B are waveform diagrams used to explain the operation of FIG. 8. 2, 5, 12, 14, 25, 27...hybrid (HYB), 22...first variable phase shifter, 23,
23'...90°hybrid (90°HYB), 24,
26...Sub variable phase shifter, 28...Second variable phase shifter, 29...0-π phase shifter, 32, 32'...Variable attenuator, 40...Amplitude equalizer, IN...IF signal input , OUT...Automatic equalization IF signal output.

Claims (1)

[Claims] 1. Microwaves that have undergone amplitude distortion due to fading are heterodyne detected by a radio device and
Even in the signal, primary amplitude distortion and secondary amplitude distortion occur,
In order to equalize this, the main signal output from the central tap among the three taps, the leading wave signal which is advanced by a predetermined time with respect to the main signal and the main signal output from the remaining two taps, respectively. In a 3-tap transversal equalizer that generates a delayed wave signal delayed by a predetermined time and further synthesizes these three types of signals, the advanced wave signal is used as the first input, and the first-order amplitude distortion is equalized. a first variable phase shifter whose second input is a control voltage to be converted to A 0-π phase shifter controlled by an equalizing control voltage is provided, and a synthesis circuit is provided for synthesizing the outputs of the first variable phase shifter and the second variable phase shifter, and the output from the synthesis circuit is The synthesized signal passed through the 0-π phase shifter is combined with the main signal to become an IF signal output.
The frequency bands are divided into three types using band-pass filters in the middle and high ranges, each detected to obtain three types of detected signals, and the difference between the detected signals corresponding to the low and high ranges is calculated. generate the control voltage based on the difference between 1/2 of the sum of the detected signals corresponding to the low range and the high range, and the detected signal corresponding to the middle range. do,
Due to the fading at the control voltage and
The primary amplitude distortion and the secondary amplitude distortion of the IF signal are equalized, and the first and second variable phase shifters are used to generate a 90° hybrid signal and a 0° phase shifted signal from the 90° hybrid signal, respectively. It consists of a hybrid as an input, and a sub variable phase shifter which receives a 90° phase shift signal from the 90° hybrid and applies it to the hybrid under control by the control voltage, and An amplitude equalizer comprising a variable attenuator that makes the amplitude of the 90° phase-shifted signal variable.