JPS642033B2 - - Google Patents

Description

[Detailed description of the invention]

[Industrial Field of Application] The present invention provides an energy source that depends on a predetermined relationship between a value proportional to the square of the instantaneous voltage value of polyphase AC and a control amount at intervals sufficiently shorter than the cycle of polyphase AC input. Concerning AC-DC conversion to obtain output. [Prior Art] When it is desired to obtain a desired value of DC voltage from a multiphase AC input, for example, a commercial three-phase AC input, a conventional typical switching mode rectifier is shown in Fig. 1. There is. This conventional rectifier has an inductor connected in series to each three-phase AC input terminal.
L _u , L _v , L _w A three-phase full-wave rectifier circuit Rec that consists of six rectifiers D ₁ to _{D 6} and performs full-wave rectification of three-phase AC input.
1. Inductor L ₁ and capacitor C ₁ that constitute a smoothing circuit that smoothes the output of this three-phase full-wave rectifier circuit,
Switching transistors Q ₁ and Q ₁ ' are connected in series with each other via the primary winding N ₁ of the transformer T, and are conductive during the off period of these switching transistors to transfer the energy in the transformer to the capacitor C ₁
diodes D ₇ and D ₈ , a rectifier D ₉ that rectifies the voltage of the secondary winding N ₂ of the transformer T, a flywheel diode D ₁₀ , an inductor L ₂ and a capacitor C ₂ that form the output filter circuit, and the output terminal O ，
It consists of a control circuit C _po that controls the pulse width of switching transistors Q ₁ and Q ₁ ' so as to keep the DC output voltage between O' and output terminals O and O' constant. [Problems to be Solved by the Invention] In such a conventional rectifier, the three-phase AC input is rectified by the three-phase full-wave rectifier circuit Rec1, and then the inductor _L1
Since the DC current is smoothed by a smoothing circuit consisting of a capacitor and a capacitor _C1 , a pulsed current flows through each phase via the diode of the phase with the highest voltage, so it contains many harmonic components. . Even with the inductors L _u , L _v , and L _w provided in the rectifier circuit Rec1, at least low-order harmonic components cannot be removed. These harmonic components not only cause inductive disturbances in communication circuits and cause them to malfunction, but they also have an adverse effect on peripheral devices connected to the same power supply, and cause power loss in multiphase AC power supplies. Increasing the capacity of the generator and combining it with a generator results in an increase in capacity due to an increase in the power loss of the generator. Another disadvantage is that a smoothing inductor L ₁ and a capacitor C ₁ are required. [Means for Solving the Problems] The present invention eliminates such conventional drawbacks, and provides a symmetrical polyphase AC and two-phase three-wire system input side and its DC output side with a constant load. Based on the knowledge that the instantaneous value of the energy flow is constant, the multiphase AC input current is controlled in a specific manner by controlling the conduction period of each switching element so that the input current flows in a sinusoidal waveform. It is characterized by being able to reduce harmonic components of the current and obtain a constant DC output with a small ripple component. Symmetrical n-phase AC device with symmetrical loads (n≧3)
In this case, the instantaneous voltage V _j on the input side is expressed as V _j =√2V _n sin {ωt−2π(j−1)/n} (1). However, j=1, 2, 3, . . . n, V _n is the effective value of the input voltage V _j . Here, the load connected between each phase has a resistance value R
In the case of a resistive load with a resistive load, the instantaneous power P _i delivered to the input is P _i = _o 〓 ^j=1 V _j ² /R = 2V _n ² /R _o 〓 ^j=1 sin ² {ωt− 2π(j-1)/n} ...(2), and _o 〓 ^j=1 sin ² (ωt−2π(j-1)/n} is n/
2, so the instantaneous power P _i of the input section becomes P _i =n·V _n ² /R (3). This equation (3) shows that the energy flowing in the symmetrical n-phase AC device is constant. Therefore, if the load on the AC input is linear and the DC output side is controlled to obtain constant energy, a sinusoidal multi-phase AC input current will flow on the input side, so harmonics will be sufficiently suppressed. Components can be reduced. Furthermore, in order to extract a certain amount of energy to the DC output side, it can be seen from equation (2) above that it is sufficient to extract energy that is proportional to the square of the instantaneous voltage value of each phase at each point in time of the multiphase AC input. In order to obtain this instantaneous voltage value more precisely, it is desirable to compare the conversion frequency with the frequency of the polyphase AC input and select a value as high as possible. The present invention is based on the above-mentioned findings.
It provides AC-DC conversion. [Embodiments] Each embodiment of the AC-DC conversion according to the present invention will be described in detail below with reference to the drawings. First, one embodiment of the present invention will be explained with reference to FIGS. 2 to 5. The main circuit of the rectifier shown in Fig. 2 consists of inductors L _u , L _v , L w connected in series to the lines of each phase U, V, and W of the three-phase AC input, and capacitors C u and _{L w} connected between the phases. L _v , C _w single-phase full-wave rectifier D and capacitor
Rectifier circuits with the same configuration consisting of C ₁ ′ Rec1, Rec
2, Rec3, a pair of switching transistors Q ₁ and Q ₁ ' that perform switching operations at the same time, primary winding
A transformer T having a secondary winding N ₁ and a secondary winding N ₂ and switching transistors Q ₁ and Q ₁ ′ conduct when they are off, and the excitation energy stored in the transformer T is fed back to the capacitor C ₁ ′. Conversion sections G1, G2, G3 with the same configuration consisting of diodes D ₇ and D ₈ to
D ₉ , D ₉ ′, D ₉ ″, a flywheeling diode D ₁₀ , inductors L ₂ , L ₂ ′ forming a smoothing circuit, capacitors C ₂ , C ₂ ′, and capacitor C ₃ ,
A load F is connected to its output end. One of the structural features of the rectifier in this embodiment is that there is no need to provide a smoothing circuit for the input frequency between the rectifier circuit that rectifies each phase voltage and the conversion section, and this is a completely new control system that will be described later. By switching the switching transistors Q ₁ and Q ₁ ′ to open and close the rectified sinusoidal voltage, harmonic components of the input current can be significantly reduced, and the ripple component can be extremely low in stability. This is to obtain a DC output of 100%. Fig. 3 shows an example of the block configuration of a control circuit that performs this control method, Fig. 4 shows signals indicating the operation timing of each part, and Fig. 5 A and B
2A and 2B are diagrams for explaining input and output waveforms of the conversion section, respectively. In FIG. 3, a reference signal oscillator 1 generates a reference pulse signal at a frequency sufficiently higher than the frequency of the polyphase AC input, for example 20 KHz. This reference pulse signal is shown in FIG. 4 as signal a generated at time _t1 . At the rise of this signal a, the reset pulse forming circuit 2 generates a predetermined pulse width, for example, 1μ.
Generates a reset pulse with a pulse width of seconds. This reset pulse is used for reset via OR circuit 5.
It is applied to the gate of FET 6, turns on FET 6 by the width of the pulse, and discharges the charge in capacitor 7 to almost zero. The delay circuit 3 drives the on signal b at a time delayed by a time approximately equal to the carrier accumulation time of the switching transistors Q ₁ and Q ₁ ' from the fall of the reset pulse, that is, at a time t ₂ delayed from the signal a by a time τ. applied to the latch circuit 4. Accordingly, the drive latch circuit 4 applies a drive signal S ₁ to the bases of the switching transistors Q ₁ and Q ₁ ' of the first conversion section G1 to turn them on as shown by the signal d. A capacitor 7 connected in parallel to the FET 6 is charged by a constant current from a controllable constant current source 9 whose constant current value is controlled by an error signal from an error amplifier 8. The error signal output by the error amplifier 8 is a detection signal proportional to the DC output voltage in the rectifier.
Depends on the difference between S _d and the reference value. Therefore, after the charging voltage of the capacitor 7 drops to almost zero value by the reset pulse based on the reference signal a, the control amount, in this embodiment, the magnitude of the difference between the output voltage detection signal S _d and the reference value It increases with a proportionate rate of increase. The charge voltage of the capacitor 7 is determined by each comparator 10,
10', 10''. Full-wave rectifiers 11, 11', 1 are applied to the negative terminals of these comparators, respectively.
The DC side terminal 1'' of the full-wave rectifier 11 is connected to the resistors 12, 12', and 12'', respectively, and the AC side terminal 1 of the full-wave rectifier 11
A voltage proportional to the U-V inter-phase voltage of the three-phase AC input is applied between 3 and 14, and a voltage proportional to the V-W inter-phase voltage is applied between the AC side terminals 13' and 14' of the full-wave rectifier 11'. Also, the AC side terminal 13'' of the full wave rectifier 11'',
Since a voltage proportional to the voltage between the W and U phases is applied between 14" and 14", a sinusoidal voltage obtained by full-wave rectification of the alternating current voltage between the U and W phases appears at the negative terminal of the comparator 10. , Similarly, the negative terminals of the comparators 10' and 10'' are supplied with the AC voltage between the V and W phases, and the AC voltage between the V and W phases.
A sinusoidal voltage obtained by rectifying the alternating current voltage between the phases is applied. Respective comparators 10, 10', 1
0'' compares the aforementioned voltages applied to the positive and negative terminals, and outputs an off signal when the voltage at the positive terminal becomes equal to the voltage at the negative terminal.The off signal c of the comparator 10 is output at time t. ₃ is input to the drive latch circuit 4,
Accordingly, the drive latch circuit 4 stops supplying the base drive signal S ₁ to the switching transistors Q ₁ and Q ₁ ' in the conversion section G1. Therefore, transistors Q ₁ and Q ₁ ' are turned off at time t ₄ after the accumulation time has elapsed, as shown by signal d in FIG. Next, the output signal c from the comparator 10 is input to a reset pulse forming circuit 2' having the same configuration as the circuit 2. Accordingly, the circuit 2' applies a reset pulse similar to the reset pulse generated by the circuit 2 to the gate of the FET 6 via the OR circuit 5, turns it on, and discharges the charge in the capacitor 7. Further, a delay circuit 3' having the same configuration as circuit 3 receives the reset pulse and supplies an on signal e to the drive latch circuit 4' at time _t5 delayed by a time τ approximately equal to the carrier accumulation time from the signal c. Accordingly, circuit 4' sends a drive signal to the switching transistor of converter G2.
Apply S ₂ to turn it on (signal g).
The capacitor 7 is again charged by a constant current that depends on the magnitude of the output voltage detection signal S _d during the on-operation period of the converter G2. This charge voltage is comparator 1
0', the voltage proportional to the V-W phase voltage is compared with the full-wave rectified sinusoidal voltage as described above, and at the time _t6 when both voltages become equal, the signal f
is applied to the drive latch circuit 4' and also to the reset pulse forming circuit 2''. When the drive latch circuit 4' receives the signal f, it immediately stops sending out the base drive signal _S2 , and accordingly, the converter G2 changes its switch. The reset pulse forming circuit 2'' is turned off at time _t7 after the accumulation time of the switching transistor has elapsed.
0', a reset pulse is applied to the gate of FET 6 via OR circuit 5, and this is turned on to instantly discharge the charge in capacitor 7. The third delay circuit 3'' receives the reset pulse from the circuit 2'' and drives the on-signal h to the latch circuit 4'' at time _t8 delayed by the time τ from the signal f.
give to Accordingly, the circuit 4'' applies the drive signal _S3 to the switching transistor of the converter G3 to turn it on (signal j).The capacitor 7 is once discharged by turning on the FET 6, and the output voltage detection signal is output again. It is charged with a constant current proportional to S _d.The charging voltage of the capacitor 7 is compared with the sinusoidal voltage obtained by full-wave rectification of the voltage proportional to the W-U phase voltage of the three-phase AC input by the comparator 10''. , the comparator 10'' detects when these two voltages become equal.
At _t9 , an off signal i is applied to the drive latch circuit 4''.
Accordingly, the circuit 4'' stops supplying the drive signal _S3 ,
The converter G3 turns off when the storage time of its switching transistor has elapsed (signal j). In the section where all of the rectifiers 9, 9', and 9'' are non-conducting, the energy stored in the inductor _L2 is transferred to the diode _D10.
Flow through. Each part operates in this way, and one cycle T ends. In FIG. 3, reference numeral 15 indicates a compensation diode for canceling the adverse effect of forward drop in each full-wave rectifier 11, 11', 11''.To summarize the above operation, the switch in each conversion section The switching transistors operate in a time-division manner, that is, they perform switching operations sequentially within each cycle that is sufficiently smaller than one cycle of the multiphase AC input voltage, and moreover, as the switching transistor that is on is turned off, the next switching transistor is turned on. Furthermore, the switching transistor in each conversion section is controlled with a pulse width proportional to the product of the output voltage detection signal and a sinusoidal voltage proportional to each phase voltage, and each switching element is controlled by a pulse width proportional to the product of the output voltage detection signal and a sinusoidal voltage proportional to each phase voltage. The rectified sinusoidal voltage is opened and closed.Therefore, the switching transistors in each converter switch the energy taken out from each phase of the polyphase AC to the output side at a cycle sufficiently shorter than the cycle of the polyphase AC. It can be seen that switching control is performed so that it is proportional to the product of the control amount and a value proportional to the square of the instantaneous voltage value of the polyphase AC (Ksin ² θt: K is a constant).Furthermore, understand this control method. For the sake of simplicity, the three-phase AC input waveform of the converter in a 180° interval centered on the U phase will be explained with reference to Fig. 5. In Fig. 5A, V ₁ is the voltage waveform between UV and V phases, and V ₂ is In the V-W phase voltage waveform, V ₃ indicates the W-U phase voltage waveform, and in the same figure B, U ₁ , U ₂ , and U ₃ indicate the three-phase AC input waveform.
The output waveforms of the rectifiers D ₉ , D ₉ ′, and D ₉ ″ in each period starting from time t ₀ to _{t 5} corresponding to an interval of 30° are shown enlarged. Each converter G1 to G3 is
For example, each converter G operates at a conversion frequency of 20KHz.
1 to G3 are the corresponding phase-to-phase voltages of the three-phase AC input.
Opens and closes at a conversion frequency of 20KHz. As shown by the chain line in FIG. 5, each output voltage waveform in one cycle starting from time t ₀ when the UV phase-to-phase voltage V ₁ has a zero value is determined by the output voltage detection signal S as described in detail in the previous embodiment. The switching transistors of each conversion unit G1 to G3 are controlled with a pulse width determined by comparing the charging voltage of the capacitor 7, which depends on _d , with the corresponding instantaneous value of the voltage between each phase, and these switching Considering that the transistor opens and closes a sinusoidal voltage obtained by full-wave rectification of the corresponding voltage between each phase, the output U ₁ corresponding to the input V ₁ at time t ₀ has a very small pulse width and amplitude, and the input V ₂
Output U ₂ corresponding to and output U ₃ corresponding to input V ₃
It is easily understood that both are approximately equal, and both the pulse width and amplitude are sufficiently large compared to the output _U1 . Next, for the sake of illustration, there is a period starting at time t ₁ corresponding to 30° of the U phase of the phase AC input (of course, each conversion section is operating at a conversion frequency of 20 KHz even between times t ₀ and t ₁ ). To explain, the output U ₁ increases in both pulse width and amplitude as the input V ₁ rises in a sinusoidal waveform, and the output U ₂ becomes maximum in both pulse width and amplitude as the input V ₂ reaches its maximum amplitude. . Further, as the input V ₃ becomes approximately equal to V ₁ at time t ₁ , the output U ₃ becomes approximately equal to the output U ₁ in both pulse width and amplitude. Below, each cycle starting from time t ₂ , t _{3 ,} . . . t ₅ will be similarly explained. In other words, in this example, the frequency of the polyphase AC input is
Assuming 50 Hz, each period is divided into 400 equal parts with a conversion frequency of 20 KHz, and in each of these 400 equal parts, the conversion units G1 to G3 are time-divisionally controlled using the control method described above. The sum of the output voltages U1 to U3 in each of the 400 equally divided conversion cycles is controlled according to the error signal that is the difference between the output voltage detection signal V _d and the reference value, so this detection signal V _d is constant. If so, the output voltage U in each period is determined by the above equation (3).
It can be seen that the sums of 1 to U3 are all equal to each other. This situation can be seen in Figure 5B, where the output voltage U
If we consider that 1, U2, and U3 are arranged in a line over time and the conversion frequency is made high enough, we can see that the envelope will approach DC as much as possible. Therefore, from equation (3) above, since the instantaneous power P _i on the input side is constant, the instantaneous power obtained on the output side is constant, and the pulse width control described above can be performed at a conversion frequency that is sufficiently high relative to the input frequency. If so, a sinusoidal current will naturally flow on the input side simply by removing the high frequency component. In this embodiment as well, the sum of the outputs U ₁ to _{U 3} in each cycle due to the conversion frequency of 20 KHz, which is sufficiently higher than the frequency of the multiphase AC input, is almost constant, so it is possible to obtain a DC output voltage with small ripples, and the input side It can be easily understood that a sinusoidal current with extremely few harmonic components flows through the . Furthermore, according to this AC-DC conversion, as can be seen from the above, when the switching transistor of the converting section G1 is turned off, the switching transistor of the converting section G2 is turned on, and as the switching transistor of the converting section G2 is turned off, the switching transistor of the converting section G3 is turned on. Since it can be controlled so that it turns on, the output side can be connected in parallel using an on-on type converter, and it is easy to generate harmonics at low voltage and large current, and the maximum duty of the converter used in the example is It is possible to increase the utilization rate by up to 1/2. In order to enable this kind of follow-up control, in order to eliminate the influence of the carrier accumulation time of the switching transistor in each conversion section, the on-signal is delayed by approximately the same amount of time, and the pulse width is In order to improve the proportional accuracy with the input voltage, it is desirable to start charging the capacitor 7 before this delay time starts. Next, we will discuss specific examples of characteristics obtained by such AC-DC conversion. However, harmonics range from the 2nd to the 45th order. Input is 3-phase AC (50Hz) 200V, output is
For DC48V, 30A: Harmonics of input voltage - U - V: 1.67%, V -
W: 1.41%, W-U: 2.22% Harmonic component of input current-U: 5.02%, V: 4.07
%, W: 4.75% Power factor - 0.982, efficiency - 88.3%. The control method as described above can also be applied to the rectifier having the configuration shown in FIG.
The outputs of D ₉ , D ₉ ′, and D ₉ ″ are connected in series via diodes D ₁₁ , D ₁₁ ′, D ₁₁ ″, and the like. Next, FIG. 7 shows a case in which the converters G1 to G3 are configured as a bridge by switching elements, and setter-tap double-wave rectifier circuits are connected in series. The basic control method in this rectifier is exactly the same as the method in the previous embodiment, so it will not be described in detail. Next, in the rectifier shown in Fig. 8, the rectifier circuit on the output side is a full-wave rectifier bridge D ₉ , D ₉ ′, D ₉ ″, and one rectifier arm d ₁ , d ₁ ′, d ₁ ″ is connected in parallel. Connect the first inductor L in series with ₂ and the other rectifier arm
d ₂ , d ₂ ′, and d ₂ ″ are connected in parallel and connected in series with another second inductor L ₂ ″. The basic control method for this device is also the same as in the first embodiment, but the maximum duty of the voltage waveform that each converter supplies to one inductor L ₂ or L ₂ ' is limited to 1. Positive,
The feature is that the current is taken out from separate inductors L ₂ and L ₂ ' so that it can be divided into negative cycles and used up to 1/2 of the maximum duty of the output voltage waveform of each conversion section. It is possible to improve the rate. Next, Fig. 9 shows a basic on-on type three-phase rectifier in which six bidirectional switching elements S1 to S6 are connected in a three-phase full-wave rectification configuration, and the smoothing inductor _L2 is connected to this. The inductance is large enough to keep the flowing current constant during the conversion period. The basic control technology is the same as the control method described in relation to Figure 2, so it will not be described in detail, but the table below shows
An example of a sequence of switching operations of switching elements S1 to S6 during a period from 0° to 2π of the V-phase voltage is shown.

〔Effect of the invention〕

As described above, according to the present invention, the switching element is switched at a conversion frequency considerably higher than the frequency of the multiphase AC, and the energy taken out from each phase to the output side for each period according to the conversion frequency is the square of the multiphase AC. Since it is controlled so that the total energy extracted from each phase is proportional to the product of the control amount and the control amount, it is possible to flow an input current with very small harmonic components. Inductive disturbances due to harmonic components in circuits can be prevented, and a constant output voltage with small high frequency ripples can be obtained. Further, according to the present invention, there is no need to provide a smoothing circuit between the rectifier circuit on the input side and the converter, so the device can be made smaller. It goes without saying that a thyristor or the like can be used in addition to a transistor as a switching element, and although the above embodiment describes a symmetrical three-phase AC input, the same applies to other polyphase inputs or two-phase three-wire systems. Even if the input waveform is slightly deformed, ripples at the output can be reduced by controlling the energy flow at the input to be constant in the feedback loop.

[Brief explanation of drawings]

Fig. 1 is a diagram for explaining a conventional method of controlling a rectifier, Fig. 2 is a diagram showing an example of a rectifier for carrying out an embodiment of the present invention, and Fig. 3 is a block diagram of the control circuit. Figure 4 is a diagram showing the configuration, Figure 4 is a diagram showing signals indicating the operation timing of each part, and Figure 5 is a diagram showing the operation timing of each part.
The figure is a diagram to explain the input and output side waveforms.
FIGS. 8 to 8 and 9 to 11 are diagrams showing different examples of rectifiers for implementing the rectification control method according to the present invention, respectively. Rec1~Rec3... Rectifier circuit, G1~G3...
Conversion section, F...Load, C _po ...Control circuit, 1...
Reference signal generator, 2, 2', 2''...Reset pulse forming circuit, 3, 3', 3''...Delay circuit, 4,
4', 4''...drive latch circuit, 5...OR circuit,
8...Error amplifier, 9...Controllable constant current source, 1
0, 10', 10''... Comparator.

Claims (1)

[Scope of Claims] 1. A method for controlling a rectifier which receives multiphase alternating current as input and obtains an output voltage by sending energy to an output through the switching element when the switching element is turned on, wherein the switching element of each phase is At a conversion frequency higher than the frequency of the multiphase alternating current, and in each period determined by this conversion frequency, the switching elements of each phase are turned on as the switching elements of the other phases are turned on as the switching elements of the phases are turned off. control so that the switching operation is performed in sequence, and the power that can be extracted from each phase of the multiphase AC to the output side in each cycle determined by the conversion frequency is proportional to the square of the instantaneous voltage of the multiphase AC. 1. A method for controlling a rectifier, characterized in that control is performed in a manner dependent on a control amount according to a detection signal and a control amount according to a detection signal. 2. The method of controlling a rectifier according to claim 1, wherein the switching elements of each phase perform switching operations in a predetermined sequence. 3. Control of the rectifier according to claim 1, wherein the switching elements of each phase perform switching operations in the order of the phases in which the detected voltage value in each phase of the multiphase AC input is larger or smaller. Method.