JPH0257874B2 - - Google Patents

Description

[Detailed Description of the Invention] [Technical Field of the Invention] This invention provides a moving target detection function (Moving target detection function).
This relates to a pulse radar device equipped with Target Indication (hereinafter referred to as MTI).
The purpose is to dramatically improve MTI performance.

[Prior art]

Conventionally, there has been a device of this type as shown in FIG. This is an example of a coherent MTI radar, and in the figure, 1 is the radar antenna, 2 is the transmitter/receiver switch, 3 is the transmitting klystron amplifier (a TWT amplifier may be used instead), and 4 is the klystron power amplifier 3. 5 is a mixer that converts the high frequency signal received by the radar into an intermediate frequency; 6 is a stable local oscillator; 7 is a pulse modulator for driving the radar;
8 is a mixer for converting an oscillation signal from a coherent oscillator 10 to a transmission high frequency signal, which will be described later.
is an intermediate frequency amplifier, and 9 is a phase detector (Phase) for detecting the Doppler frequency shift signal.
10 is a coherent oscillator used as the radar's original oscillator; 11 is a trigger oscillator that down-regulates the output of the coherent oscillator 10 to create a trigger signal at the radar's transmission repetition frequency. It is.

In addition, 12 is an ultrasonic oscillator (Carrier Wave) that oscillates ultrasonic waves so that it can be input to the MTI delay line 14, which is a delay element that uses ultrasonic waves.
13 is a modulator for modulating the output signal of the phase detector 9 with the ultrasonic wave output from the ultrasonic oscillator 12. 14 is a delay line for MTI, which is a circuit that delays the input signal by one period of the radar repetition period and outputs the delayed signal. 15 is an amplifier for recovering the amplitude attenuation in the delay line 14; 16
is a detector, and the radar signal is extracted as a video signal at its output. 17
is an attenuator for providing attenuation corresponding to the delay line 14, and 18 and 19 are similar to the amplifier 15 and detector 16, respectively. 2
0 is a subtractor that subtracts a delayed signal and a non-delayed signal to extract a difference.

Next, the operation will be explained.

If the transmission frequency of the radar is t, and when the transmitted wave hits a moving target and is received as a reflected wave again, it has undergone a Doppler frequency shift d, then the frequency of the reflected wave is expressed as t±d. The Doppler frequency shift d is expressed by the following equation.

d=2Vr/λ=2Vr・t/c Vr: Velocity of the moving target λ: Transmission wavelength c: Speed of light This Doppler frequency deviation d is extracted as the output of the phase detector 9 in FIG. The principle is as follows.

That is, let the frequency of the reference signal from the coherent oscillator 10 be c, and the frequency of the received signal from the intermediate frequency amplifier 8 be c±d, and for simplicity, express each as continuous sinusoidal signals Vcoho and Vi, then Vcoho=Asin2πct Vi=Bsin(2πct±φd) φd=2πdt A, B are amplitudes t is time Therefore, the output Vpsd of the phase detector 9 is, Vpsd=Vcoho・Vi={Asin2πct}{Bsin(2πc
t±φd)}=AB/2{cosφd−cos(4πct+φd)}. In the above equation, the component including the intermediate frequency c in the second term is removed by a filter after the detection circuit, and only the component (AB/2)·cosφd in the first term is extracted from the phase detector 9. Here, φd=
Since 2πdt, (AB/2)·cosφd is a video signal whose amplitude changes depending on the value of the Doppler frequency shift d.

When this video signal is fed into a normal MTI made up of the components 12 to 20 described above, only the changes in amplitude due to the value of the Doppler frequency deviation d are extracted. Therefore, only the moving target signal is output, and the fixed target signal is eliminated.

FIG. 2 shows an example of frequency response characteristics in a single cancellation method using one delay line in a normal MTI. The maximum value of the output amplitude is normalized to 1. In the figure, T represents the pulse transmission repetition period of the radar, and the delay time of the delay line 14 is also T. It is a well-known fact that in this type of MTI, there is a Doppler frequency at which the moving target signal corresponding to the blind speed is also canceled every integer multiple of 1/T.

Conventional pulse radar equipment is configured as described above, and if the reflected signal of a fixed target is an ideal one with no Doppler frequency shift, the reflected signal of the fixed target will be completely erased and it will move. Only the reflected signal M of the target is detected. However, in reality, reflected signals from fixed targets such as land and mountains are
As shown by diagonal lines in FIG. 3, the spectrum is spread out on the frequency axis. Such a clutter spectrum S could not be sufficiently eliminated using the MTI response characteristics as shown in FIG.

As a conventional method to solve this problem, a multiple cancellation system is constructed using multiple delay lines, and negative feedback technology is also adopted to form a system as shown in Figure 4.
There is one that uses MTI response characteristics to improve the fixed target cancellation ratio. However, even with this method, the effect was small considering the complexity of the MTI configuration.

[Summary of the invention]

The present invention was made in order to eliminate the drawbacks of the conventional ones as described above, and it applies frequency modulation to the transmitted wave of the radar, extracts the demodulated wave of the modulated signal from the received wave reflected by the transmitted wave, and The purpose of the present invention is to provide a pulse radar device that can dramatically improve MTI characteristics compared to conventional ones by using it as an MTI input.

[Embodiments of the invention]

An embodiment of the present invention will be described below with reference to the drawings. In FIG. 5, the same reference numerals as in FIG. 1 indicate the same parts. 22 is a modulated wave oscillator for FM, 23
is a frequency modulator, 40 is a pulse type FM demodulator, 5
0 comprehensively shows the configuration (moving target detection means) having the MTI function of the component parts 12 to 20 in FIG.

Next, the operation will be explained.

In the device of this embodiment, a frequency modulator 23 is newly provided between the coherent oscillator 10 and the mixer 7, and the frequency of the coherent signal for transmission is modulated by the frequency of the modulated wave oscillator 22 (hereinafter referred to as FM).
). Then, the frequency-modulated output of the coherent oscillator 10 passes through the mixer 7 and reaches the klystron 3, where it becomes a transmission high-frequency pulse and is transmitted from the antenna 1 into space. The carrier frequency of the transmitted high-frequency pulse is c,
When the modulation frequency is m and the maximum frequency deviation is Δ, the modulation index m is given by m=Δ/m. Here, the modulation index m is 0.3 to 0.4
If the relationship between the modulation frequency m and the maximum frequency deviation Δ is determined so that it has a low value, the second
The sideband waves and above are considerably smaller, and it is necessary to consider only the first sideband wave. Therefore, the frequency spectrum becomes as shown in FIG.

Figure 6 conceptually shows the spread of the spectrum of each component due to pulse modulation in addition to FM.
This spread is determined by the pulse modulation waveform. The voltage e(t) of this transmitted high-frequency pulse
Expressed numerically, it becomes as follows.

e(t)=Eo {1/2 _∞ 〓 〓 ⁿ⁼¹ 2/nπ[sinnπ/K]cosn2π/T}・cos(ωot+
msinPt)...(1) where Eo: voltage amplitude K: T/r T: transmission period r: transmission pulse width ωo=2πc, c is carrier frequency m=Δω/P=Δ/m Δω=2πΔ, Δ is maximum frequency Deviation P=2πm, m is modulation frequency In the above equation, the term in square brackets is a term representing the waveform of pulse modulation, and the cos term is a term representing frequency modulation.

The voltage e'(t) when this pulse FM modulated wave undergoes a Doppler frequency shift is expressed as follows.

e′(t)=aEo{1/2 _∞ 〓 〓 ⁿ⁼¹ 2/nπ[sinnπ/K′]cosn2π/Τt}・cos{(
ωo±ωd)t+Δω/Pdsin(P±Pd)t}...(2) However, a: Value determined by the reflection area of the target ωd=(4π/λo)・Vr, λo is the wavelength of the carrier wave,
Vr is the axial velocity Pd = (4π/λp)・Vr, λp is the wavelength of the modulated wave,
Vr is the same as above K'=T/(1±ωd/ωo)τ If we conceptually represent the spectrum of the pulsed FM wave according to equation (2) above, as shown by the dotted line in Fig. 6, Doppler deviation d from the spectrum of
The shape will be shifted. This figure shows a case in which the frequency is shifted toward the higher side. The magnitude of this shift differs slightly by the difference in frequency between the carrier wave and the upper and lower sideband waves, and this is shown as d' and d'' in the figure.

These pulsed FM waves are then received by radar. The received wave is transmitted from the antenna 1 to the transmission/reception switching section 2
The signal is then converted to an intermediate frequency by a mixer 5, amplified by an intermediate frequency amplifier 8, and input to a pulse-type FM demodulator 40. This demodulator 40 is constructed in accordance with the spectra shown in FIGS. 6 and 7, and is shown in FIG.

In FIG. 8, 40a is an input terminal, and 42 is an input terminal.
IF signal splitter, 43 is a bandpass filter for upper sideband, 44 is a bandpass filter for carrier wave, 4
5 is a band pass filter for the lower side band, and 46 is passed through each band pass filter 43, 44, 45.
An IF signal adder for adding IF signals, 47 an amplitude limiter for making the amplitude constant before the received wave is demodulated by a frequency discriminator 48, and 49 a demodulator for amplifying only the frequency band of the demodulated signal. wave amplifier, 4
0b is an output terminal.

In the pulse-type FM demodulator 40, it is necessary to improve the signal-to-noise ratio by using a bandpass filter that passes only the frequency component of the pulse before performing frequency discrimination. As shown in the figure,
Each band filter 43 to 45 may extract only each component. Here, the modulation index is ~.
In a low range of about 3 to 0.4, other sideband components are sufficiently small, so even if they are ignored, waveform distortion is small and the purpose of this embodiment is not hindered.

The demodulator 40 having such a configuration generates pulses.
It will be explained below that when the FM wave is demodulated, the modulated wave and its Doppler frequency component are extracted.

What is shown in FIG. 9 is the input/output characteristic Fd of the frequency discriminator 48. The horizontal axis shows the input angular frequency,
The output voltage V is shown on the vertical axis. For simplicity, below:
The input and output waves are explained as continuous waves (cw).

The instantaneous angular frequency shift of the FM wave subjected to Doppler shift is from equation (2), phase angle φ = (ωo±ωd)t +Δω/P±Pdsin (P±Pd)t, so the instantaneous angular frequency is , dφ/dt=ω =ωo±ωd+Δωcos(p±pd)t (3).

Here, if the characteristic Fd with respect to the change in of the input signal angular frequency of the frequency discriminator 48 is a straight line within the maximum angular frequency deviation width ±Δω as shown in FIG. 9, then the characteristic is V=k( ω−ωo). Therefore, the voltage V of the output wave out with respect to the input wave in expressed by equation (3) is V=k(ω-ωo) =k{ωo±ωd+Δωcos(P±Pd)t-ωo)} =±kωd+kΔωcos( P±Pd)t However, k is a constant Here, kΔω is a constant value, so if this is set as Vc, V=±kωd+Vccos(P±Pd)t.

Here, if the modulation angular frequency P is taken as P≫ωd with respect to the Doppler angular frequency deviation ωd of the carrier wave, kωd in the first term of the above equation becomes DC-like. This frequency discriminator 4
The output voltage waveform of No. 8 is shown in FIG. 10, and the kωd component is easily removed by the demodulated wave amplifier 49 at the subsequent stage. Therefore, the output terminal 40b has
The component of Vccos(P±Pd)t will be extracted. This means that the original modulation angular frequency P and its Doppler angular frequency deviation component Pd have been extracted. Note that it is very easy to satisfy the above condition P≫ωd in this embodiment.

Next, the effects of this embodiment will be explained from the definitions of ωd and Pd used in equation (2) above.

Here, ωd=4π/λoVr, Pd=4π/λpVr, and λo=c/c, λp=c/m (c is the speed of light), so Pd/ωd=λo/λp=m/c becomes. This shows that the Doppler frequency shift is directly proportional to the carrier frequency c and the modulation frequency m.

The demodulated wave extracted by the pulse demodulator 40 and subjected to Doppler shift enters the MTI 50 via the phase detector 9. Here, the phase detector 9
In this case, the output signal of the modulation oscillator 22 is used as the reference signal instead of the output signal of the coherent oscillator 10. Further, the MTI 50 is the same as the conventional one except that it operates in the demodulated wave signal band. Note that if the frequency of the demodulated wave is the frequency of the ultrasonic wave required by the delay line 14, the ultrasonic oscillator 12 and modulator 13 in FIG. 1 are unnecessary.

Then, the value of the Doppler frequency shift of the demodulated wave entering MTI50 is determined by the modulation frequency as described above.
m, and since this modulation frequency m is considerably lower than the carrier frequency c,
The Clutter spectrum S in Figure 3 is
It has a compressed form with a ratio of m/c, and as shown in FIG. 11, it becomes almost a line spectrum.
On the other hand, the Doppler frequency deviation of the moving target signal is also compressed by the same ratio, but in the case of a moving target, since it is moving, it is inherently much smaller than the clutter, as shown in Figure 2. has a large Doppler frequency shift. Therefore, assuming the maximum velocity Vrmax of the target to be captured by the radar in the radar axis direction, the modulation frequency m should be adjusted to such an extent that the Doppler frequency deviation of the target signal at this time does not become smaller than 1/T in Fig. 11. By selecting , the clutter spectrum can be compressed by the m/c ratio without impairing the detection characteristics of the moving target signal.

Therefore, the fixed target erasure ratio of MTI50 can be improved by the m/c ratio. At this time, the detection range R of the signal corresponding to the speed of the moving target is only between 0 and 1/T, as shown in FIG.

Here, the modulation frequency m has the following relationship with respect to the maximum axial velocity Vrmax of the moving target.

That is, if md is the Doppler frequency shift that modulation frequency m undergoes, then 2πmd=Pd=4π/λpVr, so substituting Vr=Vrmax and λp=c/m gives md=2m/cVrmax.

Since md at this time should be 1/T, 1/T=2m/cvrmax Therefore, m=c/2Vrmax can be obtained.

As an example, assuming that the maximum speed in the radar axis direction of the aircraft target is 3 Matsuha, and substituting T = 5 ms for trial calculation, m = 30.2 MHz. Therefore, in the case of a pulse radar that uses a frequency of c=3000MHz as a carrier wave, the clutter spectrum will be compressed to m/c≈30MHz/3000MHz=1/100. In addition, the target detection characteristics are 1/T = 200Hz at Vrmax = 3 Matsuha.
Since the maximum point of Doppler frequency deviation will occur at this point, no loss will occur in the target detection characteristics.

Also in the case of this embodiment, if the MTI characteristics are shaped using multiple delay lines as shown in Figure 4, the compression effect of the clutter spectrum and the detection characteristics of the target signal can be further improved. will be done. In addition, the modulation frequency in the above embodiment
30MHz is an easily implementable frequency.

In the above embodiment, a case was described in which frequency modulation was applied to the transmission pulse, but even if other amplitude modulation or phase modulation is used, the same effect as in the above embodiment can be obtained in principle.

〔Effect of the invention〕

As described above, according to the present invention, the pulse transmission wave of the pulse radar device is subjected to frequency modulation and transmitted, the demodulated wave of the modulated signal is extracted from the received pulse signal, and the demodulated wave is moved with respect to this demodulated wave. Since the target detection function is activated, the clutter spectrum can be compressed without impairing the moving target detection characteristics, and the fixed target cancellation ratio can be dramatically improved compared to the conventional one. be.

[Brief explanation of the drawing]

Fig. 1 shows the configuration of a conventional pulse radar device, Fig. 2 shows the response characteristics of a conventional MTI, and Fig. 3 shows the clutter response characteristics of a conventional MTI.
FIG. 4 is a diagram showing a conventional shaped MTI response characteristic, FIG. 5 is a diagram showing the configuration of a pulse radar device according to an embodiment of the present invention, and FIG. 6 is a diagram showing the configuration of a pulse radar device according to an embodiment of the present invention. Figure 7 shows the spectrum of the pulsed FM wave that has undergone Doppler frequency shift in the device. Figure 8 shows the pulse shape of the device.
A diagram showing an example of the configuration of an FM demodulator, FIG.
Figure 10 is a diagram showing the input/output characteristics of the frequency discriminator in the demodulator, and Figure 10 is a diagram showing its output voltage waveform.
FIG. 1 is a diagram showing the clutter spectrum compression effect of a pulse radar device according to an embodiment of the present invention. 23...Frequency modulator (transmission wave frequency modulation means),
40...pulse type FM demodulator (modulation signal demodulation means),
50...MTI (moving target detection means). In the figures, the same reference numerals indicate the same or equivalent parts.

Claims (1)

[Claims] 1. Transmission wave frequency modulation means that performs frequency modulation on a pulse transmission wave having a predetermined carrier frequency, and a modulation signal that extracts only the pulse carrier wave component and pulse upper and lower sideband components from the received pulse signal component. demodulating means; receiving a modulated frequency component m and a frequency component d corresponding to the speed of the moving target, which are the outputs of the demodulating means;
comprising a phase detector that eliminates the modulation frequency component m and outputs only the Doppler frequency component d, and a moving target detection means that receives the output of the phase detector and extracts a change in the amplitude component corresponding to the Doppler frequency component, A pulse radar device characterized in that it detects reflected waves of.