JPH0332313B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION (a) Description of the Technical Field The present invention relates to a drive control device for a synchronous motor that is capable of controlling active power and reactive power to arbitrary values as seen from the power source side.

(b) Description of Prior Art FIG. 1 is a configuration diagram of a conventional drive control device for a synchronous motor. In the figure, BUS is the electrical line of the 3-phase AC power supply, CAP is the phase advance capacitor, SSL is the reactive power compensator, L is the DC reactor, TR is the power transformer, CC is the cycloconverter, SM is the synchronous motor body, and Load is the machine. It's a load. Also, PS is the rotor position detector, PG is the rotation pulse generator, PTG is the three-phase sine wave generator, VR〓 is the speed setting device, H _N (s)
is a control compensation circuit, ML _U , ML _V , ML _W are multipliers,
C _N , C _U , C _V , _CW are comparators, K _U , K _V , K _W are operational amplifiers, PH _U , PH _V , PH _W are phase control circuits,
CT _M is a motor input current detector.

The cycloconverter CC converts constant voltage, constant frequency AC power into variable voltage, variable frequency AC power, and connects the 3-phase armature winding of the synchronous motor SM to
It is controlled to supply a sinusoidal current synchronized with the back electromotive force induced in the motor.

The operation will be briefly explained below.

PS is the rotor (field) position detector, and the motor SM
generates a rectangular wave signal synchronized with the back electromotive force (speed electromotive force) induced in the armature winding. in electrical angle
The three-phase rectangular wave signal shifted by 120 degrees is input to the three-phase sine wave generator PTG and converted into a three-phase unit sine wave.

On the other hand, the electric motor is controlled by the rotating pulse generator PG.
The rotational speed ω of the SM is detected and compared with the command value ω ^* from the speed setting device VRω. Output of comparator C _N ε _N = ω ^*
-ω, the amplitude value of the armature current supplied to the motor SM is controlled. H _N (s) is a control compensation circuit for a speed control system, and an integral element is often used so that the steady-state value of the deviation ε _N becomes zero. The output of H _N (s) gives the amplitude value I ^* _O of the current command value.

The multiplier ML _U multiplies the above amplitude value I ^* _O and the above 3
Multiply the output of the phase sine wave generator PTG by the U phase component.
As a result, I ^* _U I=I ^* _O ·sinωt is obtained as the command value of the current _IU to be supplied to the U-phase armature winding. Similarly, the V-phase current command value I ^* _V and the W-phase current command value I ^* _W are each given as follows.

I ^* _V = I ^* _O・sin (ωt−2π/3) I ^* _W = I ^* _O・sin (ωt + 2π/3) where ω is the rotational angular frequency of the electric motor SM.
CT _M is a current transformer for detecting armature currents I _U , I _V , I _W , and the detected values are input to comparators C _U , C _V , _CW , and the above command values I ^* _U , I Compare with ^* _V and I ^* _W . Each deviation ε _U = I ^* _U − I _U , ε _V = I ^* _V − I _V , ε _W = I ^* _W − I _W is amplified by operational amplifiers K _U , K _V , K _W and phase control circuit _PHU ,
Input to PH _V and PH _W.

FIG. 2 shows a typical configuration diagram of a three-phase output cycloconverter, and the portion surrounded by a broken line corresponds to CC in FIG. 1. CC-U is a U-phase cycloconverter, which consists of a positive group converter SSP and a negative group converter SSN, and supplies a U-phase armature current _iU . When a forward current i ⁺ _U is supplied from SSP in a so-called non-circulating current type cycloconverter,
SSN is gate-blocked, and conversely, when SSN supplies negative current i ^- _U , SSP is gate-blocked. from SSP to SSN, or
When the operation changes from SSN to SSP, the current is temporarily reduced to zero and the switch is made to prevent a short circuit in the power supply.

The phase control circuit PH _U shown in Fig. 1 controls the firing phase of the U-phase cycloconverter CC-U shown in Fig. 2, and the voltage proportional to the aforementioned deviation ε _U = I ^* _U − I _{U is} in the positive group. Converter SSP or negative group converter SSN
The ignition phase angle α _U is controlled so that it occurs from .

CC-V and CC-W are similarly controlled.

In this way, the currents I _U , I _V , I _W flowing through the armature windings of the motor SM are set to the command values of the currents I ^* _U , I ^* _V ,
Controlled to be equal to I ^* _W.

Further, speed control is performed as follows.

In other words, when ω ^* > ω, ε _N = ω ^* −ω becomes a positive value, increasing the current amplitude value I ^* _O and increasing the motor SM
Increases the generated torque and accelerates. On the contrary ω ^*
<ω, then ε _N <0, reducing I ^* _O ,
Deceleration is performed by reducing the torque generated by the SM. As a result, it settles down to ω ^* ≒ ω.

The conventional drive control device for a synchronous motor as described above has a drawback that the reactive power at the power receiving end of the cycloconverter fluctuates greatly as the speed of the motor is controlled. The reactive power fluctuations cause voltage fluctuations in the power supply system, causing adverse effects such as flicker failure on other electrical equipment.

Therefore, recently it has been proposed to install a reactive power compensator SSL at the power receiving end, as shown in FIG. First of all, this is the phase advance capacitor CAP
Taking a constant leading reactive current Icap, the lagging reactive current of the cycloconverter Icc(REACT),
The sum of the delayed reactive currents _IQ of the compensator SSL is the above Icap
The current _IQ of the compensator SSL is controlled so that the current IQ becomes equal. That is, when Icap=Icc(REACT)+ _IQ is satisfied, the power supply current _IS does not include a reactive component, and the fundamental wave power factor at the receiving end can be controlled to 1.

However, since the delayed reactive current Icc (REACT) of the cycloconverter CC varies greatly from zero to the maximum value, the capacity of the compensator SSL needs to be comparable to the capacity of the cycloconverter. As a result, the capacity of the semiconductor elements that make up SSL also increases, resulting in a very expensive system.

(c) Purpose of the Invention The present invention has been made in view of the above, and has an object to make it possible to make the fundamental wave power factor at the receiving end of a cycloconverter 1 without providing an expensive reactive power compensator that is conventionally required. The object of the present invention is to provide a drive control device for a synchronous motor. Another object of the present invention is to provide a synchronous motor drive control device that can control the value of active power exchanged between an AC power supply and a synchronous motor to an arbitrary value.

(d) Configuration of the Invention FIG. 3 is a configuration diagram showing an embodiment of the synchronous motor drive control device of the present invention. In the diagram, BUS is 3
Electrical line of phase AC power supply, CAP is phase advancing capacitor,
TR is a power transformer, CC is a cycloconverter,
SM is the synchronous motor body, and Load is the mechanical load.

Also, PS is rotor position detector, PG is rotation pulse generator, F/V is frequency-voltage converter, PTG
is a three-phase sine wave generator, CT _S and CT _M are current detectors,
PT _S is a voltage detector, PQC is an active/reactive power calculation circuit, VR _Q is a reactive power setter, VR is a rotation speed setter, C ₁ , C ₂ , C _U are comparators, H _Q (s), H _P
(s), H _N (s) are control compensation circuits, ML and ML _U are multipliers, DVM is a divider, LIM ₁ and LIM ₂ are limiter circuits, K _U is an operational amplifier, and PH _U is a phase control circuit. be.

The current control circuit CONT-U surrounded by the broken line is
This shows a current control circuit for the U phase of the cycloconverter CC, and although not shown, the current control circuits for the V and W phases are similarly configured.

The following explanation will be given assuming that the cycloconverter CC has the configuration shown in FIG. 2, for example.

First, the three-phase sine wave generator PTG will be explained.

FIG. 4 shows a specific embodiment of the three-phase sine wave generator PTG. In the figure, SQ is a square calculation circuit, SQR is a square root calculation circuit, V/F is a voltage-frequency converter,
PC is a counter, ROM is a memory circuit, D/A is a digital to analog converter, PLL is a phase synchronization circuit,
_M1 to _M4 are multipliers, _A1 to _A6 are adders, and _K1 and _K2 are operational amplifiers.

The inputs are the three-phase rectangular wave signal γ _O from the rotor position detector PS, the rotational speed ω from the rotary pulse generator PG, and the cosine value cosγ of the armature current phase angle γ with respect to the back electromotive force of the motor, and the output is 3-phase unit sine wave φ _U ,
φ _V and φ _W.

First, the output signal (pulse train) of the PG is converted into a voltage amount ω (rotational speed detection value) via an F/V converter. The detected rotational speed value ω is input to the V/F converter via the adder _A2 . Also phase synchronization circuit
The correction amount Δω output from the PLL is also input to the V/F converter via the adder _A2 . Therefore, the V/F converter generates a pulse train with a frequency proportional to the input ω+Δω, and counts the counter PC. Next, the PC's count value is made to correspond to the address of the memory circuit ROM, and its contents are output. The storage circuit stores a sine function sin θ, a cosine function cos θ, and a three-phase rectangular wave function, and generates a digital quantity according to the count value of the counter PC. The digital amount
sin θ and cos θ are converted into analog quantities φ _a and φ _b by a D/A converter.

The above correction amount Δω is 3 from the rotor position detector PS.
This is intended to achieve phase synchronization between the phase rectangular wave signal γ _O and the three-phase rectangular wave signal γ _O ' output from the storage circuit ROM, and is determined by the phase synchronization circuit PLL.

FIG. 5 is a time chart for explaining the operation of the phase synchronization circuit PLL. The rotor position detector PS outputs three-phase rectangular wave signals _PSU , _PSV , and _PSW . Moreover, PS _U ′, PS _V ′, and PS _W ′ are three-phase rectangular wave signals generated from the storage circuit ROM.
The figure shows a steady state, where PS _U ′ stabilizes 90° behind PS _U. PS _V ′ and PS _W ′ are also PS _V , PS _W
90° behind. EX _U is the exclusive OR of PS _U and PS _U ′. Similarly, EX V is the exclusive OR of PS _V and PS _V ′, and EX _V is the exclusive OR of PS _W and PS _W ′.
Expressed as EX _W. These EX _U , EX _V and
When EX _W signals are added in an analog manner, the result is as shown by SUM. This can be obtained by adding a negative bias voltage by an average value of 1.5 and integrating it. Here, as shown by the broken line _, PS _U
Consider the case where the delay is δ. “2” of SUM
The period of becomes longer by δ, and as a result, the integral value becomes
It increases by Δω like SUM′. Therefore, the fourth
The counter PC in the figure counts faster by the correction amount Δω, advancing the phases of PS _U ′, PS _V ′, PS _W ′, etc. It is controlled so that δ=0 finally. vice versa
When PS _U ′ leads PS _U , Δω becomes a negative value, and the counter PC is slowly counted by the correction amount Δω, delaying the phase of PS _U ′, and finally δ=0. control so that

φ _a and φ _b in Fig. 5 indicate the output _signals of the D _{/ A} converter in a steady _state . Output in phase. In a steady state where phase synchronization is achieved with the signal from the position detection value PS, φ _a is in phase with PS _U.

The two-phase unit sine wave φ _a obtained in this way,
φ _b is input to multipliers M ₁ to M ₄ in FIG. 4.

On the other hand, the cosine value cosγ of the control phase angle γ is subjected to the following calculation via the square calculation circuit SQ, the adder _A1 , and the square root calculation circuit SQR to obtain the sine value sinγ.

sinγ=√1−() ² The corresponding cosine value cosγ and sine value sinγ are applied to the multiplier _M1
Enter ~ _M4 . The output φ _a · cos γ of M ₁ and the output φ _b · sin γ of M ₂ are input to adder A ₃ to obtain φ _a ′ = φ _a · cos γ + φ _b · sin γ. Further, the output φ _b ·cosγ of M ₃ and the output φ _a ·sinγ of M ₄ are input to the adder A ₄ to obtain φ _b ′=−φ _a ·sin γ + φ _b ·cos γ.

The following operational amplifiers K ₁ (=1/2) and K ₂ (=√3/
2) and adders A ₅ and A ₆ are for converting from two phases to three phases. That is, φ _U , φ _V , φ _W are φ _a ′,
It is obtained from φ _b ′ by performing the following calculation.

φ _U =φ _a ′ φ _V =−(1/2)φ _a ′−(√3/2)φ _b ′φ _W =−(1/2)φ _a ′+(√3/2)φ _b ′ This is the output signal of the three-phase sine wave generator.

Figure 6 shows the above unit sine waves φ _a , φ _b and φ _a ′, φ _b ′
, and a vector diagram of φ _U , φ _V , and φ _W .

φ _a ′ and φ _b ′ are delayed from φ _a and φ _b by a phase angle γ, respectively, and φ _U , φ _V , and φ _W are created from the corresponding φ _a ′ and φ _b ′.

As mentioned earlier, φ _a is controlled to be in phase with PS _U , so φ _U becomes a sine wave that lags PS _U by the phase angle γ. Similarly, φ _V and φ _W are each PS _V
And PS becomes a sine wave delayed by phase angle γ from _W.
That is, by changing the value of input cosγ,
The phase angle γ can be given an arbitrary value to the output signal of the position detector PS. By the way, cosγ=
1, γ=0, φ _a and φ _a ' match, and the conventional synchronous motor drive device operates at a constant value with γ _O =0°.

Returning to FIG. 3, the explanation of the device of the present invention will be continued. The three-phase power supply voltage and three-phase power supply current are detected by the transformer _PTS and current transformer _CTS installed at the receiving end of the three-phase power supply. PQC is the active power P and reactive power Q at the receiving end from the voltage detection value and current detection value.
This is a computing unit that calculates That is, active power P is obtained by multiplying the instantaneous values of voltage and current and adding the three phases, and reactive power Q is obtained in the same manner by shifting the voltage phase by 90 degrees.

The detected value Q of the above reactive power is determined by the comparator _C1 .
and its command value Q ^* , and calculate the deviation ε _Q = Q ^* −Q
is input to the control compensation circuit H _Q (ω). H _Q
(s) is usually _expressed as
Integral elements are used. The output of H _Q (s) is the motor
Amplitude values of armature currents I _U , I _V , I _W supplied to SM
It becomes I ^* _O .

In addition, the comparator _C2 compares the detected speed value ω of the motor SM with its command value ω ^* , and the deviation ε _N =
ω ^* −ω is input to the control compensation circuit H _N (ω).
The output of H _N (s) is passed through the limiter circuit LIM ₂ ,
It is assumed that the generated torque command value of the electric motor SM is T ^* . The detected speed value ω and the torque command value T ^* are input to the multiplier ML, and the active power command value P ^* =ω·T ^* is determined.

The comparator _C3 compares the active power detection value P and the command value P ^* , and inputs the deviation ε _P =P ^* -P to the next control compensation circuit H _P (s). H _P (s)
An integral element is generally used to make the steady-state deviation ε _P zero. The output v _P of H _P (s) is divided by the following divider
Input to DVM and find cosγ=v _P・(V _S /Vm). Here, V _S is the power supply voltage, and Vm is the motor terminal voltage. The next limiter circuit LIM ₁ is for limiting the cosine value cosγ of the phase angle γ to a range of −1cosγ1.

The corresponding cosine value cosγ is converted to the following three-phase sine wave generator PTG
By inputting _to
As mentioned above, φ _V and φ _W can be obtained.

The U-phase unit sine wave φ _U and the current amplitude value I ^* _O are input to a multiplier M _U to obtain a U-phase armature current command value I ^* _U expressed by the following equation.

I ^* _U = I ^* _O・φ _U = I ^* _O・sin (ωt−γ) On the other hand, the U-phase armature current I _U is detected by the current transformer CT _M , and _the above Compare with command value I ^* _U. The deviation ε _U = I ^* _U − I _U is amplified by the operational amplifier K _U , and the phase control circuit PH _U of the U-phase cycloconverter is
Enter. The output voltage of the U-phase cycloconverter is controlled so that the armature current _IU becomes equal to the command value I ^* _U . The V-phase and W-phase cycloconverters are similarly controlled. At this time, the V-phase and W-phase armature current command values are each expressed by the following equations.

I ^* _U = I ^* _O・φ _V I ^* _O・sin (ωt−γ−2π/3) I ^* _W = I ^* _O・φ _W I ^* _O・sin (ωt−γ+2π/3) Synchronous motor SM The generated torque Te can be expressed as follows.

Te=k _e・I・I _O・cosγ Where, k _e is the proportionality constant, I is the field current I _O is the peak value of the armature current, and the output P _O of the motor SM is the rotational angular frequency ω and the above generation As is well known, it is expressed as the product of torque Te. The effective power P supplied from the power source is the output P _O plus various losses such as copper loss and iron loss due to armature resistance. Here, for convenience of explanation, the various losses described above are ignored as being small. In other words, consider P≒P _O.

When the active power P at the receiving end is smaller than its command value P ^* , ε _P =P ^* -P becomes a positive value, and v _P is increased via the control compensation circuit H _P (s). Therefore cosγ
increases, increasing the generated torque Te of the electric motor SM. As a result, the output P _O =ω·Te of the electric motor SM increases, and the effective power P at the receiving end also increases. Finally P ^* ≒ P
I feel calm. Conversely, if P ^* < P, ε _P
< 0, reducing cosγ and P≒P _O ,
After all, it settles down to P ^* ≒ P.

(e) Effect of the invention Looking at this from the standpoint of controlling the rotational speed of the electric motor SM, it is as follows.

If the speed command value ω ^* is larger than the speed detection value ω,
The deviation ε _N = ω ^* −ω is a positive value, and the control compensation circuit
H _N (s) and the torque command T ^* obtained via the limiter circuit LIM ₂ are increased. As a result, the active power command value P ^* =ω·T ^* increases, causing the generated torque Te of the electric motor SM to increase. Therefore, the electric motor SM is accelerated and settles down to ω ^* ≒ ω. On the contrary, ω ^* ＜
In the case of ω, ε _N <0, which reduces T ^* and lowers P ^* . As a result, the torque generated by the electric motor SM decreases and is decelerated, settling down to ω ^* ≒ ω.

On the other hand, the cycloconverter CC seen from the power supply side
The delayed reactive power Q _CC can be expressed as follows.

Q _CC =k _S・V _S・(｜I _U ｜・sinα _U +｜I _V ｜・sinα _V +｜I _W ｜・sinα _W ) where k _S is the proportionality constant, V _S is the power supply voltage, |I _U |, |I _V |, |I _W | are the absolute values of the armature currents, α _U , α _V , α _W are the firing phase angles, where the proportionality constant k _S and the power supply voltage V _S are constant values,
Assuming that the armature currents I _U , I _V , I _W are controlled equal to their command values I ^* _U , I ^* _V , I ^* _W , I _U = I _O・sin(ωt−γ) I _V =I _O・sin(ωt−γ−2π/3) I _W =I _O・sin(ωt−γ+2π/3). Therefore, Qcc is Q _CC =k _S・V _S・I _O {|sin(ωt−γ)|・sinα _U +|sin(ωt−γ−2π/3)|・sinα _V +|sin(ωt−γ −2π/3) |・sinα _W.

At this time, the terminal voltages of each phase of the armature winding of the motor SM, V _U , V _V , and V _W , are expressed as in the following equations.

V _U =Vm・sinωt=k _V・cosα _U V _V =Vm・sin(ωt−2π/3)=k _V・V _S・cosα _V V _W =Vm・sin(ωt+2π/3)=k _V・V _S・cosα _W Therefore, the firing phase angles α _U , α _V , α _W are α _U = cos ^-1 {Vm/k _V V _S・sinωt} α _V = cos ^-1 {Vm/k _V V _S・sin (ωt−2π/3)} α _W =cos ⁻¹ {Vm/k _V V _S ·sin(ωt+2π/3)}.

The terminal voltage peak value Vm of the armature winding is proportional to the product of the rotational speed ω of the motor SM and the field current I. Therefore, if I=constant, when the motor SM starts at low speed, the value of Vm is small, and α _U ≒ α _V ≒ α _W ≒ 90°. As ω increases, α _U , α _V , α _W become
It begins to fluctuate in the direction of 0° or 180° with 90° as the center.

FIG. 7 shows the firing phase angle α U and its sine value sin _{α U with} respect to the armature voltage V _U and armature current I _U for the U-phase cycloconverter.
At the time of starting, the peak value Vm≈0, so the ignition phase angle α _U becomes a constant value of 90° as shown below. Therefore sinα _U
indicates a constant value of 1 like ''. As the speed increases, Vm increases and α _U fluctuates as →→. Therefore, sinα _U also ′→′→
'. Actually, since the cycloconverter CC commutates depending on the power supply voltage (natural commutation method), the firing phase angle α _U is controlled within the range of α limit ≦ α _U ≦ β limit. Usually αlimit≒20° and βlimit≒150° are selected. Therefore, it can be considered that even at the maximum speed, the curve fluctuates at most.

The reactive power consumed by the U-phase cycloconverter is Q _CC-U =k _S・V _S・｜I _U |・sinα _U =k _S・V _S・I _O・｜sin(ωt−γ)｜・sinα _U becomes. At startup, α _U ≒90° = constant, so
Although it is not related to the phase difference γ between voltage V _U and current I _U ,
As the speed increases, α _U , that is, sin α _U changes, so it becomes influenced by γ. As can be seen from Figure 7, when γ = 90°, Q _CC-U becomes maximum. Also, when γ=0°, Q _CC-U becomes minimum.

The same applies to the V phase and W phase.

These can be summarized as follows.

Q _CC is proportional to the current peak value I _O. In addition, Q _CC is large when the rotational speed ω of the motor is low, and becomes smaller as ω becomes higher. Furthermore, Q _CC has a minimum value when γ=0° or γ=180°, and a maximum value when γ=90°.

The reactive power Q at the receiving end is the delayed reactive power Q _CC of the relevant cycloconverter CC and the phase advance capacitor CAP.
is the sum of the leading reactive power Qcap. If the reactive power detection value Q (makes the delay a positive value) at the relevant power receiving end is smaller than the command value Q ^* = 0 (in the case of lead), the deviation ε _Q
=Q ^* -Q becomes a positive value and increases the current peak value _IO . Therefore, Q _CC increases and is controlled so that Q≈Q ^* =0. However, when I _O increases, the generated torque Te of the electric motor SM increases, and the active power P at the receiving end also increases. As a result, P ^* < P, and
The control system works so that ε _P becomes negative and cos γ decreases, so that P ^* ≒ P. Therefore, this time γ approaches 90°, further increasing Q _CC . Therefore, Q>
Q ^* = 0 and I _O decreases slightly. After repeating this vibration phenomenon several times, finally Q ^* ≒ Q, P ^*
It settles on a new peak value I _O and phase angle γ such that =P. The same applies when Q>Q ^* .

Furthermore, the vibration phenomenon described above also occurs when controlling the effective power P. It goes without saying that the compensation constants of the control compensation circuits H _Q (s) and H _P (s) are selected to be optimal values in order to reduce and calm down this vibration phenomenon.

(f) Other Embodiments FIG. 8 is a block diagram showing another embodiment of the apparatus of the present invention. Only the parts that differ from FIG. 3 will be explained.
VAR is a reactive power calculation circuit, SQ ₁ and SQ ₂ are square calculation circuits, A ₁ is an adder, SQR is a square root calculation circuit,
DIV is a divider.

The reactive power calculation circuit VAR calculates the reactive power Q from the voltage and current detection values at the receiving end. The reactive power detection value Q and the command value Q ^* are compared by the comparator C ₁ , and the deviation ε _Q = Q ^* −Q is calculated by the control compensation circuit H _Q
(s). The output I ^* _Q of H _Q (s) becomes the reactive current command value. Furthermore, the output I ^* _P of the multiplier ML becomes the effective current command value. square calculation circuit SQ ₁ and
Square it by SQ ₂ , input it to the adder, and use it as the armature current peak value I ^* _O of the motor SM via the next square root calculation circuit SQR. That is, I ^* _O = √ ( ^* _P ) ² + ( ^* _a ) ² . Also, the above I ^* _O result and the above effective current command value
Input I ^* _P to the divider DIV, calculate v _P = I ^* _P / I ^* _O , and pass it through the next divider DVM and limiter circuit LIM ₁ ,
It is taken as the cosine value cosγ of the phase delay angle γ of the armature current.

When Q ^* > Q, ε _Q becomes positive and increases I ^* _Q. Therefore, I ^* _O = √ ( ^* _Q ) ² + ( ^* _P ) ² increases, and the delayed reactive power Q _CC of the cycloconverter CC increases and Q ^*
It is controlled so that ≒Q. At this time, as I ^* _O increases, v _P =I ^* _P /I ^* _O decreases, reducing cosγ. As mentioned before, the output P _O of the electric motor SM is P _O = ω・Te=ω・_ke・I・I _O・cosγ, so if we decrease cosγ by the amount that I _O increases, we get P _O
= held constant. Therefore, increasing I ^* _Q has no effect on the active power control system.

Next, consider the case where I ^* _P is increased. As a result
I ^* _O = √ ( ^* _P ) ² + ( ^* _Q ) ² increases, cosγ = I ^* _P / I
^* _O also increases. Therefore, the output P _O of the electric motor SM increases in proportion to I * _P as P _O = ω・_ke・I・I _O・cosγ = ω・_ke・I^・I ^* _P. At this time, the delayed reactive power Q _CC of the cycloconverter CC increases as I ^* _O increases, and decreases as cosγ increases. Therefore, changing I ^* _P has little effect on Q _CC . Further, even if the reactive power control system is affected, the active power control system is not affected, so that it is easier to handle than the control system shown in the embodiment of FIG. In other words, a control system with less mutual interference can be obtained.

Note that the cycloconverter CC is not limited to a non-circulating current type cycloconverter, and it goes without saying that it can be similarly achieved with a partially circulating type cycloconverter or a circulating current type cycloconverter.

(g) Effects of the Invention As described above, the synchronous motor drive control device of the present invention can eliminate reactive power fluctuations at the receiving end of a cycloconverter without providing an expensive reactive power compensator that is conventionally required. Furthermore, by combining it with a phase advance capacitor that takes a constant amount of advance reactive power, it has the advantage that the fundamental wave power factor at the receiving end can always be controlled to 1. Furthermore, since the value of active power exchanged between the AC power source and the synchronous motor is directly controlled, power running and regenerative operation can be easily performed, and operation under constant power control or constant torque control can also be performed using conventional devices. It has the advantage of being easier. Especially in speed control, by giving the power command value P ^* as the product of the generated torque command value T ^* and the rotational speed ω, it becomes possible to control the generated torque of the electric motor without being affected by the rotational speed ω. , the speed control response is improved.

[Brief explanation of the drawing]

FIG. 1 is a block diagram of a conventional synchronous motor drive control device, FIG. 2 is a typical block diagram of the cycloconverter shown in FIG. 1, and FIG. 3 is an embodiment of the synchronous motor drive control device of the present invention. FIG. 4 is a configuration diagram showing a specific example of the three-phase sine wave generator shown in FIG. 3, FIG. 5 is a time chart for explaining the operation of FIG. 4, and FIG. The figure is a vector diagram for explaining the operation of the embodiment shown in FIG. 4, FIG. 7 is a waveform diagram of voltage current and phase angle for explaining the operation of the embodiment of FIG. 3, and FIG. FIG. CAP...phase advance capacitor, TR...power transformer,
CC...cycloconverter, SM...synchronous motor,
Load…mechanical load, PS…rotor position detector, PG
…Rotary pulse generator, CT _S , CT _M …Current transformer, PT _S
...Transformer, PQC...Active power reactive power calculation circuit,
H _Q (s), H _P (s), H _N (s)...control compensation circuit,
ML, ML _U ...multiplier, DVM...divider, LIM ₁ ,
LIM ₂ …Limiter circuit, C ₁ to C ₃ …Comparator, VR _Q …
Reactive power setting device, VR〓...Speed setting device, K _U ...Operation amplifier, PH _U ...Phase control circuit, PTG...3-phase sine wave generator, C _U ...Comparator, F/V...Frequency-voltage converter, VAR...Reactive power calculation circuit, SQ ₁ , SQ ₂ ...
Square calculation circuit, _A1 ...adder, SQR...square root calculation circuit, DIV...divider.

Claims (1)

[Scope of Claims] 1. A synchronous motor, a cycloconverter that is interposed between the synchronous motor and an AC power source and supplies variable voltage and variable frequency power to the synchronous motor, and an armature current that is supplied to the synchronous motor. In order to control the reactive power at the receiving end of the cycloconverter by controlling the peak value of In order to control the generated torque of ^the synchronous motor, the phase angle γ of the armature current with respect to the speed electromotive force induced in the armature winding of the synchronous motor or the position of the rotating magnetic pole is provided _. means for giving the current peak value command;
means for determining a current command value to be supplied to the armature winding of the synchronous motor based on I ^* _O and current phase angle γ; and means for controlling the armature current of the synchronous motor in accordance with the current command value. A drive control device for a synchronous motor, comprising: 2. A synchronous motor, a cycloconverter that is interposed between the synchronous motor and the AC power source and supplies variable voltage and variable frequency power to the synchronous motor, and a cycloconverter that controls reactive power at the receiving end of the cycloconverter. Means for giving a current command I ^* _Q and an effective current command for controlling the generated torque of the synchronous motor
I ^* _P , a peak value command I ^* _O of the current supplied to the armature winding of the synchronous motor from the reactive current command I ^* _Q and the active current command I ^* _P , and a speed electromotive force of the synchronous motor. or means for giving the phase angle γ of the armature current with respect to the position of the rotating magnetic pole, and the wave height command I ^* _O
and means for determining a command value of the current to be supplied to the armature winding of the synchronous motor based on the current phase angle γ, and means for controlling the armature current of the synchronous motor in accordance with the current command value. A drive control device for a synchronous motor consisting of: