JPH0334314B2 - - Google Patents

Description

[Detailed Description of the Invention] [Technical Field of the Invention] The present invention is a slip frequency control device for an induction motor, which prevents a decrease in the output of the induction motor by correcting the slip frequency according to the current control result of the induction motor. This invention relates to a slip frequency control method.

[Technical background of the invention and its problems]

Figure 1 is a circuit configuration diagram showing an equivalent circuit for one phase of an induction motor when the induction motor is subjected to slip frequency control.As is clear from the figure, the motor excitation inductance 1 and the secondary equivalent load 2 of the motor It consists of parallel circuits. Here, the motor excitation inductance 1 has a value of L ₀ and the motor secondary equivalent load 2 has a value of R ₂ /S.
Note that R ₂ is the motor secondary winding resistance, and S is the slip of the motor. On the other hand, the motor primary winding resistance, primary leakage inductance, excitation conductance, and secondary winding inductance can be ignored if the control is simple.

In FIG. 1, an angular frequency ω ₁ between terminals A and B,
When an AC voltage of magnitude E ₁ is applied, the current flowing through the motor excitation inductance 1, that is, the motor excitation current I ₀ is I ₀ =E ₁ /ω ₁ L ₀ (1). Moreover, the current I ₂ flowing through the motor secondary winding is I ₂ =E ₁ /R ₂ /S=SE ₁ /R ₂ (2). As can be seen from Figure 1, the exciting current _I0 is the current flowing through the inductance load, so it lags 90 degrees with respect to the applied voltage _E1 , and the current _I2 flowing through the secondary winding is the current flowing through the resistive load. Therefore, it is in phase with the applied voltage _E1 . On the other hand, since the magnitude Φ of the rotating magnetic field within the motor is proportional to the exciting current I ₀ , Φ=K ₁ I ₀ (3). However, K ₁ is a proportionality constant. here,
Magnetic flux magnitude Φ at any arbitrary frequency
In order to control the excitation current I ₀ to be constant, in other words, to keep the excitation current I 0 constant, E ₁ /
It is sufficient to control so that ω ₁ is constant. On the other hand, since the torque T generated by the motor is proportional to the product of the magnitude of the magnetic flux Φ and the current I ₂ flowing through the secondary winding conductor in the magnetic flux, T=K ₂ ΦI ₂ (4). However, K ₂ is a proportionality constant. Therefore,
When the magnitude of magnetic flux Φ is controlled to be constant, the magnitude of torque T is equal to the motor secondary winding current I ₂
It can be seen that it is proportional to the size of In addition, as mentioned above, the applied voltage E ₁ that fixes the frequency at any frequency is uniquely constant, so equation (2) becomes I ₂ = K ₃ S ...(5) at that frequency, and the motor The torque T is T=K ₄ S (6). Here, K ₃ is a constant, and K ₄ is a constant when the frequency is fixed. Also, the magnitude of the motor primary current I ₁ is the vector sum of the excitation current I〓 ₀ and the rotor current I〓 ₂ , and since the excitation current I〓 ₀ and the rotor current I〓 ₂ are orthogonal, I ₁ = √ ₀ ² + ₂ ² …(7). Therefore, if the excitation current I ₀ is controlled to be constant, that is, the magnetic flux is constant, and the torque T required by the motor is found, from equation (2), I ₂ = T / K ₂ Φ ... (8) Find _{I 2} so that
By providing a power supply frequency obtained by adding a slip frequency corresponding to the slip S obtained from the formula to the motor synchronous rotation frequency, the motor can generate the required torque and perform speed control.

Based on these properties of induction motors, induction motor control apparatuses have heretofore been used which control the slip frequency of induction motors using inverters, and FIG. 2 shows a block diagram thereof. In FIG. 2, a variable DC power supply 3 receiving power from an AC power supply 3A converts AC into DC and supplies it to an inverter circuit 6 via a reactor 4. Note that the inverter circuit 6 is composed of a plurality of switching elements and outputs three-phase alternating current of U phase, V phase, and W phase.
AC power from the inverter circuit 6 is supplied to the induction motor 7
is supplied to and drives it. The speed of the induction motor 7 is detected by a speed detector 8 and a speed detection signal is output.
Output as V _S. The speed setter 9 compares and calculates the speed setting signal _VSR and the speed detection signal _Vs to set the speed of the induction motor 7, and outputs a torque reference signal T ^* . The function generator 11 also outputs a torque reference signal.
The average current reference signal I ₁ ^* of the induction motor 7 is generated as a signal having the relationship of equation (7) with respect to T ^* . The current control circuit 12 compares and amplifies the current reference signal I ₁ ^* and the current output signal I ₁ of the current detector 5 that detects the main circuit current, and controls the voltage of the variable DC power supply 3 via the variable DC power supply control circuit 13. The motor current can be adjusted by varying the
Control I ₁ so that it becomes the current reference value I ₁ ^* . on the other hand,
The torque reference signal T ^* is converted via the regulator 14 into a voltage signal S _V corresponding to the slip frequency. This voltage signal S _V is summed with the speed detection signal V _S and input to the voltage-frequency converter 15 where it is converted into a frequency signal ₀ . Frequency signal obtained in this way
₀ is input to the AC voltage generation control circuit 16, and the inverter circuit 6 is controlled based on this to generate the necessary AC voltage.

Here _, the torque reference T ^* is determined by the motor 7
is proportional to the rotor current _I2 . Furthermore, the rotor current I ₂ of the motor 7 is proportional to the product of the slip S and the motor voltage as shown in equation (2), and is inversely proportional to the rotor winding resistance R ₂ of the motor 7, but at a fixed angular frequency ω ₁ In this respect, E ₁ /R ₂ is controlled by changing the slip S in proportion to the torque reference T ^* as a constant.

In general, in a Leonard device or inverter device that performs speed control by controlling current in a minor loop, the rated voltage of the DC or AC motor driven by the device is lower than the maximum output voltage of the device. In particular, the difference is expected to be even larger in order to achieve high-speed response. Increasing the difference between the device's maximum output voltage and the motor's rated output voltage means lowering the motor's rated voltage, since the device's maximum output voltage is determined by the magnitude of the device's input power supply voltage. Therefore, since the motor voltage is lowered and the motor rated current is increased, the current capacity of the main circuit switching element of the device increases, which increases the cost of the element and the price of the device. Therefore, the price of the device itself is greatly influenced by how small the difference between the maximum output voltage of the device and the rated voltage of the motor can be made without impairing high-speed response. Therefore, the device maximum output voltage and the motor rated voltage are set to approximately the same value.

On the other hand, in the slip frequency control device (5), (6),
Slip S, rotor current I ₂ , and primary current I ₁ are calculated from the required torque T using equation (7), and each value is controlled to generate torque T, but the device input power supply voltage may drop, etc. Therefore, when the maximum output voltage of the device decreases, the slip S is corrected for the required torque T, but the required primary current I ₁
does not flow as calculated, which may result in a torque shortage. The motor primary current I ₁ is the inverter output voltage, that is, the motor terminal voltage V 1 because the actual induction motor has a primary leakage inductance 17 and a primary winding resistance 18 as shown in Fig. 3, compared to the equivalent circuit shown in Fig. ₁ .
The voltage difference between E and motor induced voltage _E1 is determined by the primary leakage inductance and the impedance of the primary winding resistance. Therefore, when the inverter output voltage decreases due to a decrease in the inverter input power supply voltage, the required primary current cannot flow, resulting in a disadvantage that torque is insufficient.

[Purpose of the invention]

As described above, an object of the present invention is to provide a slip frequency control method that allows a drive motor to generate a desired torque even if the inverter output voltage decreases due to a decrease in the inverter input power supply voltage or the like.

[Summary of the invention]

In order to achieve this objective, the present invention detects that the current control system has reached the limit for increasing its output current, for example, when the deviation between the current reference and current feedback is at a predetermined value, and in this case, the slip frequency As a result, the applied frequency of the induction motor is increased to increase the torque.

[Embodiments of the invention]

FIG. 4 shows a block diagram of the present invention. Portions indicated by the same numbers as in FIG. 2 indicate the same parts. The output of the current control circuit 12 is input to the variable DC power supply control circuit 13 and also to the current control saturation detection circuit 19.
When the output of is above a certain value, a value corresponding to the difference is synthesized with the speed detection signal V _S and the voltage signal S _V corresponding to the slip frequency, and is input to the voltage-frequency converter 15 .

In FIG. 4, when the main circuit element of the variable DC power supply 3 is a thyristor, the variable DC power supply control circuit 13
is a thyristor phase control circuit, which operates by changing the firing phase of the thyristor to change the output voltage of the variable DC power supply 3 so that the current I ₁ becomes equal to the current reference I ₁ ^* . In other words, for the current reference I ₁ ^* ,
When the current I ₁ is small, the output of the current control circuit 12 becomes large, and the variable DC power supply control circuit 13 advances the firing phase of the thyristor of the variable DC power supply 3 to increase the output voltage of the variable DC power supply 3 and increase the current. It operates to increase I ₁ to bring it closer to the current reference I ₁ ^* . Similarly, when the current I ₁ is larger than the current reference value I ₁ ^* , the output of the current control circuit 12 becomes small, and the variable DC power supply control circuit 13 delays the firing phase of the thyristor of the variable DC power supply 3 to make the variable The output voltage of the DC power supply 3 is reduced, the power supply I ₁ is reduced, and the current reference I ₁ ^*
It operates in a way that brings it closer to . Therefore, if the current I ₁ does not flow as per the reference value due to a drop in the power supply voltage, etc., the output of the current control circuit 12 becomes even larger, and the variable DC power supply control circuit 13 controls the variable DC power supply 3.
An attempt is made to further advance the firing phase of the thyristor to increase the output voltage of the variable DC power supply 3 and increase the current _I1 , but even if the firing phase of the thyristor is advanced the most, the voltage of the AC power supply 3A itself Since the voltage is low, the output voltage of the variable DC power supply 3 does not increase, and as a result, the output of the current control circuit 12 increases to the maximum value of the control circuit output voltage.

Therefore, the maximum value of the current control circuit output voltage within the normal operating range is determined in advance, and when the current control circuit output exceeds the maximum value, an output voltage is generated in the current control saturation detection circuit 19 according to the proportion. Then, it is added together with the voltage signal S _V corresponding to the slip frequency and the speed detection signal V _S , resulting in an increase in the inverter output frequency. Therefore, although the output voltage of the induction motor is reduced, the induction motor power supply frequency, that is, the inverter output frequency is increased, so that the slip of the induction motor increases and the torque generated by the motor increases. That is, in general, in an induction motor, under the same load torque condition, the slip S decreases in proportion to the square of the voltage magnitude.
Therefore, as the inverter output voltage decreases due to a decrease in the input power supply voltage, etc., the current control saturation detection circuit output operates in the direction of increasing the inverter output frequency, increasing the slip of the motor and increasing the load torque. do. The increase in the slip frequency that increases the slip frequency due to the inverter output voltage drop above the value determined from the torque reference signal T ^* is the degree of the output voltage drop, that is, the difference between the current reference I ₁ ^* and the load current I ₁ It shall increase in proportion to the degree of difference.

In the following embodiments, the variable DC power supply 3 is shown as a thyristor, but as shown in Fig. 5, it includes a rectifying element 3B that converts AC power to DC, a switching element 3C such as a transistor that converts DC power to variable DC voltage, and a load. In order to keep the current constant, it is possible to replace the reactor 4 with a freewheel diode 3D for discharging its energy.
In this case, the variable DC power supply control circuit 13 determines the on period of the switching element 3C in a constant period in proportion to the magnitude of the output voltage of the current control circuit 12, and determines the pulse width that determines the magnitude of the output voltage of the variable DC power supply. It includes a modulation circuit and a drive circuit that turns on and off the switching element.

FIG. 6 shows a block diagram in which the present invention is applied to a sine wave current inverter. Portions indicated by the same numbers as in FIGS. 2 and 5 indicate the same parts. This system does not have a variable DC power supply, but instead consists of a rectifying element 3B that converts the AC power supply 3A into DC, and a capacitor 3E that absorbs the reactive power on the load side. Three sets of switching elements 6A, such as two pairs of transistors, are connected in parallel between the DC power source and the positive side bus line and the negative side bus line of the DC power source, and a rectifier element 6B is connected in antiparallel to each switching element. Further, two wires from the neutral points of the two pairs of switching elements are connected to the induction motor 7 via current detectors 20 and 21, and the other wire is directly connected to the induction motor 7 from the neutral points of the two pairs of switching elements. Ru.

In this block diagram, a sine wave current reference generation circuit 22 synthesizes a current reference signal I ₁ ^* and a frequency signal ₀ to generate a sine wave current reference I _R ^* whose current magnitude is I ₁ and output frequency is ₀ . let sinusoidal current reference
Generate a sinusoidal current reference I _S ^* with the same magnitude and 120° phase delay with respect to I _R ^* . The sine wave current reference I _R ^* is compared and amplified by the R-phase current control circuit 12A with the output I _R of the current detector 20, which is the R-phase load current, and becomes the R-phase voltage reference V _R ^* , which is pulse width modulated. The R-phase switching element is controlled via the circuit 24 so that the load current I _R flows in accordance with the current reference I _R ^* . Similarly, the sine wave current reference I _S ^* is compared and amplified by the S-phase current control circuit 12B with the output I _S of the current detector 21, which is the S-phase load current, to become the S-phase voltage reference V _S ^* , and the pulse width The S-phase switching element is controlled via the modulation circuit 25, and the load current is
U _S is controlled so that it flows in accordance with the current reference I _B ^* .

Since this inverter is a three-phase, three-wire load, the sum of the voltages of each phase at any given time is zero, so the T-phase voltage reference V _T ^* is from V _R ^* +V _S ^* +V _T ^* = 0.
It is necessary that V _T ^* =−(V _R ^* +V _S ^* ). Therefore, the T-phase voltage reference V _T ^* is generated by the voltage reference generation circuit 26 from the R-phase voltage reference signal V R *, which is the output of the R-phase current control circuit 12A, and the S-phase voltage reference signal V _R ^* , which is the output of the S-phase current control circuit 12B. The voltage reference signal V _S ^* is synthesized and generated, and the T-phase switching element is driven via the pulse width modulation circuit 27.

Here, since each output of the R-phase current control circuit 12A and the S-phase current control circuit 12B is a voltage reference signal of the R-phase and S-phase, it increases when the load current increases. Therefore, if the input power supply voltage drops and the load current no longer flows according to the current standard,
The voltage reference of each circuit increases, which acts to increase the inverter output voltage and increase the inverter output current. At the same time, the output voltage of the voltage reference generation circuit 26, which combines the R-phase and S-phase voltage references and generates the T-phase voltage reference, also rises. Therefore, the output voltage saturation detection circuit 19 detects a sudden increase in the voltage reference of each phase, and depending on the degree of increase in the voltage reference, that is, the degree of saturation of the output voltage, generates a voltage signal corresponding to the slip frequency. This is to correct the slip frequency by increasing a component S _V ' that increases the inverter output frequency together with S _V and the speed detection signal V _S.

In the above explanation, the voltage S _V corresponding to the slip frequency from the torque reference T ^* and the voltage corresponding to the slip frequency correction due to inverter output voltage saturation.
Although S _V ' is shown separately, the same effect can be achieved by varying the gain of the regulator 14 based on the output of the current control saturation detection circuit 19 as shown in FIG. When the current control is operating normally, that is, when the load current I ₁ has a one-to-one relationship with the current reference I ₁ ^* , the output of the current control saturation detection circuit 19 is not sent out; The gain of the regulator 14 is a gain that can output a slip frequency that matches the driving machine, and the voltage S _V corresponding to the slip frequency is output in proportion to the torque reference T ^* . In the range in which the current control saturation detection circuit 19 operates, that is, in the range where the load current I ₁ decreases with respect to the current reference I ₁ ^* , the current control saturation detection circuit 19 operates.
From the output of , the voltage S _V corresponding to the slip frequency is output in proportion to the torque reference T ^* as above, but
The ratio increases as the output of the current control saturation detection circuit 19 increases, that is, as the inverter output voltage becomes saturated.

Expressing these relationships in a formula, when the current control is operating normally, the voltage S _V corresponding to the slip frequency is S _V = K _S T ^* (9) where K _S is a constant. In the range in which the current control saturation detection circuit 19 operates, the voltage S _V corresponding to the slip frequency is controlled so that S _V =(K _S +ΔK _S )T ^* (10). Here K _S is in equation (9)
ΔK _S is a variable that changes depending on the saturation rate of current control, that is, the saturation rate of output voltage, and changes depending _on the output of the current control saturation detection circuit 19. That is, in equation (10), K _S T ^* is the original slip frequency calculation component S _V , and ΔK _S T ^* corresponds to the voltage S _V ' corresponding to the slip frequency correction due to inverter output voltage saturation.

In equation (10), the voltage S _V ′ corresponding to the slip frequency correction due to inverter output voltage saturation is ΔK _S T ^* , and although it changes depending on the magnitude of the torque reference T ^* , the original inverter output voltage drop Since the purpose is to prevent the motor output from decreasing in the torque reference T ^* , the motor output will not decrease and this will not be a problem. Therefore, if the torque reference T ^* is large, we only need to consider the case where the motor output is large, so ΔK _S T ^*
Of these, the torque reference T ^* can be considered a constant.

〔Effect of the invention〕

By implementing the present invention, it is possible to prevent a decrease in motor output due to a decrease in input power supply voltage, etc., without increasing the current capacity of the main circuit switching element, and as a result, it is possible to reduce the cost of the device.

[Brief explanation of the drawing]

Figure 1 is a simple equivalent circuit diagram for one phase in an induction motor, Figure 2 is a block diagram of a conventional slip frequency control device, and Figure 3 is a simple equivalent circuit diagram for one phase in the induction motor shown in Figure 1. Fig. 4 is an equivalent circuit diagram with added winding resistance and primary leakage inductance; Fig. 4 is a block diagram showing an embodiment of the present invention; Fig. 5;
FIGS. 6 and 7 are block diagrams showing other different embodiments of the present invention. 1...Induction motor excitation inductance, 2...Induction motor secondary equivalent load, 3...Variable DC power supply, 3A...
AC power supply, 3B... Rectifying element, 3C, 6A... Switching element, 3D, 6B... Freewheel diode, 3E... Capacitor, 4... Reactor, 5,
20, 21... Current detector, 6... Inverter circuit,
7... Induction motor, 8... Speed detector, 9... Speed setter, 10... Amplifier, 11... Function generation circuit, 12,
12A, 12B... Current control circuit, 13... Variable DC power supply control circuit, 14... Regulator, 15... Voltage-frequency converter, 16... AC voltage generation control circuit, 17...
Induction motor primary leakage inductance, 18... Induction motor primary winding resistance, 19... Current control saturation detection circuit, 22, 23... Sine wave current reference generation circuit, 2
4, 25, 27... Pulse width modulation circuit, 26... Voltage reference generation circuit.

Claims (1)

[Claims] 1 A speed control device that obtains a torque reference by integrating the speed difference between the commanded speed and the actual speed, and controls the frequency applied to the induction motor by the sum of the slip frequency obtained from the torque reference and the rotation frequency of the induction motor. In an induction motor control device that compares a current flowing through the induction motor with a current reference obtained from the torque reference and controls the current of the induction motor according to the deviation, the current control system of the induction motor controls the output current of the induction motor. When it is detected that the current reference and current feedback deviation signal exceeds a predetermined value, the current control saturation detection circuit generates an output voltage corresponding to the slip frequency according to the ratio. A slip frequency control method comprising: correcting the slip frequency and increasing an applied frequency of an induction motor to increase torque.