JPH0351038B2 - - Google Patents

Description

[Detailed Description of the Invention] This invention converts the output signal of a detector that detects physical quantities such as pressure, temperature, displacement, etc. into a corresponding current,
The present invention relates to a two-wire transmission circuit that transmits data to a remote receiving unit via a two-wire transmission line.

As is well known, a two-wire transmission circuit converts the output signal of a detector into a corresponding current and then supplies it to a two-wire transmission line, and because it only requires two transmission lines, it is widely used in the field of industrial measurement. Widely used.

This two-wire transmission circuit modulates the amplitude of a carrier wave according to the electrical change state of a detector that detects a physical quantity, detects and smoothes this signal, and obtains a DC signal corresponding to the physical quantity (for example, the U.S. patent No.
3646583, etc.), or one that measures the charging time of the capacitance using a capacitance type detector and generates an electrical signal based on this measured time (for example, U.S. Patent No. 518536). Are known.

However, in the former case, it is necessary to compensate for the temperature characteristics of the detection diode, and it is necessary to select and use diodes with uniform temperature characteristics, and to balance the input and output using a circuit network using high-precision resistors. The latter method requires a high-speed and high-precision level comparator, which makes it difficult to reduce production costs. It has some drawbacks.

Therefore, in order to solve the above-mentioned drawbacks, the present applicant developed a signal transmitter that does not use a high-precision resistor or a high-precision level comparator (Japanese Patent Laid-Open No. 56-132695,
We have developed the "signal transmitter" shown in Publication No.-132696. This signal transmitter generates first and second oscillation signals in which at least one of the frequencies changes according to a process quantity (physical quantity), and further includes a counter that alternately counts the first and second oscillation signals. , the pulse signal output from the output terminal of a predetermined bit of this counter is controlled intermittently by the pulse signal output from the output terminal of other bits and supplied to the integrating circuit, and the pulse signal output from the output terminal of the predetermined bit is controlled intermittently by the pulse signal output from the output terminal of other bits. When the pulse signal is disconnected, a return signal is supplied to the integrating circuit, and the output signal of this integrating circuit is supplied to the output section that controls the transmission current, which is useful for circuit adjustment and component selection. It has the advantage of being problem-free and easy to manufacture.

However, this signal transmitter had the following drawbacks.

Since this method divides the signal output from the output terminal of a predetermined bit of the counter and inserts a return signal (under intermittent control), a large number of switch elements are required to divide the signals, and the number of these switch elements increases. Switching timing control must be accurate, making the circuit complex. If the output signal peak value of the counter is not accurately managed, measurement errors will occur. The signal input to the integrator circuit is a signal (pulse signal) with a voltage value corresponding to the process amount, and if noise is mixed into this signal, measurement errors will occur, so this part must be shielded or the integrator circuit The distance between the sensor circuit and the sensor circuit (a circuit consisting of a detector, a counter, etc.) must be kept as short as possible, which is extremely inconvenient when it is desired to provide the sensor circuit in isolation.

In view of the above-mentioned circumstances, this invention eliminates the need for a switch element to separate the output signals of the sensor circuit.
Furthermore, it does not require accurate control of the output signal peak value of the sensor circuit, and also provides a two-wire transmission circuit that allows the sensor circuit to be installed in isolation. a transmission current control section that controls the transmission current and generates a return signal according to the transmission current; a signal conversion section that outputs a pulse signal whose duty ratio changes according to the physical quantity to be measured; The present invention is characterized by comprising a switch means that performs intermittent control using a pulse signal, and a smoothing circuit that smoothes the output of the switch means and applies the smoothed output to the input of the transmission current control section.

Embodiments of the present invention will be described below with reference to the drawings.

FIG. 1 is a block diagram showing the configuration of a first embodiment of the present invention. In this figure, reference numeral 1 denotes a single variable capacitance detector consisting of a movable electrode 1a and fixed electrodes 1b and 1c, and the movable electrode 1a is configured to move in response to an external force. Further, the capacitance C ₁ formed by the movable electrode 1a and the fixed electrode 1b and the capacitance C ₂ formed by the fixed electrodes 1b and 1c are each expressed by the following equations, assuming that the displacement of the movable electrode 1a is ΔP.

(However, k ₁ is a constant C ₀ is the capacitance of C ₁ and C ₂ when △P=O) Next, in FIG. 1, 2 is a signal conversion section,
2a and 2b are power supply terminals thereof, and 2c is an output terminal. As shown in FIG. 2, this signal conversion section 2 is comprised of switch elements SW ₁ and SW ₂ , an oscillation section 3, and a binary counter 5 of arbitrary bits. In the circuit shown in the figure, when the n-th output terminal Qn of counter 5 is at a high level, SW ₁ is ON and SW ₂ is ON.
is OFF, and when the output terminal Qn is at a low level, SW ₁ is OFF and SW ₂ is ON. Oscillator 3 outputs signal S 1 of frequency ₁ corresponding to capacitance C ₁ when switch element SW 1 is ON, and switch element SW ₂ outputs signal S ₁ of frequency 1 corresponding to capacitance C ₁ .
When ON, outputs signal S ₂ of frequency ₂ corresponding to capacitance C ₂ . According to this configuration, the counter 5 alternately counts the signals _S1 and _S2 , and the output terminal Qn becomes high level when counting the signal _S1 and becomes low level when counting the signal _S2 .

Here, the pulse Ps output from the output terminal Qn
(hereinafter referred to as the detection pulse signal) is shown in FIG.
As shown in this figure, the detection pulse signal Ps alternates between a period _T1 during which the counter 5 is counting the signal _S1 and a period _T2 during which the counter 5 is counting the signal _S2 . In this case, the ratio between periods T ₁ and T ₂ corresponds to the physical quantity (process quantity) to be measured. This detection pulse signal Ps is applied to the switch means 8 shown in FIG.
Supplied as OFF control signal, detection pulse signal
When Ps is at a high level, the switch means 8 is in the ON state,
When the level is low, the switch means 8 is turned off. On the other hand, 10 is a transistor Q ₁ , Q ₂ , a resistor R ₉ ,
R ₁₀ , diodes D ₁ , D ₂ and Zener diode
This is a constant voltage circuit that is composed of ZD and outputs a constant voltage Es. 11 is an operational amplifier, which controls the transistor _Q3 for controlling the transmission current. The emitter of this transistor _3Q is connected to the common line COM via a diode _D3 and a resistor _R11 , and a feedback resistor Rf is connected between the common line COM and the negative output terminal 21.
is inserted. Resistors R ₃ and R ₄ divide the constant voltage Es, and the resulting voltage V ₂ is applied to the operational amplifier 1.
The resistors R ₁ and R ₂ divide the voltage difference between the constant voltage Es and the voltage Ef across the return resistor Rf, and supply the voltage V ₁ to one terminal of the switch means 8. . The voltage V ₁ is supplied to a smoothing circuit 30 consisting of a resistor R ₅ and a capacitor C ₅ every time the switch means 8 is turned on, and the output voltage V ₃ of this smoothing circuit 30 is supplied to the non-inverting input terminal of the operational amplifier 11. be done. Also, in the circuit described above, the resistance R ₁
and R ₂ are set to sufficiently large values, and the current flowing through the resistors R ₁ and R ₂ is configured to be sufficiently negligible compared to the current flowing through the return resistor Rf. In FIG. 1, reference numerals 15 and 16 are a power source and a resistor provided in a remote receiving section, respectively, and these are inserted between the positive output terminal 20 and the negative output terminal 21.

Next, the operation of the above-mentioned circuit will be explained.

First, since the output voltage V ₃ of the smoothing circuit 30 is determined by the voltage V ₁ and the ON time of the switch means 8, V ₃ =V ₁・T ₁ /T ₁ +T ₂ =(Es・R ₂ /R ₁ +R ₂ −Ef・R ₂ /R ₁ +R ₂ )T ₁ /T ₁ +T ₂ …(2), and as can be seen from the figure, V ₂ becomes V ₂ =EsR ₄ /R ₃ +R ₄ …(3). Become.

Here, when the voltage V ₃ becomes larger than V ₂ , the output voltage of the operational amplifier 11 increases, and the transistor Q ₃
The current flowing through increases. As a result, the current flowing through the return resistance Rf increases and the voltage Ef increases. Voltage Ef
When V 1 becomes large, the voltage V ₁ decreases, and as a result, the output voltage V ₃ of the smoothing circuit 30 decreases. and,
The current flowing through transistor Q ₃ is controlled so that V ₃ =V ₂ . Further, even when the voltage V ₃ becomes smaller than V ₂ , the current flowing through the transistor Q ₃ is controlled so that V ₃ =V ₂ as in the case described above.

Thus, voltage V ₃ is always equal to voltage V ₂ in this circuit.

Therefore, assuming that equations (2) and (3) are equal, (EsR ₂ /R ₁ +R ₂ −EfR ₁ /R ₁ +R ₂ )−T ₁ /T ₁ +T ₂ =EsR ₄ /R ₃ +R ₄ …(4) is obtained. By sequentially transforming this equation (4), EsR ₂ /R ₁ +R ₂ −EfR ₁ /R ₁ +R ₂ = EsR ₄ /R ₃ +R ₄ ×T ₁ +T ₂ /T ₁ …(5) EsR ₂ /R ₁ +R ₂ = Es{R ₂ /R ₁ +R ₂ −R ₄ /R ₃ +R ₄・T ₁ +T ₂ /T ₁ } …(5)′ Ef＝Es{R ₂ /R ₁ −R ₄ ( R ₁ + R ₂ )/R ₁ (R ₃ + R ₄ )×T ₁ +
T ₂ /T ₁ } ...(5)'' Here, if R ₂ /R ₁ = A, R ₄ (R ₁ + R ₂ ) / R ₁ (R ₃ + R ₄ ) = B ... (6) , the above equation (5)'' becomes Ef=Es{A-B(1+ _T2 / _T1 )}=Es{(A-B) _-BT2 / _T1 }...(7). Since the current flowing through the resistors R ₁ and R ₂ can be ignored as mentioned above, the transmission current I ₀ becomes I ₀ = Ef / Rf (8), and this equation (8) is replaced by the equation (7). By substituting , _I0 =(A-B)Es/Rf-BEs/Rf-・_T2 / _T1 ...(9) is obtained. Now, the times T ₁ and T ₂ are proportional to the capacitances C ₁ and C ₂ , respectively, so if the proportionality constant is k ₂ , then T ₁ =
k ₂ C ₁ , T ₂ = k ₂ c ₂ , and by substituting the above equation (1) into this relationship, we get becomes. Here, when the movable electrode 1a (see FIG. 1) is not energized and Δp=0, T ₁ =T ₂ as seen from equation (10). By the way, in a two-wire transmitter that outputs a transmission current of 4 to 20 mA, △
Adjust the transmission current I ₀ to 4 mA when p = 0 (hereinafter referred to as 0
Therefore, the right side of the following equation I ₀ =(A-2B)/R·Es (11) obtained as T ₁ =T ₂ in equation (9) corresponds to 4 mA. And (9), (10),
The following equation representing the transmission current I ₀ can be obtained from equation (11).

I ₀ = (A-2B) Es/Rf+BEs/Rfk ₁・△p =4mA+B Es/Rfk ₁・△P …(12) In this equation (12), (B・Es・k ₁ /Rf) is a constant. Therefore, the transmission current I ₀ is ultimately proportional to the displacement ΔP of the movable electrode 1a.

Next, a case will be described in which a differential capacitance detector 1' as shown in FIG. 4 is used in place of the single variable capacitance detector 1 in this embodiment. As is well known, in the differential capacitance detector 1', the center electrode 1b is a movable electrode, and the capacitances C ₁ and C ₂ change complementarily in accordance with the displacement ΔP of the movable electrode 1b. In this case, the capacitances C ₁ and C ₂ are expressed by the following equations.

Therefore, the periods T ₁ and T ₂ are each becomes. Substituting this equation (14) into the above equation (9), I ₀ = (A-B) Es/Rf-B・Ef/Rf・1-k ₁ △P/1
+k ₁ △P = (A-B) Es/Rf-B・Ef/Rf (1-2k ₁ △P/1+k ₁ △P) = (A-2B) Es/Rf+B.Ef/Rf×2k ₁ △P /1+k ₁
△P = 4mA + B・Ef/Rf×2k ₁ △P/1+k ₁ △P …(15) As shown in this equation, the transmission current I ₀ is not a linear function with respect to ΔP, that is, I ₀ is non-linear as shown by the solid line in FIG. In this case, if a compensation resistor R _N is inserted between the inverting input terminal of the operational amplifier 11 and the common line COM shown in FIG. 1, I ₀ can be changed linearly with respect to ΔP ( (See dashed line in Figure 5).

FIG. 6 is a block diagram showing the configuration of a second embodiment of the invention. In this figure, the same reference numerals are given to the parts corresponding to those in FIG. 1, and the explanation thereof will be omitted.

In this figure, 2' is a signal converter, the configuration of which is shown in FIG. The signal converter 2' shown in FIG. 7 is the signal converter 2 shown in FIG.
The difference is that the electrode 1c is directly connected to the input terminal of the oscillator 3 without using a switch element. According to such a configuration, when the switch element SW ₁ is ON, the oscillation unit 3 outputs the signal S ₁ of frequency ₁ corresponding to (C ₁ + C ₂ ), and the switch element SW 1
If SW ₁ is OFF, frequency ₂ corresponding to capacitance C ₂
Outputs the signal _S2 . Therefore, the high level period T ₁ and the low level period T ₂ of the pulse Ps in this case
(see FIG. 3) are each expressed by the following equations.

T ₁ =k ₃ (C ₁ +C ₂ ) T ₂ =k ₃ C ₂ (16) Then, this pulse Ps is sent to the switch means 26 as an ON-OFF control signal via the resistor 25 as shown in FIG. Supplied. The switch means 26 connects the terminals 26a and 26c when the pulse Ps is at a high level, and connects the terminal 26b when the pulse Ps is at a low level.
and 26c. Next, the resistors 30 and 31 each have the same resistance value, and have a constant voltage
The difference between the voltage Vf obtained at the sliding terminal of Es and the variable resistor Rs is divided into 1/2 (note that it is not necessarily necessary to divide the voltage into 1/2), and the voltage _V5 is applied to the non-inverting input terminal of the operational amplifier 25. supply to. Therefore, the voltage V ₅
is expressed by the following equation.

V ₅ = 1/2 (Es-Vf) (17) Further, the operational amplifier 25 constitutes a voltage follower, and outputs the voltage V ₅ as it is to the output terminal. The terminal 26c of the switch means 26 is a resistor.
It is connected to the common line COM via R ₂₅ and also to the smoothing circuit 30 . In this case, the output voltage V ₃ of the smoothing circuit 30 is expressed by the following equation.

V ₃ = V ₅ · T ₁ / T ₁ + T ₂ ...(18) On the other hand, R ₀ is a variable resistor used to adjust the reference value (4 mA) of the transmission current I ₀ , and the voltage at its sliding terminal is becomes V ₀ . Resistors R ₂₀ and R ₂₁ divide the difference between the voltages V ₀ and V ₃ to the operational amplifier 11.
is supplied to the non-inverting input terminal of In the case of this embodiment, the resistances R ₂₀ = R ₂₁ = R ₃ = R ₄ are set, and the potentials of the non-inverting and inverting input terminals of the operational amplifier 11 V ₇ ,
V ₂ is expressed by the following formulas.

V ₇ = 1/2・(V ₀ −V ₃ )+V ₃ …(19) V ₂ =1/2・Es …(20) A constant voltage Es is supplied to one end of the resistor R ₂₇ , and this resistor R ₂₇ The other end is connected to the negative output terminal 21 via a resistor _R28 . The resistance value of each of these resistors R ₂₇ and R ₂₈ is such that when the transmission current I ₀ is adjusted to a reference value of 4 mA, the voltage Ef across the return resistor Rf is equal to the voltage E ₁ across the resistor R ₂₈ . It is set so that Therefore, when the transmission current I ₀ is 4 mA, the voltage difference between both fixed terminals of the variable resistor Rs becomes 0, and no current flows through the variable resistor Rg. Therefore, when the transmission current I ₀ is 4 mA, the position of the sliding terminal of the variable resistor Rs has no effect on the transmission current I ₀ . On the other hand, when the transmission current exceeds 4 mA, the voltages Ef and E ₁ become different, so a current corresponding to the transmission current I ₀ flows through the variable resistor Rs. As a result, the voltage Vf becomes the transmission current I ₀
The value corresponding to Therefore, if the voltage E ₁ when I ₀ =4 mA is E ₁₀ , the transmission current I ₀ in this example is expressed by the following equation.

I ₀ = 4mA + Vf / α・Rf = E ₁₀ /Rf + Vf / α・Rf …(21) However, α is a constant determined by the position of the sliding terminal of variable resistor Rs. When changing the position, the value of voltage Vf changes and at the same time the transmission current I ₀
The ratio of voltage Vf to voltage Vf changes. That is, by adjusting the variable resistor Rs, the ratio of the voltage Vf, which is the return amount, can be changed, and the value of the constant α in equation (21) can be changed (hereinafter referred to as span adjustment). In this embodiment, a diode _D5 is inserted between the positive output terminal 20 and the collector of the transistor _Q3 in the forward direction.

Next, the operation of this embodiment will be explained.

In the circuit shown in Figure 6, when V ₇ > V ₂ ,
The output voltage of operational amplifier 11 increases, the current flowing through transistor Q ₃ increases, and the transmission current I ₀ increases. As a result, the voltage Vf increases, the voltage _V5 decreases (see equation (17)), and the voltage _V3 decreases. As a result,
The voltage V ₇ decreases, and the value of the transmission current I ₀ is controlled so that V ₇ =V ₂ . In this way, the operational amplifier 11 operates so that the voltage difference between both input terminals is 0, as in the above-described embodiment, and the voltage V ₂ is always maintained.
= _V7 . Therefore, when formula (19) = formula (20) and transform the formula, 1/2・(V ₀ −V ₃ )+V ₃ =1/2・Es …(22) V ₀ +V ₃ =Es …( 22′). Then, from equations (17), (18), and (22) mentioned above,
When Vf is calculated by eliminating V ₃ and V ₅ , Vf=Es−(Es−V ₀ )・2(T ₁ +T ₂ )/T ₁ …(23), and this equation (23) is replaced by (16) Substituting the formula and eliminating T ₁ and T ₂ , Vf=Es−2(Es−V ₀ )k ₃ (C ₁ +2C ₂ )/k ₃ (C ₁ +C ₂ )
…(24) becomes. Next, by substituting equation (13) into equation (24), △
Finding the relationship between P and Vf, Vf=Es-2(Es-V ₀ )(C ₀ 1/1-k ₁ △P+2・C ₀ 1
/1+k ₁ △P/(C ₀ 1/1-k ₁ △P+C ₀ 1/1+k ₁ △P
)=Es−2(Es−V ₀ )(3−k ₁ −△P)/2 =Es−(3Es−Esk ₁ △P−3V ₀ +V ₀ k ₁ △P)=3V ₀ −2
Es+(Es−V ₀ )k ₁ △P…(25) Here, since the voltage V ₀ can be arbitrarily determined by changing the position of the sliding terminal of the variable resistor R ₀ , V ₀ =2/3Es… (26), equation (25) becomes Vf=(Es−V ₀ )・k ₁・△P (27). Substituting equations (26) and (27) into equation (21), I ₀ = E ₁₀ /Rf + (Es-V ₀ )/α・Rf・R ₁・△P = E ₁₀ /Rf+Es/3α・Rf・R ₁・△P …(28) As can be seen from this equation (28), the transmission current I ₀
is proportional to △P.

In this example, if capacitances C ₁ and C ₂ are exchanged, that is, electrode 1 in FIG.
a directly to the input terminal of the oscillator 3, and the electrode 1c
When connected to the input terminal of the oscillation section 3 via the switch element _SW1 , the circuit configuration is as shown in FIG. 8, for example. In this figure, the operational amplifier 25 constitutes an inverting amplifier with a gain of 1 with resistors _R32 to _R35 ,
Further, the resistor R ₃₂ =R ₃₃ is set, and Es/2 (V) is supplied to the non-inverting input terminal of the operational amplifier 25. That is, the configuration is such that the potential of the output terminal of the operational amplifier 25 becomes (Es-V ₀ ). On the other hand, the voltage V ₇ is supplied to the inverting input terminal of the operational amplifier 11, and the voltage V ₂ is supplied to the non-inverting input terminal. Note that the other parts are similar to the circuit shown in FIG.

In this way, if the capacitances C ₁ and C ₂ are swapped, the period
T ₁ and T ₂ (see Figure 3) are each T ₁ = k ₃ (C ₁ + C ₂ ) T ₂ = k ₃ C ₁ (29), and the voltage V ₃ is V ₃ as shown in the figure. = (Es−V ₅ )×T ₁ /T ₁ +T ₂ (30). Next, △P in the circuit shown in this figure and
Derive the relationship with I ₀ . In the circuit shown in this figure, only the periods T ₁ and T ₂ are changed as shown in equation (29), and other operations are the same as in the previous embodiment.
From equations (17), (22)′, and (30), Vf=Es−2V ₀ +2(Es−V ₀ )T ₂ /T ₁ (31). By substituting the above equation (29) into this equation (31), we get Vf=Es−2V ₀ +2(Es−V ₀ )k ₃ C ₁ /k ₃ (C ₁ +C ₂ ) …(32), and this ( Substituting the above equation (13) into equation 32), Vf=Es−2V ₀ +2(Es−V ₀ )C ₀・1/1−k
₁ △P/C ₀・(1/1−k ₁ △P+1/1+k ₁ △P) = Es−2V ₀ +2(Es−V ₀ )・1/2(1
+k ₁ △P) =2Es−3V ₀ +(Es−V ₀ )k ₁ △P…(33)
becomes. Here, when V ₀ =2/3Es, Vf=1/3Esk ₁ ΔP (34), and by substituting this equation (34) into the above equation (21), the following equation is obtained.

I ₀ = E ₁₀ /Rf+Es/3α・Rf・R ₁・△P (35) From this equation (35), it can be seen that the transmission current I ₀ is proportional to △P.

Here, if we transform the above-mentioned equation (23), we get Vf=Es-(Es-V ₀ )・2・(1+T ₂ /T ₁ )...(36), and in this equation (36), Es and V ₀ are Since it is a constant, it can be seen that the larger the change in period T ₂ , the larger the change in voltage Vf, which is the return signal component, which is advantageous in signal processing. Further, the same can be said of the above equation (31).

Therefore, when increasing the amount of change in the voltage Vf, a countdown circuit 40 shown in FIG. 9 is installed between the output terminal of the oscillation section 3 in the signal conversion section 2' shown in FIG. Interpose. In this case, the input terminal 40a is connected to the output terminal of the oscillator 3, the output terminal 40b is connected to the input terminal CL of the counter 5, and the control terminal 40c is connected to the output terminal Qn of the counter 5. As shown in FIG. 9, this countdown circuit 40 is composed of NAND gates 41 to 44 and a T-type flip-flop _FF1 , and has a control terminal 4.
When 0c is at a high level (during period _T1 ), the pulse signal supplied to the input terminal 40a is outputted to the output terminal 40b via the NAND gates 43 and 42 sequentially, and when the control terminal 40c is at a low level ( During period _T2 ), the pulse signal supplied to the input terminal 40a is frequency-divided by 1/2 by FF ₁ , and then outputted to the output terminal 40b via the NAND gates 41 and 42. According to such a configuration, the time width of the period T ₂ is doubled, and as can be seen from the above equation (36), the amount of change in the voltage Vf can be increased, which is extremely advantageous in terms of signal processing. In addition, in the circuit described above, FF ₁ is set to 1
Although only one is used, if two, three, etc. are connected in series, the time width of the period T ₂ can be increased by four times, eight times, etc.

Here, an example of a specific circuit of the signal converter 2' is shown in FIG. In this figure, 51 to 54 are NAND gates, 56 is a constant current source, 57 is a low-pass filter, 58 is an operational amplifier, and C ₁₀ and C ₁₁ are capacitors. In the circuit shown in this figure, if the time width of period T ₂ is not to be increased, the points shown in the figure are directly connected, and if the time width of period T ₂ is to be doubled, the input terminal 40a shown in FIG. , the output terminal 40b and the control terminal 40c, respectively.
Connect points. Then, the time width of period T ₂ is set to 2
Waveforms at points, points, and points in the case of doubling are shown in FIGS. 11A to 11C, respectively. Note that the waveform at the point is the detection pulse signal Ps.

Note that an integrating circuit 60 shown in FIG. 12 may be used in place of the smoothing circuit 30 used in the circuits shown in FIGS. 6 and 8. This integrating circuit 60 is composed of a resistor R ₄₀ , a capacitor C ₁₅ , a bias power supply Va, and an operational amplifier 59, as shown in the figure.

In addition, in place of the variable resistor _R0 for O adjustment used in the circuit shown in FIG. 6, a variable resistor is used to appropriately divide and supply the voltage Es to the inverting input terminal of the operational amplifier 11, as shown in FIG. R ₀ ′ may also be provided. According to such a configuration, since the inverting input terminal voltage of the operational amplifier 11 can be changed by changing the position of the sliding terminal of the variable resistor R ₀ ', the reference value of the transmission current I ₀ can be adjusted. Further, a variable resistor Rs' shown in FIG. 13 may be used instead of the variable resistor Rs for span adjustment shown in FIG. 6. One fixed terminal of this variable resistor Rs' is connected to the output terminal of the operational amplifier 25, and the other fixed terminal is connected to the common line.
Connected to COM. According to such a configuration, by adjusting the position of the sliding terminal of the variable resistor Rs', the voltage V ₅ and the output voltage of the operational amplifier 25 can be adjusted.
The proportionality constant of V _OUT can be changed, and the transmission current I ₀
It is possible to change the ratio of the return amount to the ring, that is, the span.

FIG. 14 is a block diagram showing the configuration of the main parts of a third embodiment of the present invention. Note that the other parts of this embodiment are the same as those of the second embodiment described above (see Fig. 6).
It is similar to

In this figure, reference numeral 65 denotes a switch means having the same structure as the switch means 26, and the voltage
_V2 and the common potential are alternately switched and supplied to a smoothing circuit 70 consisting of a resistor _R40 and a capacitor _C40 . This switch means 65 has a detection pulse signal.
Ps (see FIG. 3) is inverted and supplied by an inverter 66. According to such a configuration,
The output voltage V ₃ of the smoothing circuit 30 is V ₃ = (Es - Vf) × T ₁ /T ₁ + T ₂ (37), and the output voltage V' ₂ of the smoothing circuit 70 is V' ₂ = V ₂ × T ₂ /T ₁ +T ₂ =Es/2×T ₂ /T ₁ +T ₂ (38). Then, the operational amplifier 11 outputs a voltage V′ ₂ =V ₃
Since it operates as follows, we obtain the following equation by setting equation (37) = equation (38).

(Es−Vf)×T ₁ /T ₁ +T ₂ = Es/2×T ₂ /T ₁ +T ₂ …(39) Transforming this equation (39), Vf=Es・(1−T ₂ /2・T ₁ ) ...(40) Substituting the above equations (13) and (16) into equation (40), Vf=Es・(1−1−k ₁ △P/4) =Es・(3 /4+k ₁ △P/4) =Es/4(3+k ₁ △P) ...(41). From this equation (41), it can be seen that the voltage Vf corresponds to ΔP (it is a linear function of ΔP). That is, the transmission current I ₀ corresponds to ΔP.

FIG. 15 is a block diagram showing the configuration of a fourth embodiment of the present invention.

In this figure, 70 is a current transformer, and a resistor R ₄₅ is inserted at both ends of the secondary winding. One end of this resistor _R45 is connected to the common line COM, and the other end is connected to one end of _{the resistor R5 via} a diode D5. The center tap of the primary winding of the current transformer 70 is the common line.
COM, one end is connected to the negative side output terminal ₂₁ via the switch element SW5, and the other end is connected to the negative side output terminal ₂₁ via the switch element SW6. switch element
SW ₅ and SW ₆ are each controlled by the detection pulse signal Ps so that when one is ON, the other is OFF. Further, 69 is a constant current source.

According to the configuration described above, the current transformer 70
Switch element SW ₅ provided in the primary side circuit of
By complementarily and alternately turning ON and OFF SW ₆ , a positive and negative pulse-like signal whose peak value corresponds to the duty ratio of the transmission current _I0 and the detection pulse signal Ps is obtained in the secondary side circuit. And the diode
Only positive signals are transmitted to the smoothing circuit 30 by D ₅ . As a result, the output voltage V ₃ of the smoothing circuit 30 is determined by the duty ratio of the transmission current I ₀ and the sensor pulse Ps, and the operational amplifier 11 is configured to transmit the voltage so that V ₅ =V ₃ as in each of the embodiments described above. Since the current I ₀ is controlled, the transmission current I ₀ corresponds to the duty ratio of the sensor pulse Ps. That is, the transmission current I ₀ corresponds to the aforementioned displacement ΔP.

FIG. 16 is a block diagram showing the configuration of the main parts of a fifth embodiment of the present invention.

In this figure, SW ₇ is a switch element that is ON-OFF controlled by the detection pulse signal Ps.
It is inserted between resistors R ₃₀ and R ₃₁ . Also,
Capacitor C ₂₀ is connected in parallel with resistor R ₃₀ , resistor
A capacitor _C21 is inserted between one end of _R31 (switch element _SW7 side) and the common line COM. According to such a configuration, switch element _SW7
Since the voltages Vf and Es are added in the ON state, the voltage applied to the non-inverting input terminal of the operational amplifier 11 is (Vf + Es) × - T ₁ /T ₁ + T ₂ , and the operational amplifier 11 is Since it operates so as to make the voltage difference between both input terminals zero, the transmission current I ₀ corresponds to the duty ratio of the detection pulse signal Ps.

In addition, the sensor pulse in each of the above-mentioned examples
Since Ps is a switching signal for switching the switch means 8, 26 or the switch elements _SW5 to _SW7 , even if noise or the like is mixed in and the peak value changes somewhat, it does not affect the measurement results. Therefore, the signal converters 2, 2' can be provided separately, and thereby a plurality of signal converters 2, 2' can be switched and used as shown in FIG. 17.

In this diagram, la, la, ... and lb, lb...
are power supply lines for the signal converters 2, 2, . . . , and lc, lc, . . . are signal lines to which sensor pulses Ps are output. Reference numeral 81 denotes an output stage circuit comprising the operational amplifier 11, smoothing circuit 30, switch means 8, 26, feedback circuit, etc. shown in each of the above embodiments, lc' is the input line of the sensor pulse Ps, la', lb ' are each signal converter 2
This is the power output line that supplies power to the 80 is a switching unit which selects one of the signal converters 2, 2... and connects the power supply lines la and lb of the selected signal converter 2 to the power output line of the output stage circuit 81.
la' and lb' respectively, and also connects the signal line lc to the input line lc'.

As explained above, according to the present invention, there is provided a transmission current control unit that controls the transmission current based on the signal supplied to the input and generates a return signal according to the transmission current, and a signal conversion unit that outputs a pulse signal with a varying duty ratio; a switch unit that intermittently controls the return signal using the pulse signal; the output of the switch unit is smoothed, and the smoothed output is input to the transmission current control unit; Even if the peak value of the detected pulse signal fluctuates somewhat, extremely high measurement accuracy can be maintained. Further, since there is no need to accurately control the peak value of the detection pulse signal and it is less susceptible to noise, the detector and the signal converter can be provided in isolation. In addition, since the detection pulse signal is used as a simple switching element, a circuit that separates this detection pulse signal,
There is no need for any switch elements, and the advantage is that the circuit configuration can be simplified.

[Brief explanation of drawings]

FIG. 1 is a block diagram showing the configuration of a first embodiment of the present invention, and FIG. 2 is a signal conversion section 2 shown in FIG.
Fig. 3 is a waveform diagram showing the waveform of the detection pulse signal Ps, Fig. 4 is a schematic block diagram showing the structure of the differential capacitance detector 1', and Fig. 5 shows the differences in the same embodiment. A diagram showing the relationship between displacement ΔP and transmission current _I0 when using the dynamic capacitance detector 1', FIG. 6 is a block diagram showing the configuration of the second embodiment of the present invention, and FIG. Figure 6 is a block diagram showing the configuration of the signal converter 2', Figure 8 is a block diagram showing a modified example of the circuit when capacitors _C1 and _C2 are swapped in the same embodiment, and Figure 9 is a countdown diagram. circuit 4
10 is a block diagram showing a specific configuration example of the signal converter 2', and FIG. 11 A, B, and C are points of the circuit shown in FIG. 10, respectively.
12 is a waveform diagram showing waveforms at points. FIG. 12 is a block diagram showing the configuration of an integrating circuit 60 that can be used in place of the smoothing circuit 30 shown in FIGS. 1, 6, and 8. 14 is a block diagram showing another embodiment of the O adjustment method and span adjustment method used in the circuit shown in FIG. 6, FIG. 14 is a block diagram showing the configuration of a third embodiment of the present invention, and FIG. 16 is a block diagram showing the configuration of the fourth embodiment of this invention, FIG. 17 is a block diagram showing the configuration of the fifth embodiment of this invention, and FIG. 17 is a block diagram showing the configuration of the fourth embodiment of this invention. Or 2'
FIG. 3 is a block diagram showing an example of a configuration in which a plurality of devices are provided. 1...Single variable capacitance detector (detector), 1'...
... Differential capacitance detector (detector), 2, 2' ... Signal converter, 8, 26 ... Switch means, 11 ... Operational amplifier (transmission current control section), 30 ... Smoothing circuit,
40...Countdown circuit, 60...Integrator circuit (smoothing circuit), 70...Count transformer (return voltage generation section), _SW5 , _SW6 , _SW7 ...Switch element (switch means), _C1 , C ₂ ...Capacitance (impedance element pair), Rf, F ₄₅ ...Resistance (return voltage generation section), Rs...Variable resistance (return voltage generation section).

Claims (1)

[Claims] 1. A transmission current control section that controls a transmission current based on a signal supplied to an input and generates a feedback signal according to the transmission current, and a duty ratio that changes according to a physical quantity to be measured. a signal converter that outputs a pulse signal; a switch that controls the feedback signal intermittently using the pulse signal; and a smoothing circuit that smoothes the output of the switch and provides the smoothed output to the input of the transmission current controller. A two-wire transmission circuit characterized by: