JPH0510035B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION [Field of Industrial Application] The present invention relates to a control device for an AC motor driven by a thyristor power converter.

[Conventional technology]

FIG. 7 is a configuration diagram showing an example of a thyristor motor for driving a conventional synchronous motor described in Japanese Patent Publication No. 59-1077.

In Fig. 7, 1 is a first converter that converts AC from a commercial AC power supply into DC, 2 is a second converter that converts the DC into variable frequency AC, 3 is a synchronous motor, and F is its synchronous motor. Field winding, 4 is synchronous motor 3
A position detector 5 outputs a position signal with a phase corresponding to the rotational angular position of the rotating shaft of the second transducer 2. a γ control circuit that controls the control advance angle γ; 6 a gate output circuit that outputs a gate signal for the second converter 2 based on the output signal of the γ control circuit 5;
7 is a speed generator, 8 is a speed command circuit, 9 is a speed deviation amplifier that matches and amplifies the speed command signal of the speed command circuit 8 and the speed feedback signal which is the output signal of the speed generator 7, and 10 is a first converter. 1 is a current detector that detects the AC input current; 11 is a current deviation amplifier that matches and amplifies the output signal of the speed deviation amplifier 9 and the current feedback signal of the current detector 10; 12 is a current deviation amplifier that detects the output signal of the current deviation amplifier 11; 1 converter 1
gate pulse phaser for controlling the firing phase of 13;
14 is a field command circuit that outputs a command signal Ifp that commands the magnitude of the field current If, and 14 is a thyristor circuit 1.
7, a current detector that detects the magnitude of the AC input current; 15, a current deviation amplifier that matches and amplifies the field command signal Ifp and the output signal of the current detector 14;
16 is a gate pulse phase shifter that controls the firing phase of the thyristor circuit 17, and 17 is a thyristor circuit that supplies field current If to the field winding F.

Next, to explain its operation, part numbers 7 to 12
is a speed control circuit that controls the input current of the first converter 1 according to the speed deviation, that is, the magnitude of the armature current of the motor 3 that is in a proportional relationship with this, and part numbers 4 to 6 are current detectors 10. control angle γ of the second converter 2 depending on the output signal, i.e. the armature current
The circuit that controls the circuit, part numbers 13 to 17 are field currents.
A field control circuit is configured to cause If to flow in proportion to the field command signal Ifp. Since these operations are similar to those of the already well-known so-called thyristor motor device, detailed explanation will be omitted.

FIG. 8 is a vector diagram showing the relationship between voltage and current of the motor in FIG. 7. The figure a shows the no-load condition, the figure b shows the load condition when the field current If is kept constant and the control angle γ is controlled so that the power factor is constant, and the figure c shows the field current If separately. The armature current Ia
This is a vector diagram when the vehicle is controlled so that it is proportional to , and γ is kept constant.

As is clear from Figure 8b, even if the power factor can be maintained at a predetermined value, the terminal voltage V
As Ia increases (from Ia ₁ to Ia ₂ ), it decreases (from V ₁ to
_V2 ). Due to this voltage drop, the maximum current value that can be commutated in the second converter 2 is reduced. As a result, sufficient output cannot be obtained from the electric motor 3.

In addition, in the case of c in the same figure, _the terminal voltage V increases ₍ V ₁
V ₂ ), so there is no problem like that shown in b in the same figure.

However, at the time of overload, the terminal voltage V becomes higher than at the rated time, so the thyristor of the second converter 2 needs to have a high withstand voltage. Furthermore, because the electric motor itself undergoes magnetic saturation, it may not be possible to obtain as large an output as expected. Furthermore, when the load is light, the terminal voltage V decreases, resulting in a disadvantage that the power factor (power supply power factor) of the first converter 1 decreases accordingly.

In addition, as a means of solving the above problems,
-No. 1077 describes the magnitude of the no-load induced voltage E ₀ obtained by vectorial addition of the terminal voltage and synchronous reactance drop, and the control of the phase difference between this no-load induced voltage E ₀ and the armature current Ia. A method for controlling the terminal voltage to be constant regardless of the armature current is described in detail.

Although FIG. 9 is a vector diagram showing the principle of this operation, this operation will be briefly explained here. In order to keep the terminal voltage V _M constant, the magnitude of the no-load induced voltage E ₀ and the E ₀ are determined according to the magnitude of the armature current Ia.
The phase difference θ (phase difference angle) between the armature current Ia and the terminal voltage is controlled, and the phase (γ + θ) of the second converter is controlled while maintaining the relationship of γ + θ so that the phase difference γ between the armature current Ia and the terminal voltage is constant. is under control.

However, in this method, since the terminal voltage is controlled to be constant, the commutation overlap angle u of the second converter changes depending on the magnitude of the armature current, and the arm element of the second converter changes. The application period (γ-u) of the reverse voltage to the thyristor changes.

At this time, the second converter is multiphase (for example, 12
phase) to reduce torque pulsation and drive a large-capacity thyristor motor, in order to perform commutation every 30 degrees, the arm element is The reverse voltage period of a certain thyristor is 30°-u even if γ > 30°, and in order to achieve stable commutation of the second converter, this commutation overlap angle must be adjusted to increase the armature current. It is necessary to set the terminal voltage so that it does not become extremely large.

In addition, in order to control this voltage accurately,
The magnetic saturation characteristics of the AC motor 3 must be taken into consideration, and the method disclosed in Japanese Patent Publication No. 59-1077 has a problem in accuracy.

[Problem that the invention seeks to solve]

Conventional AC motor control devices are configured as described above, so the terminal voltage and power factor fluctuate significantly due to load fluctuations, causing the commutation of the second converter to become unstable or insufficient output. There were problems such as not being able to obtain

This invention was made to solve the above-mentioned problems, and it is an alternating current that can prevent fluctuations in terminal voltage and power factor due to load fluctuations, perform stable commutation, and obtain sufficient output. The purpose is to obtain a control device for an electric motor.

[Means for solving problems]

The AC motor control device according to the present invention controls the phase difference θ (phase difference angle) between the terminal voltage and the no-load induced voltage and the field current according to the armature current, and
The motor is equipped with a vector calculator that controls the magnitude of the terminal voltage to ensure a predetermined commutation margin angle, and the terminal voltage signal of the AC motor is used as the phase signal of the AC motor.

[Effect]

The AC motor control device in this invention controls the locus of the terminal voltage by a vector calculator so that it is parallel to the axis (d-axis) of the field current, and the field current is a magnetizing current for generating the terminal voltage. The terminal voltage of the AC motor is controlled by the sum of the d-axis component of Let this be the phase signal of the AC motor.

〔Example〕

An embodiment of the present invention will be described below with reference to the drawings. In FIG. 1, 18 is a power factor angle command circuit that commands the advance angle φ (power factor angle) of the armature current with respect to the terminal voltage of the motor 3, and 19 is a no-load command circuit that commands the terminal voltage of the motor 3 when no load is applied. terminal voltage command circuit,
Reference numeral 20 denotes a vector calculator, into which the commands of the power factor angle command circuit 18 and the no-load terminal voltage command circuit 19 and the armature current detection signal Ia are input, and the field current command Ifp and the second converter 2 are inputted. Outputs the phase command β. 21 is a phase control circuit, and PT serves as a terminal voltage detector for detecting the terminal voltage of the motor 3.
The conduction phase angle of the second power converter 2 is controlled based on the output signal of the power converter 22 and the command of the vector calculator 20.

FIG. 2 shows a detailed configuration diagram of the vector calculator 20. As shown in FIG. In Fig. 2, 201 is a θ function table that outputs a signal θ (phase difference angle) using V ₀ , Ia, and φ;
202 is the output of this θ function table 201 and V ₀
V calculation circuit 203 that calculates the terminal voltage V by
is the magnetizing current from the output signal of this V calculation circuit 202.
A no-load saturation curve table 204 of the motor 3 for calculating iμ is this no-load saturation curve table 203
205 is an iμd calculation circuit that outputs iμd from the output signal of θ and θ; 205 is an Eaq calculation circuit that calculates the q-axis armature reaction voltage component Eaq from Ia and φ;
an ifa calculation circuit that calculates a compensation field current component ifa of armature reaction from the output signal of the Eaq calculation circuit 205;
207 is an adder as a field current command generating circuit that adds the output signals of this ifa calculation circuit 206 and the iμd calculation circuit 204, 208 is a u calculation circuit that calculates the commutation overlap angle u from V and φ, and 209 is a u calculation circuit. This adder serves as a phase command generation circuit that adds the output signal u/2 of the u calculation circuit 208 and φ.

Next, the principle of operation of the above embodiment will be explained with reference to the vector diagram shown in FIG. Assuming that the direction of the field current is the d-axis and the axial direction perpendicular thereto is the q-axis as a reference axis, a no-load induced voltage of the motor 3 is generated in the Q-axis direction.

The basis of the control means in this invention is that the vector locus of the terminal voltage V, with respect to the no-load terminal voltage V ₀ on the q-axis, is
This is to control the movement so that it changes parallel to the d-axis direction. If the phase difference (phase difference angle) between the terminal voltage V and the q-axis is θ, and the phase difference (power factor angle) between the armature current Ia and the terminal voltage V is φ, then the terminal voltage V is the no-load terminal voltage
V ₀ and armature reaction voltage component Ead generated in the d-axis direction
= XaqIacos (φ+θ), and the following relationship holds true.

V ₀ tanθ=XaqIacos(φ+θ)...(1) Transform equation (1) to obtain equation (2).

XaqIa/V ₀ = tan θ/COS (φ + θ) ...(2) Here, the left side of equation (2) is the ratio of the d-axis armature reaction voltage component to the no-load terminal voltage with respect to the no-load terminal voltage V ₀ [per unit ( Perunit) value]. The θ function table 201 is a table that uses the power factor angle φ as a parameter and calculates the phase difference angle θ from tanθ/cos (φ+θ). By inputting the Perunit value on the left side of equation (2), θ for a given φ can be calculated. can be found.

FIG. 4 is a graph showing an example of this θ function table.

In other words, if the trajectory of the terminal voltage V is controlled to be parallel to the d-axis, the no-load terminal voltage of the d-axis armature reaction voltage component due to the q-axis armature reaction
The relationship between the ratio with V ₀ , phase difference angle θ, and power factor angle φ is shown in (2) above.
It becomes like the expression.

Here, in Figure 4, the horizontal axis is the phase difference angle θ, the vertical axis is the value of the right side of equation (2), and the vertical axis is when the phase difference angle θ is changed from 0° to 50° using the power factor angle θ as a parameter. Calculate the value of and store it in ROM as a table value.

During control, the left side of equation (2) is calculated, and the power factor angle φ is a set value, so for example, Xaq−
When Ia/V ₀ =3.0 and the power factor angle φ is controlled to be constant at 40°, the necessary θ is derived to be 36° by referring to the table in the order of the dotted line arrows in FIG.
Regarding the parameter θ, for example, θ=30° to 40°
It has been turned into a flat table.

The terminal voltage V is determined as a function of θ using the following equation.

V=V ₀ /cosθ (3) The V calculation circuit 202 calculates the terminal voltage V according to equation (3). Next, the magnetizing current iμ generated in the direction orthogonal to this terminal voltage signal V is determined using the no-load saturation curve table 203. This no-load saturation curve table shows the relationship between the induced voltage and the field current at a predetermined speed, taking into account the magnetic saturation of the motor 3, as shown in the graph of FIG. 5 with curve 1 as an example. This magnetizing current iμ corresponds to the composite magnetomotive force of the electric motor 3.

The d-axis component iμd of this magnetizing current iμ is calculated according to the following relational expression, and the iμd calculation circuit 204 executes the calculation of equation (4).

iμd=iμcosθ (4) On the other hand, the armature reaction voltage component Eaq in the q-axis direction is given by the following relational expression, and is calculated in the Eaq calculation circuit 205.

Eaq=XadIasin(φ+θ) (5) This q-axis armature reaction voltage component Eaq is controlled to be compensated by the field current component ifa in the d-axis direction. The conversion from Eaq to ifa in this case is executed by the ifa calculation circuit 206, as shown in the following equation, using the tangent Kfa of the no-load saturation curve shown in FIG. 5 as a coefficient.

ifa=Kfa・Eaq (6) The d-axis field current components iμd, ifa obtained according to the above equations (4) and (6) are added by the adder 207, and the field is calculated as shown in the following equation. The magnetic current command Ifp is obtained.

Ifp=iμd+ifa (7) The firing phase command γ of the second converter 2 is determined by the power factor angle φ with respect to the phase of the terminal voltage V according to the following relational expression.
and the commutation overlap angle u.

γ=φ+u/2 (8) At this time, the firing phase angle β of the second transducer 2 with respect to the q-axis direction is as follows.

β=θ+φ+u/2…(9) Here, the commutation overlap angle u is expressed by the following equation.

Note that equation (10) can be obtained by eliminating γ from cos(γ−u)−COSγ=√2XcId/V and equation (9).
Furthermore, since equation (10) is a function of the DC current Id of the second converter 2, it is necessary to convert this Id into the fundamental wave effective value Ia of the armature current. If the commutation overlap angle u is considered, the armature current will have a trapezoidal waveform as shown in FIG. 6, and the basic effective value Ia of the armature current at this time will be a function of u as follows.

Ia=√6/π sinu/2/u/2Id...(11) However, in large-capacity thyristor motors with 12 phases or more, the commutation overlap angle u is generally u20° to 25°.
If the voltage is not limited, it will not be possible to secure the reverse voltage period for turning off the thyristor. In this case, sinu/2/u/2 in equation (11) is 1 to 0.992, and it is practically acceptable to set Ia≈√6/πId.

Therefore, if we transform equation (10), we get The u calculation circuit 208 executes the calculation according to equation (12).

As described above, since the device of the present invention is controlled according to the vector relational expressions (1) to (3), in order to secure the thyristor commutation margin angle (reverse voltage application period) of 30°-u, Above power factor angle φ and no-load terminal voltage V ₀
It is only necessary to select an appropriate value.

The phase control circuit 21 uses the phase signal of the terminal voltage of the motor 3 detected by the PT 22 as a reference.
It is sufficient to perform a phase operation to advance the phase by the phase command γ. Various types of this phase control method have been put into practical use, and since this is a well-known technique, a description thereof will be omitted here.

In the above embodiment, the constants Xad, Kaq, and Xc mean the d-axis armature reaction reactance, the q-axis armature reaction reactance, and the commutation reactance, respectively, and these constants are proportional to the frequency of the motor 3. Although this is omitted for convenience of explanation, it may be changed in accordance with the output signal of the speed generator 7. Similarly, when calculating the magnetizing current iμ using the no-load saturation curve table 203, the input signal, the terminal voltage V, is converted into a signal that is inversely proportional to the speed of the motor 3. Good too.

Further, in the above embodiment, the detection signal of the armature current Ia is used as the input signal of the vector calculator 20, but the output signal of the speed deviation amplifier 9 may also be used. If the response characteristics of the current deviation amplifier are improved so that the deviation between the detection signal of the armature current Ia and the reference signal of the armature current, which is the output signal of the speed deviation amplifier 9, is reduced, the same effect as in the above embodiment can be obtained. play.

Further, in the above embodiment, the vector calculator 20
The calculation may be digitally processed by a microcomputer or the like, and in this case, the calculation accuracy is improved compared to analog processing. In addition, in the above embodiment, a 6-phase rectifier circuit is shown as the second converter 2 in FIG. Even if the structure is configured as shown in FIG.

〔Effect of the invention〕

As described above, according to the present invention, a table of the phase difference angle θ is used such that the vector locus of the terminal voltage changes parallel to the d-axis direction with respect to the no-load terminal voltage, and By performing vector calculations and using a no-load saturation curve to calculate the magnetizing current, the accuracy of the device can be improved.
Further, there is an effect that a device capable of stable commutation operation can be obtained.

[Brief explanation of the drawing]

Fig. 1 is a block diagram showing a control device for an AC motor according to an embodiment of the present invention, Fig. 2 is a detailed block diagram of the vector calculator in Fig. 1, and Fig. 3 is a diagram for explaining the operating principle of the present invention. vector illustration, 4th
The figure is a characteristic diagram of the θ calculation circuit, Figure 5 is a characteristic diagram showing the no-load saturation curve, Figure 6 is a waveform diagram of the armature current,
Fig. 7 is a configuration diagram of the conventional device, Fig. 8 is a vector diagram showing the relationship between voltage and current of the motor, and Fig. 9 is a diagram of the configuration of the conventional device.
A vector diagram for explaining the operation of the device shown in the figure, and FIG. 10 is a voltage waveform diagram of a thyristor. 1 is a first power converter, 2 is a second power converter, 3 is an AC motor (synchronous motor), 4 is a position detector, 18 is a power factor angle command circuit, 19 is a no-load terminal voltage command circuit, 20 is a vector calculator, 22 is a terminal voltage detector (PT), 201 is a phase difference angle calculation table, 202 is a terminal voltage calculator, 203 is a no-load saturation curve table, 204 is a d-axis component magnetizing current calculator, 205 is a q-axis armature reaction voltage calculator, 20
6 is a field current calculator, 207 is a field current command generation circuit (adder), 208 is a commutation overlap angle calculator,
209 is a phase command generation circuit (adder). In addition, in the figures, the same reference numerals indicate the same or equivalent parts.

Claims (1)

[Claims] 1. First and second power converters that convert alternating current to direct current and convert the direct current to alternating current, an alternating current motor driven by the output of the second power converter, and the phase of the terminal voltage of the alternating current motor. a terminal voltage detector for detecting the voltage at the no-load terminal; a no-load terminal voltage command circuit for setting the magnitude of the no-load terminal voltage of the AC motor; a power factor command circuit for commanding the power factor angle of the AC motor; Vector calculation for outputting a field current command of the AC motor and a firing phase command of the second power converter according to the magnitude of the armature current of the AC motor based on the voltage command signal and the power factor angle command signal. and a phase control circuit which inputs the terminal voltage of the AC motor and the firing phase command, advances the firing phase command based on the phase signal of the terminal voltage, and supplies it to the gate output circuit, The vector calculator is configured to operate the d-axis armature in order to perform vector calculation such that the vector locus of the terminal voltage of the AC motor changes in a direction perpendicular to the no-load terminal voltage in accordance with the magnitude of the armature current. a phase difference angle calculation table that calculates a phase difference angle by inputting the ratio of the reaction voltage to the no-load terminal voltage; a terminal voltage calculator that calculates the terminal voltage from the phase difference angle and the no-load terminal voltage signal; a no-load saturation curve table of the AC motor for calculating the magnetizing current; a d-axis component magnetizing current calculator for calculating the d-axis component of the magnetizing current from the phase difference angle; a q-axis armature reaction voltage calculator for calculating a shaft armature reaction voltage; a field current calculator for armature reaction compensation that calculates a field current component that compensates for and cancels the q-axis armature reaction voltage component; A field current command generation circuit that generates the field current command by adding the child reaction compensation field current signal and the d-axis component magnetizing current, and a field current command generating circuit that generates the field current command by adding the child reaction compensation field current signal and the d-axis component magnetizing current; Control of an AC motor, characterized in that it has a commutation overlap angle calculator for calculating an angle, and a phase command generation circuit that adds a commutation overlap angle signal and a power factor angle to generate a phase command for the power converter. Device.