JPH0526435B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to a vector control device for an induction motor.

In recent years, as a control method to improve the quick response of induction motors, the primary current of the motor is controlled by dividing it into an excitation current and a secondary current, and the secondary magnetic flux and secondary current vector are always orthogonal, so that the motor is equivalent to a DC motor. A vector control method has been proposed that attempts to obtain high responsiveness. However, when a voltage source inverter such as a pulse width modulation (PWM) inverter is used in the power converter that is actually used, the voltage becomes the manipulated variable even though the primary current is controlled. There was a problem in that the primary current did not flow as per the settings, resulting in poor response and making accurate variable speed control difficult.

In the primary voltage control of the electric motor, the present invention uses vector control that can cancel mutual interference between the secondary magnetic flux and the secondary current.
It is an object of the present invention to provide a vector control device that solves the problems of the conventional technology.

EMBODIMENT OF THE INVENTION Hereinafter, following the principle explanation of the present invention, embodiments will be explained in detail.

First, the voltage equation expressed by the α-β axis that rotates the induction motor in synchronization with the primary voltage is the following equation (1), and the generated torque T is the equation (2).

Here, each symbol is the quantity shown below.

e ₁ ; Primary voltage (α, β components) e′ ₂ ; Secondary voltage (α, β components)
β component) i ₁ : Primary current (α, β component) λ′ ₂ : Secondary magnetic flux (α, β component)
β component) r ₁ ; Primary resistance r ₂ ; Secondary resistance M; Exciting inductance L ₂ ; Secondary inductance L〓; Equivalent leakage inductance (L〓=L ₁ L ₂ −M ² /L ₂ ) L ₁ ; Primary inductance P: Differential symbol d/dt ω ₁ ; Power supply angular frequency ω _r ; Rotor angular frequency i′ ₂ ; Secondary current (α, β components) Equations (1) and (2) above can be expressed as 1
As shown in the figure, for the two-phase voltages e ₁ 〓, e ₁ 〓, the α-axis and β-axis components of the primary current and secondary magnetic flux i ₁ 〓, i ₁ 〓,
This is an equivalent block diagram of a dielectric motor that generates λ' ₂ 〓, λ' ₂ 〓 and torque T.

Here, the α and β axes, which rotate in synchronization with the primary voltage, may be set at any phase, but if the α axis is set in the direction of the secondary magnetic flux, the condition that the secondary current matches the β axis, i.e. , the conditions for the secondary current to be orthogonal to the magnetic flux are λ' ₂ 〓 = constant λ' ₂ 〓 = 0 ... (3), and the primary frequency ω ₀ is ω, as clarified by vector control theory. ₀ = ω _r + Mr ₂ /L ₂ λ′ ₂ 〓・i ₁ 〓 ……(4).

In this way, when the α and β axes are determined, the primary current i ₁
The α-axis component i ₁ 〓 (=constant) is a primary current corresponding to the magnetic flux λ′ ₂ , and the β-axis component i ₁ 〓 is a primary current corresponding to the secondary current i′ ₂ .

Next, when the conditions of equations (3) and (4) above are inserted into the block diagram of FIG. 1, the block diagram shown in FIG. 2 is obtained. That is, point a in FIG. 1 is controlled to zero from the relationship of equation (4). Since point c is λ′ ₂ 〓=0, all the quantities connected to this point are zero, and similarly i′ ₂ 〓
The quantity connected from =0 to point d is also zero, and since λ' ₂ = constant, its differential at point b is also zero. _Then , when calculating the part of the broken line block A in FIG. 1 under the same conditions, L ₂ /r ₂ =τ ₂ , L ₌
M ² )/L ₂ ], (L〓+M/L ₂ r ₂・L ₂ /L ₂ P+r ₂・M/L ₂ )ω ₀ i ₁ 〓=(L
〓＋M／L ₂ ／τ ₂ P＋１）ω ₀ i ₁ 〓 ＝L〓τ ₂ P＋L ₁ L ₂ −M ² ／L ₂ ＋M ² ／L ₂ ／τ ₂ P＋１ω ₀ i ₁ 〓
=L〓/L ₁ τ ₂ P+1/τ ₂ P+1L ₁ ω ₀ i ₁ 〓 Here, since i ₁ 〓=constant, let P=0 =L ₁ ω ₀ i ₁ 〓 In this way, Fig. The block diagram of
This becomes the block diagram shown in the figure. As is clear from Fig. 2, the secondary magnetic flux λ′ ₂ 〓 cannot be uniquely set by the α-phase primary voltage e ₁ 〓, but is determined by −L 〓 by the β-phase primary current i ₁ 〓.
There is interference of ω ₀ i ₁ 〓, and the secondary current i ₂ 〓 cannot be uniquely set by the β-phase primary voltage e ₁ 〓, but −L ₁ ω ₀ i due to the α-phase primary current i ₁ 〓. There is an interference of ₁ 〓. Therefore,
In the present invention, the primary voltage is adjusted so that the control amount is compensated in advance for the interference caused by the primary currents i ₁ 〓 and i ₁ 〓.
Correct e ₁ 〓, e ₁ 〓. For this correction, L〓ω ₀ i _{1 〓 added to the primary voltage e 1 〓} is taken into account when setting the voltage e ₁ 〓.
Subtract L〓ω ₀ i ₁ 〓, and add L ₁ ω ₀ i ₁ 〓 to the setting of the voltage e ₁ β in anticipation of L ₁ ω ₀ i _{1 〓} subtracted from the primary voltage e ₁ 〓. In addition, since the current _{i 1 〓} is accompanied by a first-order lag of [1/(L〓P+1r ₁ )] with respect to the voltage e 1 〓, the delay is included in the correction calculation of the voltage e ₁ 〓. Please make any corrections. By this correction, it is possible to set the primary voltages e 1 〓 and e ₁ 〓 that control the secondary magnetic flux λ′ ₂ 〓 and the secondary current i ₂ 〓 in a non-interfering _manner .

FIG. 3 is a block diagram showing one embodiment of the present invention. In controlling the electric motor 1 by supplying a voltage-controlled primary voltage from the PWM inverter 2 so that the magnetic flux and the secondary current run orthogonally to each other, the α and β phase voltages e ₁ 〓 and e ₁ 〓 are set. The correction calculation circuit 3 performs the above-mentioned correction. The correction calculation circuit 3 uses a coefficient multiplier 3 1 of primary resistance r 1 to set the α-phase primary voltage e _{1 〓} and to set the α-phase primary current setting value i ^* 〓 for controlling the secondary magnetic flux λ′ _{2 of} the motor to be constant. A correction value for correcting the interference amount shown in FIG. 2 due to the β-phase primary current i ₁ 〓 is subtracted from the value passed through the β-phase primary current i 1 . This correction value is calculated by adding the β-phase primary current setting value i ^* ₁ 〓 for controlling the secondary current i′ ₂ to the first-order lag [1/(L〓P
+r ₁ )] and its coefficient r ₁ The integrator 3 ₂ takes out the delay correction value i ^** ₁ 〓 for the i ^* ₁ 〓, and adds the power supply angular frequency ω ₀ to this correction value i ^** ₁ 〓. Multiply by multiplier 3 ₃ ,
This multiplication result is obtained by passing it through a coefficient unit ₃₄ having an equivalent leakage inductance L as a coefficient.

In addition, the correction calculation circuit 3 sets the β-phase primary voltage e ₁ 〓 by applying the α-phase primary current i ₁ 〓 to the value obtained by passing the set value i ^* ₁ 〓 through a coefficient unit 3 ₅ of the primary resistance r ₁ 〓. A correction value for correcting the interference shown in FIG. 2 is added in advance. This correction value is obtained by multiplying the current setting value i ^* ₁ 〓 by the power supply angular frequency ω ₀ in a multiplier 3 ₆ and passing the multiplication result through a coefficient multiplier 3 ₇ having a primary inductance L ₁ as a coefficient.

The β-phase primary current set value i ^* ₁ 〓 is a speed adjustment that performs proportional integral calculation (PI) of the deviation between the speed set value V ^* _S and the detected value (rotor angular frequency ω _r ) of the speed detector 4 coupled to the motor 1. It is obtained as the output of the device 5. The power supply angular frequency ω ₀ is obtained by the angular frequency calculation circuit 6. This arithmetic circuit 6 ^is a divider _{6 1} that divides the set value i ^* ₁ 〓 by the set value i ^** _{1 〓} which is corrected for the delay shown in FIG. 2.
and a coefficient unit ₆₂ that multiplies this division result i ^* ₁ 〓/i ^* ₁ 〓 by a coefficient 1/τ2 to calculate the slip angular frequency ω _S , and add the rotor angular frequency to this slip angular frequency ω _S. ω _r
, to obtain the power source angular frequency ω ₀ . Calculation of the slip angular frequency ω _S by the divider 6 ₁ and the coefficient unit 6 ₂ is based on the following conditions mentioned above in the second term on the right side of equation (4) and from _{FIG .} 〓・M is introduced and i ^** ₁ 〓/(i
^* ₁ 〓・
τ ₂ ).

L〓=L ₁ L ₂ −M ² /L ₂ τ ₂ =L ₂ /r _2The α and β phase primary voltages e ₁ 〓 and e ₁ 〓 with the interference corrected in this way are obtained by the phase voltage calculation circuit 7 2 phase -
Three-phase conversion is performed, the three-phase voltage setting values e ^* _a , e ^* _b , e ^* _c of inverter 2 are taken out, and according to these setting values
By controlling the primary voltage of the PWM waveform, speed or torque control is realized in the motor 1 with magnetic flux and secondary current not interfering with each other. Note that for the two-phase to three-phase conversion in the phase voltage calculation circuit 7, a sine wave SIN _{ω 0 t} and a cosine wave COS ω ₀ t are obtained from the trigonometric function generation circuit 8 using the power supply angular frequency ω 0 . In addition, in order to obtain a PWM waveform in the inverter 2, a constant multiplied triangular wave synchronized with the power source angular frequency ω ₀ is extracted from the triangular wave generation circuit 9 using the power supply angular frequency ω ₀ , and this triangular wave and the set voltages e ^* _a , e ^* _b , by level comparison with e ^* _c
I am getting a PWM waveform.

As described above, according to the present invention, when performing vector control of an induction motor using a voltage source inverter, the inverter set voltage e is determined from the motor excitation current set value i ^* ₁ 〓 and the secondary current set value i ^* ₁ 〓. ₁ 〓, e ₁ 〓, in order to cancel the mutual interference due to the motor's primary current i ₁ 〓, i _{1 〓} , the magnetic flux and secondary current are It is possible to control the vectors so that they are always perpendicular to each other, and there is an effect that accurate vector control can be performed over a wide control range. Moreover, highly accurate control is possible by correcting the primary voltage and angular frequency ω ₀ including the first-order lag component having a time constant of L〓/r ₁ .

[Brief explanation of the drawing]

Fig. 1 is an equivalent block diagram of an induction motor with respect to two-phase voltages e ₁ 〓, e ₁ 〓, Fig. 2 is an equivalent block diagram of vector control of an induction motor, and Fig. 3 is a control device showing an embodiment of the present invention. It is a block diagram. 1...Induction motor, 2...Voltage type inverter,
3... Correction calculation circuit, 4... Speed detector, 5...
Speed regulator, 6... Angular frequency calculation circuit, 7... Phase voltage calculation circuit, 8... Trigonometric function generation circuit, 9...
Triangular wave generation circuit.

Claims (1)

[Claims] 1. An induction motor is driven by a voltage source inverter, and two-phase three-phase voltage is generated from the α-phase voltage e ₁ 〓 that sets the magnetic flux component of the induction motor and the β-phase voltage e ₁ 〓 that sets the secondary current component. By conversion, the a, b, c phases 3 of the above voltage source inverter
In the induction motor vector control device that obtains the phase voltage set values e _a *, e _b *, e _c *, the α-phase primary current set value i ₁ 〓 *, which sets the magnetic flux component of the induction motor, is set to the primary resistance r ₁ β-phase primary current setting value i ₁ 〓
* Ratio of equivalent leakage inductance L〓 to primary resistance r ₁
The value i _{1 〓} ** multiplied by _the angular frequency setting value ω ₀ of the voltage source inverter and the above inductance L〓 The above voltage e ₁ 〓 is obtained by subtracting the value passed through the coefficient machine set to , and the above set value _{i 1} 〓* is subtracted from the value passed through the coefficient machine set to the primary resistance r ₁ . An induction motor characterized in that it is equipped with a correction calculation circuit that multiplies the angular frequency setting value ω ₀ by the angular frequency setting value ω 0 and adds a value passed through a coefficient unit set to the primary inductance L ₁ to obtain the voltage e ₁ 〓. Vector control device. 2 The above value i ₁ 〓** obtained from the above setting value i ₁ 〓＊ The value divided by the above setting value i ₁ 〓* is the ratio r ₂ /L ₂ of secondary inductance L ₂ and secondary resistance r ₂ Claim 1, characterized by comprising an angular frequency calculation circuit that calculates the angular frequency setting value ω _p by adding the rotor angular frequency detection value ω _r of the induction motor to the value passed through the coefficient unit to be set. A vector control device for an induction motor as described in .