JPH06100625B2 - Circuit element measuring device - Google Patents

Description

Description: TECHNICAL FIELD The present invention relates to a circuit element measuring device for measuring impedance such as resistance, capacitor and inductance, and more specifically to a synchronous detector in the device. The present invention relates to an improvement in a measuring device in which harmonics contained in a reference phase signal are reduced without impairing linearity.

[Prior Art] Next, the configuration of a circuit element measuring device according to the prior art will be described with reference to FIG. In FIG. 1, 1 is a sine wave oscillator as a measurement signal source, 2 is an impedance to be measured, 3 is an amplifier, 4
Reference numerals 5 and 5 are changeover switches, and 6 is a synchronous detector.

The output of oscillator 1 is applied to impedance 2 from terminal 1A. 2A and 2B are an input terminal and an output terminal of a measurement signal applied to the impedance 2, respectively. The amplifier 3 is for setting the output terminal 2B of the impedance 2 to zero potential,
3A is a reference resistance.

If the current flowing from the terminal 2A to the impedance 2 is Ix and the resistance value of the reference resistor 3A is Rs, the voltage appearing at the terminal 4A of the changeover switch 4 is IxRs.
On the other hand, if the input voltage to the impedance 2 is Vx, the voltage appearing at the terminal 4B of the changeover switch 4 is Vx.

A square wave having a reference phase is sent to the terminal 5A of the changeover switch 5 via the terminal 1B of the oscillator 1, and the terminal 5B of the changeover switch 5 has a phase difference with respect to the square wave of the terminal 1B via the terminal 1C of the oscillator 1. Send a 90 ° square wave.

Therefore, when the changeover switch 4 is connected to the terminal 4A and the changeover switch 5 is connected to the terminal 5A, the direct current component I _R proportional to the real component of the current vector comes out from the synchronous detector 6. When the changeover switch 4 is connected to the terminal 4A and the changeover switch 5 is connected to the terminal 5B, the synchronous detector 6 outputs the output I _I proportional to the imaginary component of the current vector.

The same applies to the voltage vector.
When connected to 4B and the changeover switch 5 is switched, an output V _R proportional to the real component of the voltage vector and an output V _I proportional to the imaginary component come out.

An example of these relationships is shown in FIG. As is clear from FIG. 2, since the orthogonal components of Vx and IxRs are obtained, the value of impedance 2 Zx is calculated from these values.
= Vx / Ix can be calculated.

[Problems to be Solved by the Invention] Next, an example of input / output waveforms of the synchronous detector 6 will be described with reference to FIG. FIG. 3A shows the waveform of the terminal 1A which is a sine wave, and FIG. 3A shows the waveform of the terminal 1B which is a square wave. Fig. 3C shows the waveform at terminal 1C, which is 90 ° out of phase with the waveform shown in Fig. 3B.

In the conventional device as shown in FIG. 1, in order to improve the directivity of the synchronous detector 6, a square wave is used as the reference signal from the terminals 1B and 1C.

When the signal from the changeover switch 4, that is, the signal from the element to be measured is es, and the signal from the changeover switch 5, that is, the reference signal is er, the synchronous detection output V0 is V ₀ = es due to the multiplication operation of the synchronous detector. (T) · er (t) (1)

In the synchronous detector 6 in the conventional device, a square wave is used as a reference signal, and the square wave er is represented by a Fourier series: er (t) = 4 [sinωt + sin3ωt + Sin5ωt + ...]
/ Π becomes (2).

Here, assuming that the signal under measurement is a sine wave having no harmonic component, es (t) = es ₁ · sin (ωt + θ ₁ ) ... (3) where θ ₁ is expressed as a phase difference with respect to the reference signal. .

When the measured signal es is synchronously detected with a square wave er and the detected waveform is integrated with a cycle nT that is an integral multiple of the fundamental wave, Here, the average value for one cycle when θ ₁ = 0 ° and n = 1 And ask Becomes

This corresponds to an average value when the sine wave is full-wave rectified. The detection waveform at this time is the waveform of FIG. 3D, which is the output waveform when the changeover switch 4 is connected to the terminal 4B and the synchronous detection is performed in FIG.

It is to be noted that FIG. 3E shows an output waveform when the synchronous detection is performed in FIG. 3C, and the average value is zero when θ ₁ = 90 °.

Next, when e is a Fourier series when the measured signal contains harmonic components (when the waveform is distorted), e's (t) = [es ₁ · sin (ωt + θ ₁ ). + Es ₂ · sin (2ωt + θ ₂ ) + es ₃ · sin (3ωt + θ ₃ ) + ...] (5).

Performing synchronous detection with e's and er and integrating the detected waveform for nT period, Becomes

From equations (4) and (6), it can be seen that if the measured signal contains harmonic components, the influence of odd-order (3rd, 5th, ...) Harmonic components appears. .

In other words, the synchronous detector 6 when the waveform shown in Fig. 3 (a) is added calculates x1 in the range of 0 to 180 °, and 180 ° to 36
In the range of 0 °, x (-1) is calculated.

3A and 3C have odd-order harmonics, so if the waveform in FIG. 3A is distorted or the impedance 2 has a non-linear characteristic, the synchronous detector 6
The DC component is generated by the odd harmonic components included in the input signal of, and causes an error when measuring the capacitor.

The present invention uses an output of a Walsh function generator synchronized with a sine wave oscillator as a reference phase signal, and uses a synchronous detector having a weighting circuit and a multiplication circuit to reduce the influence of harmonics of the reference phase signal. An object is to provide an element measuring device.

[Means for Solving the Problem] In order to achieve this object, according to the present invention, there are provided a clock generation circuit 11A, a frequency divider circuit 11B that receives an output of the clock generator circuit 11A as an input, and a frequency divider circuit 11B. With output as input,
It has a sine wave synthesizing circuit 11C that synthesizes a sine wave, and inputs the output of the sine wave oscillator 11 as a signal source for measurement and the frequency dividing circuit 11B of the sine wave oscillator 11 to generate a Walsh function and generate 90 ° of each other. The Walsh function generator 12 that outputs an output having a phase difference and the output of the Walsh function generator 12 are used as D inputs,
A weighting circuit that inputs the output of the Walsh function generator 1 and the D-type flip-flop 15 that operates at the timing of the clock generation circuit 11A and weights the output of the measurement circuit 14.
A synchronous detector 13 having a multiplication circuit for multiplying the output of the weighting circuit 13L and the output of the Walsh function generator is provided.

[Operation] Next, the configuration of the circuit element measuring apparatus according to the present invention will be described with reference to FIG. In FIG. 4, 11 is a sine wave oscillator as a measurement signal source, 12 is a Walsh function generator which outputs outputs having a phase difference of 90 ° in synchronization with the output of the sine wave oscillator 11, 13 is a weighting circuit and a multiplier Reference numeral 15 is a D-type flip-flop having a circuit.

Further, 14 is a measuring circuit, which has an impedance of 2 in FIG.
And amplifier 3, including oscillator 1 terminal 1A to changeover switch 4
It corresponds to the circuit up to.

Next, a configuration diagram of an embodiment of the sine wave oscillator 11 and the Walsh function generator 12 will be described with reference to FIG.

The sine wave oscillator 11 includes a clock generator circuit 11A and a frequency divider circuit 1
There is a 1B and a sine synthesis circuit 11C. There are various types of circuits for generating a sine wave with the structure shown in FIG. 5, and the one described in Japanese Patent Application No. 52-31898 is an example.

By adding the output of the frequency dividing circuit 11B to the adding circuit 12A,
While maintaining synchronization between the output of the sine wave oscillator 11 and the output of the Walsh function generator 12, a 90 ° phase change is given.

The Walsh function generator 12 includes an adder circuit 12A and a Walsh function generator circuit 12B. The adder circuit 12A includes a frequency divider circuit.
The binary code output of 11B and the signal from the changeover switch 12C are added to perform binary addition.

The changeover switch 12C is for giving the reference phase of 0 ° and 90 ° to the synchronous detector 13. If the order of the frequency divider 11B is 3, 0 ° is “000”, 90 ° is “010”, and 180 ° is “10”.
"0" and 270 ° are represented as "110". Also, in FIG.
Although 0 ° and 90 ° are used, states in which they have a phase difference of 90 ° with each other, for example, 90 ° and 180 °, 180 ° and 270 ° may be used.

Next, FIG. 6 shows an embodiment circuit of the Walsh function generator 12 and the D-type flip-flop 15. In FIG. 6, there are two exclusive OR circuits EOR in the Walsh function generating circuit 12B, and WAL (1) and WAL (5) are output. In addition,
For an example of the Walsh function generation circuit 12B, see Japanese Patent Application No.
-31898.

Next, 2 corresponding to the output of the Walsh function generator 12B
A state diagram in which two D-type flip-flops are connected is shown in FIG. Reference numerals 15A and 15B in FIG. 6 are D-type flip-flops, which operate with the output of the clock generator 11A as the clock input. This is because the reference phase is changed according to the phase information from the changeover switch 12C, but the timing is shaped so that the phase angle is exactly 0 ° and 90 °.

That is, when the frequency of the function becomes high, it becomes difficult to perform accurate 90 ° phase control due to the delay time difference between the logic circuits forming the adder circuit 12A and the Walsh function generating circuit 12B. Therefore, the D-type flip-flop 15 is used to shape the timing before inputting to the synchronous detector 13. For example,
The D-type flip-flop is not necessary when the frequency handled is about 1 kHz, but the D-type flip-flop is necessary when the frequency is 10 kHz or higher.

Next, an embodiment circuit of the synchronous detector 13 will be described with reference to FIG. 13A in FIG. 7 is a phase inverting amplifier circuit having a gain of 1, and 13B to 12
E is a multiplication circuit, 13F to 13H are inverting negation circuits, and 13K is an addition circuit. The multiplication circuits 13B to 13E are analog switches,
Multiply x1 and x0.

Next, as an example of the synchronous detector using the Walsh function wave, FIG. 9 shows a waveform example in the case where the synchronous detector is configured using WAL (1) and WAL (5) of the Walsh function wave.

Assuming that the waveform of WAL (1) is er ₁ and the waveform of WAL (5) is er ₅ as the reference signal, each waveform can be expressed as a Fourier series: er ₁ (t) = (4 / π) ・ [sinωt + 1 / 3 ・ sin3ωt ＋ 1/5 ・ sin5ωt ＋…) Becomes

Here, the signals from the measurement circuit 14 include the multiplication circuits 13B to 13E.
Before entering, set the composite ratio of the reference signal WAL (1) and WAL (5). That is, weighting is performed by the weighting circuit 13L of 1: 2.414,
If the synthesized reference waveform is erw, Becomes This ratio is WAL (1) and WAL of Walsh function wave
It is the ratio when the sine waves are combined in (5).

here, Then, erw (t) = (4 / π) · kw · [sinωt + 1/7 · sin7ωt + 1/9 · sin9ωt + ...] (7)

Here, the measured signal es of equation (3) that does not have harmonic components
When the detection waveform is integrated for nT periods without performing synchronous detection with the composite wave erw of the Walsh function wave of equation (7), Becomes Expression (8) corresponds to the constant kw times of Expression (4).

Next, es of equation (5) when the measured signal contains a harmonic component is converted to the composite wave erw of the Walsh function wave of equation (7).
When synchronous detection is performed with and the detected waveform is integrated for nT period, Becomes

The principle of the synchronous detector 13 shown in FIG.
-117657.

Next, an example of the equalized spectrum for the reference signal of the synchronous detector 13 will be described with reference to FIG. Figure 8 shows a synchronous detector
DC is added to the measurement circuit 14 side of 13 and multiplication is performed, and this is observed at the output terminal 15. In FIG. 8, only the 7th harmonic of the fundamental wave is recognized in the equalized spectrum, and the 3rd harmonic and the 5th harmonic are not emitted. The level of the seventh harmonic is almost the same level as the level of the seventh harmonic of the square wave shown in FIGS. 3A and 3B, which is about 1/7 of the fundamental wave.

[Effects of the Invention] According to the present invention, since the Walsh function wave is used as the reference signal to the synchronous detector, the output signal of the device under test input to the synchronous detector contains odd harmonic components. In this case, the third and fifth harmonics of low order are eliminated, and the influence of measurement error due to the low order harmonics can be reduced.

It should be noted that the present invention can be used not only for measuring circuit elements but also for circuit analysis.

[Brief description of drawings]

FIG. 1 is a block diagram of a circuit element measuring device according to the prior art, FIG. 2 is a vector diagram of impedance 2, FIG. 3 is an example of an input / output waveform diagram of a synchronous detector 6, and FIG. 4 is a circuit according to the present invention. FIG. 5 is a block diagram of an embodiment of an element measuring device, FIG. 5 is a block diagram of an embodiment of a sine wave oscillator 11 and a Walsh function generator 12, FIG. 6 is an embodiment circuit of the Walsh function generator 12, and FIG. An embodiment circuit of the detector 13, FIG. 8 is an example of an equalized spectrum diagram for the reference signal of the synchronous detector 13, and FIG. 9 is a synchronous detector 13 when a Walsh function wave is used as the reference signal.
2 is an example of an input / output waveform of 1 ... Oscillator, 2 ... Impedance, 3 ... Amplifier,
4 ... Changeover switch, 5 ... Changeover switch, 6 ... Synchronous detector, 11 ... Sine wave oscillator, 12 ... Walsh function generator, 13 ... Synchronous detector, 14 ... Measuring circuit, 15 ... D-type flip-flop.

Claims (1)

[Claims] 1. A clock generating circuit (11A) and a frequency dividing circuit (1) for dividing the frequency of the output of the clock generating circuit (11A) as an input.
1B) and the output of the frequency divider circuit (11B) as input, and has a sine wave synthesizer circuit (11C) that synthesizes a sine wave, and uses a sine wave oscillator (11) as a signal source for measurement and a sine wave oscillator (11C). ) Input the output of the frequency divider circuit (11B), generate the Walsh function and output the Walsh function generator (12) with a 90 ° phase difference, and the output of the Walsh function generator (12). A weighting circuit (D) that inputs the D-type flip-flop (15) that operates at the timing of the clock generation circuit (11A) and the output of the Walsh function generator (12) and weights the output of the measurement circuit (14) 13L)
And a synchronous detector (13) having a multiplication circuit for multiplying the output of the weighting circuit (13L) and the output of the Walsh function generator (12).