JPH0634613B2 - AC motor controller - Google Patents

Description

The present invention relates to a control device for an AC electric motor driven by a thyristor power converter.

[Conventional technology]

FIG. 11 is a block diagram showing a conventional AC motor control device described in Japanese Patent Publication No. 59-1077.
In the figure, 1 is a first converter for converting AC from a commercial AC power source into DC, 2 is a second converter for converting the DC into AC of variable frequency, 3 is a synchronous motor, and F is its synchronous motor. The field windings 3 and 4 are position detectors 5 which output position signals of a phase corresponding to the rotational angle position of the rotary shaft of the synchronous motor 3.
Is a γ control circuit for controlling the commutation advance angle γ of the second converter 2 by phase shifting the position signal of the position detector 4 according to the magnitude of the motor armature current, and 6 is the output of the γ control circuit 5. A gate output circuit for outputting a gate signal of the second converter 2 according to a signal,
Reference numeral 7 is a speed generator, 8 is a speed command circuit, 9 is a speed deviation amplifier that amplifies the speed command signal of the speed command circuit 8 and a speed feedback signal which is an output signal of the speed generator 7, and 10 is a first converter. 1, a current detector for detecting the AC input current,
Reference numeral 11 is a current deviation amplifier for abutting and amplifying the output signal of the speed deviation amplifier 9 and the current feedback signal of the current detector 10.
Reference numeral 2 is a gate pulse phaser for controlling the firing phase of the first converter 1 based on the output signal of the current deviation amplifier 11, and 13 is a field for outputting a field current command Ifp for commanding the magnitude of the field current If. A magnetic command circuit, 14 is a current detector that detects the magnitude of the AC input current of the thyristor circuit 17 that supplies the field current If to the field winding F, and 15 is the field current command Ifp and the output of the current detector 14. A current deviation amplifier for butt-amplifying signals and a gate pulse phase shifter 16 for controlling the firing phase of the thyristor circuit 17.

The components denoted by the above-mentioned symbols 7 to 12 are speed control circuits for controlling the magnitude of the input current of the first converter 1, that is, the magnitude of the armature current of the electric motor 3 which is proportional to this, according to the speed deviation. And the components with the additional symbols 4 to 6 are current detectors 1.
A commutation lead angle control circuit for controlling the commutation lead angle γ of the second converter 2 according to the output signal of 0, that is, the armature current,
The components denoted by appendixes 13 to 17 respectively form field control circuits that allow the field current If to flow in proportion to the field current command Ifp.

The operation of each of these circuits is the same as that of the already known so-called thyristor motor device, and therefore detailed description thereof is omitted.

FIG. 12 is a vector diagram showing the relationship between the voltage and current of the electric motor 3 in FIG. In FIG. 12, (a)
Is no load, (b) is a load when the field current If is kept constant and the commutation advance angle γ is controlled so that the power factor is constant, and (c) separately shows the field current If. It is a vector diagram when controlling so that it may be proportional to the child current Ia and operating with the commutation lead angle γ constant.

As is clear from Figure 12 (b), even if the power factor as was maintained to a predetermined value, reduction In conjunction increase in the terminal voltage V is the armature current Ia (from Ia ₁ Ia ₂₎ to (V ₁ To V ₂ ). Due to this voltage drop, the maximum commutable current value in the second converter 2 drops. As a result, a sufficient output cannot be obtained from the electric motor 3.

In the case of FIG. 12 (c), the increase of the armature current Ia (Ia ₁
Since Ia ₂₎ the terminal voltage V becomes accompanied in is V ₂₎ from the raised (V ₁ from not inconveniences in FIG (b).

However, since the terminal voltage V becomes higher than that at the time of rating at the time of overload, the thyristor of the second converter 2 needs to have a high withstand voltage. In addition, since the electric motor itself causes magnetic saturation, a large output may not be obtained as expected. Further, there is a disadvantage that the power factor (power source power factor) of the first converter 1 is reduced as a result of the terminal voltage V being reduced when the load is light.

In addition, as a means for solving the above problems, Japanese Patent Publication No. 59-107
No. 7, gazette controls the magnitude of the no-load induced voltage E ₀ obtained by vector-wise adding the terminal voltage and the synchronous reactance drop and the phase difference between the no-load induced voltage E ₀ and the armature current Ia. Therefore, the method of controlling the terminal voltage to be constant regardless of the armature current is described in detail.

Although FIG. 13 is a vector diagram showing this operation principle, this operation will be briefly described here. In order to keep the terminal voltage V constant, the magnitude of the no-load induced voltage E ₀ and the phase difference θ between the E ₀ and the terminal voltage according to the magnitude of the armature current Ia.
While controlling the (phase difference angle), the phase (φ + θ) of the second converter is controlled while maintaining the relationship of φ + θ so that the phase difference φ between the armature current Ia and the terminal voltage becomes constant.

However, in this method, since the terminal voltage is controlled to be constant, the commutation overlapping angle u of the second converter changes according to the magnitude of the armature current, and the arm element of the second converter is obtained. The application period (φ-u) of the reverse voltage to the thyristor changes.

At this time, when the second converter is made multi-phase (for example, 12 phases) to reduce the torque pulsation and the large-capacity thyristor motor is driven, commutation is performed every 30 ° so that the other phase is changed. Due to the effect of commutation, as shown in FIG. 14, the reverse voltage period of the thyristor, which is an arm element, is 30 ° -even if φ> 30 °.
In order to achieve u and to perform stable commutation of the second converter, it is necessary to set the terminal voltage such that this commutation overlap angle does not become very large with respect to an increase in armature current.

Further, in order to control this voltage with high accuracy, the magnetic saturation characteristic of the AC motor 3 must be taken into consideration, and the conventional device disclosed in the above publication has a problem in accuracy.

[Problems to be solved by the invention]

Since the conventional AC motor control device is configured as described above, the terminal voltage and power factor fluctuate significantly due to load fluctuations, the commutation of the second converter becomes unstable, and sufficient output is obtained. Can't get Also, when the armature terminal voltage is detected, it is affected by the commutation of the inverter that constitutes the power converter,
There was a problem that it could not be detected accurately.

The present invention has been made to solve the above problems, and can prevent fluctuations in the terminal voltage and power factor due to load fluctuations and perform stable commutation to obtain sufficient output. Another object of the present invention is to obtain a control device for an AC motor that can detect the terminal voltage with high accuracy.

[Means for solving problems]

The control device for an AC motor according to the present invention, based on the command signal of the no-load terminal voltage command circuit and the command signal of the power factor angle command circuit, determines the phase difference between the terminal voltage and the no-load induced voltage according to the magnitude of the armature current. a vector calculator for controlling θ (phase difference angle) and field current, and for controlling the magnitude of the terminal voltage so as to secure a predetermined commutation margin angle; and a DC voltage control circuit for the second converter, It is provided with a terminal voltage control circuit of the electric motor and a data reader for reading the terminal voltage of the electric motor at a commutation start point of an inverter forming a power converter.

[Action]

The control device for the AC motor according to the present invention controls the locus of the terminal voltage by the vector calculator so as to be parallel to the axis (d-axis) of the field current, and the field current is the magnetizing current for generating the terminal voltage. The d-axis component is controlled by the sum of the field current component for compensating the armature reaction electromotive force component generated on the axis (q axis) orthogonal to the field current axis, and the DC voltage control circuit controls the second Compensation control of the ignition phase of the converter of
In addition, the terminal voltage of the motor is read at the commutation start point of the inverter that constitutes the power converter, and the terminal voltage is detected by the terminal voltage control circuit without being affected by the voltage drop due to commutation (commutation overlap angle). The commutation operation is stabilized by correcting the field current.

〔Example〕

An embodiment of the present invention will be described below with reference to the drawings. First
In the figure, 18 is a power factor angle command circuit for commanding the phase angle (power factor angle) φ of the armature current with respect to the terminal voltage of the electric motor 3,
Reference numeral 19 is a no-load terminal voltage command circuit that commands a terminal voltage of the electric motor 3 when there is no load, and 20 is a vector calculator, which includes the power factor angle command circuit 18 and the no-load terminal voltage command circuit 1 described above.
9 command and armature current Ia are input, field current command If
It outputs p and the phase command β of the second converter 2. A phase control circuit 34 controls the conduction phase angle of the second power converter 2 based on commands from the position detector 4 and the vector calculator 20. It also has a function of generating an interrupt signal INT (gate signal) at the inverter commutation start point.

Reference numeral 21 denotes a DC voltage command circuit, which is an armature current signal Ia.
And the terminal voltage signal Vfb of the electric motor 3 and the vector calculator 2
The commutation lead angle γ of 0 is input, and the DC voltage command Eref is output.

A DC voltage deviation amplifier 22 is a DC voltage signal Efb which is an output signal of the DC voltage detector 31 of the second converter 2.
The difference between the DC voltage command Eref and the DC voltage command Eref is amplified.

Reference numeral 23 is a first switch, the opening and closing of which is controlled by a first level discriminator 24 which discriminates the speed and the level of the armature current, and turns on and off the output signal of the DC voltage deviation amplifier 22.

Reference numeral 25 denotes a first adder, which is the first switch 23.
And the output signal of the vector calculator 20 are added,
The output is given to the phase control circuit 34 as the phase command β.

26 is a terminal voltage command circuit, which is a vector calculator 20.
The terminal voltage V and the armature current signal Ia are input, and the terminal voltage command Vref is output.

27 is a terminal voltage deviation amplifier, which is a terminal voltage detector 3
The difference between the terminal voltage signal Vfb which is the output signal of No. 3 and the terminal voltage command Vref of the terminal voltage command circuit 26 is amplified.

Reference numeral 28 denotes a second switch, the opening and closing of which is controlled by a second level discriminator 29 which discriminates the speed level, and turns on and off the output signal of the terminal voltage deviation amplifier 27.

Reference numeral 30 denotes a second adder, which is the second switch 28.
Output signal Ifp and the output signal Ifp of the vector calculator 20 are added, and the output is given to the current deviation amplification 15 as a field current command.

32 is a transformer for detecting the terminal voltage of the electric motor 3 (hereinafter referred to as P
T is abbreviated as T), 33 is a diode rectifier that detects the output signal of the PT as direct current, and 35 is a filter ring that filters the detection signal Vfb 'of the diode rectifier 33. A data reader 36 reads the output Vfb ″ of the filter 35 in response to an interrupt signal INT generated by the inverter phase shifter 34.

FIG. 2 shows a detailed configuration diagram of the vector calculator 20. Second
In the figure, 201 is a θ function table that outputs a phase difference angle θ based on the no-load terminal voltage V ₀ , armature current Ia, and power factor angle φ, and 202 is a terminal voltage V calculated from the output of this θ function table 201 and V _0. A terminal voltage calculation circuit 203 for converting the magnetizing current i from the output signal of the terminal voltage calculation circuit 202.
No-load saturation curve table of electric motor 3 for calculating μ, 20
Reference numeral 4 denotes an i-d which outputs the d-axis component iμd of the magnetizing current from the output signal of the no-load saturation curve table 203 and the phase difference angle θ.
A μd arithmetic circuit, 205 is an Eaq arithmetic circuit that calculates the q-axis armature reaction voltage component Eaq from the armature current Ia and the power factor angle φ,
Reference numeral 206 denotes an ifa arithmetic circuit that calculates the compensating field current component ifa of the armature reaction from the output signal of the Eaq arithmetic circuit 205.
207 is an adder as a field current command generating circuit for adding the output signals of the ifa arithmetic circuit 206 and the iμd arithmetic circuit 204, and 208 is a commutation overlapping angle u (hereinafter, u) depending on the terminal voltage V and the power factor angle φ. 209 is an adder for adding the output signal u / 2 of the u arithmetic circuit 208 and the power factor angle φ, and 210 is a commutation advance angle γ and a phase difference angle θ of the adder 209. Is an adder that adds the phase command generating circuit.

Next, the operation of the above embodiment will be described with reference to the vector diagram shown in FIG. When the direction of the field current is the d-axis and the axial direction orthogonal to this is the q-axis as the reference axis, the no-load induced voltage of the electric motor 3 is generated in the q-axis direction.

The basis of the control means in the present invention is that the vector locus of the terminal voltage V changes in parallel with the d-axis direction according to the armature current Ia with respect to the no-load terminal voltage V ₀ on the q-axis. Is to control.

If the phase difference (phase difference angle) between the terminal voltage V and the q axis is θ, and the phase difference (power factor angle) between the armature current Ia and the terminal voltage V is φ,
The terminal voltage V is obtained as the vector sum of the no-load terminal voltage V ₀ and the armature reaction voltage component Ead = XaqIacos (φ + θ) generated in the d-axis direction, and the following equation holds.

V ₀ tan θ = XaqIacos (φ + θ) (1) Equation (1) is modified to obtain Equation (2).

Here, the left side of the equation (2) shows the per unit value of the d-axis armature reaction voltage component with respect to the no-load terminal voltage V ₀ . The θ function table 201 uses the power factor angle φ as a parameter, It is a table for obtaining the phase difference angle θ from, and the phase difference angle θ for the predetermined power factor angle φ can be obtained by inputting the perunit value on the left side of the equation (2).

FIG. 4 is a graph showing an example of the θ function table 201.

The terminal voltage V is obtained from the following equation as a function of the phase difference angle θ.

The V calculation circuit 202 calculates the terminal voltage V according to the equation (3).

Next, the magnetizing current iμ generated in the direction orthogonal to the terminal voltage V is obtained by the no-load saturation curve table 203. The no-load saturation curve table 203 has an example of the curve 1 as shown in the graph of FIG.
And shows the relationship between the induced voltage and the field current at a predetermined speed in consideration of the magnetic saturation of
μ corresponds to the combined electromotive force of the electric motor 3.

The d-axis component iμd of the magnetizing current iμ is calculated according to the following relational expression, and the iμd calculation circuit 204 executes the calculation of Expression (4).

iμd = iμcos θ (4) On the other hand, the armature reaction voltage component Eaq in the q-axis direction is given by the following relational expression and calculated by the Eaq calculation circuit 205.

Eaq = XadIasin (φ + θ) (5) This armature reaction voltage component Eaq in the q-axis direction is controlled so as to be compensated by the field current component ifa in the d-axis direction.
In this case, from the armature reaction voltage component Eaq to the field current component i
The conversion of fa is executed by the ifa arithmetic circuit 206, and the tangential characteristic of the no-load saturation curve shown in FIG.
Converted using Kfa as a coefficient.

ifa = Kfa · Eaq (6) The d-axis field current component iμd, ifa obtained according to the above equations (4) and (6) is added by the adder 207,
The field current command Ifp is obtained by the following equation.

Ifp = iμd + ifa (7) The phase command β of the second converter 2 is calculated from the phase difference angle θ, the power factor angle φ, and the commutation overlapping angle u with respect to the q-axis direction by the following relational expression. Given in sum.

At this time, the commutation advance angle γ of the second converter 2 with respect to the terminal voltage V is as follows.

Here, the commutation overlap angle u is expressed by the following equation.

Note that equation (10) is And it is obtained by eliminating the commutation advance angle γ from the equation (9). Further, the equation (10) is the direct current of the second converter 2.
Since it is a function of Id, it is necessary to convert this DC current Id into the fundamental wave effective value Ia of the armature current. Considering the commutation overlap angle u, the armature current has a trapezoidal wave shape as shown in FIG. 6, and the fundamental wave effective value Ia of the armature current at this time is Ia.
Is a function of the commutation overlap angle u as follows.

However, in a large-capacity thyristor motor of 12 phases or more, the commutation overlap angle u is generally not controlled to 20 ° to 25 ° u, so that the reverse voltage period for turning off the thyristor cannot be secured. In this case, equation (11) Is 1 to 0.992, which is practical It doesn't matter. Therefore, if we transform equation (10), Then, the u arithmetic circuit 208 executes the arithmetic operation according to the equation (12).

As described above, since the present invention is controlled in accordance with the vector relational expressions of the above expressions (1) to (3), in order to secure the commutation margin angle (reverse voltage application period) of 30 ° -u of the thyristor, The power factor angle φ and the no-load terminal voltage V ₀ may be selected as appropriate values.

The phase control circuit 34 may perform a phase operation that advances the output signal of the position detector 4 set to the same phase as the q-axis direction by the phase command β. Various phase control methods are practically used. Since this is a well-known technique, it will not be described here.

However, due to the influence of the calculation accuracy and the calculation time of the vector calculator 20, the actual magnitude and phase of the terminal voltage V may differ from the calculation signal of the vector calculator 20.
The DC voltage difference amplifier 22 and the terminal voltage difference amplifier 27 are provided as means for correcting the calculation error and the response delay. The DC voltage command Eref is calculated by the DC voltage command circuit 21 by the following equation.

Here, a conversion formula that is an approximate value of formula (11) In accordance with the above, it is determined by the armature current Ia. The second term on the right side of the equation (13) represents the commutation voltage drop component, and the third term represents the drop amount of the armature resistance ra.

Note that the first term Vfb on the right side is the output signal of the data reader 36 and is proportional to the actual value of the terminal voltage of the electric motor 3. The first term is when the commutation voltage and the resistance drop are ignored. It is the average value of the DC voltage of the second power converter 2.

FIG. 7 shows a circuit configuration for detecting the motor voltage without being affected by the commutation of the inverter 2 as the second power converter. The terminal voltage of the electric motor 3 is rectified by the diode rectifier 33 through the PT 32 to generate a direct current.

Figure 8 shows the voltage and current waveforms of each part of the detection circuit.
8 (a), (b), and (c) show the output of PT32 (the input of the diode rectifier 33), and the motor terminal voltage waveform V
_{Although they are UV} , V _VW , and V _WU , due to the commutation of the inverter 2, the voltage waveform is not a sine wave but the waveform shown in the figure including a voltage drop.

When this terminal voltage is rectified by the diode rectifier 33, as shown in FIG. 8 (d), the voltage is dropped at the rectification timing due to the influence of commutation of the inverter 2. In the past, when reading this voltage, in order to avoid the influence of this commutation, the filter 35 had to be strengthened considerably, but the time constant of the filter 35 was made too large from the viewpoint of control response (phase shift delay). It was not possible to select it, and it was affected by commutation and ripple due to the rectified waveform of a sine wave, so that accurate voltage detection could not be performed and it was a disturbance.

Therefore, the phase control for performing the inverter phase shift control by the inverter phase command αc from the controller 40 (all the controllers other than the controller shown in FIG. 7 are combined into the controller 40) and the signal of the phase detector 4 The circuit 34 is provided with an inverter phase shifter 342 and an interrupt signal generation circuit 341 for generating an interrupt signal at each commutation timing of the inverter 2, and the interrupt signal is input to the data reader 36 to output the interrupt signal. By reading the electric motor voltage at a timing before the voltage drops due to commutation, it is possible to detect almost the peak value of the terminal voltage without being affected by commutation.

As a result, as shown in FIG. 8 (e), the filter 35 causes the delay time after the interrupt signal is generated until the data reader 36 operates (the processing time of S / W in the case of S / W), A terminal having a time constant sufficient to hold the terminal voltage and remove the power converter 1 and the field current lip is sufficient, and the time constant can be considerably reduced. Therefore, the dead time such as a phase shift delay with respect to the control system (voltage control system) can be minimized, and the motor terminal voltage can be detected with high accuracy.

FIG. 9 is a terminal voltage waveform diagram when the electric motor 3 has three phases. When the commutation advance angle γ fluctuates in the range of 30 ° to 90 °, it fluctuates as shown in FIG. 10 (a). When the peak value of the motor terminal voltage is E _M peak and the output of the data reader 36 is Ed _M fb, the relational expression is Ed _M fb = E _M peak sin (60 ° + γ), 30 ° γ60 ° (13− 1) Ed _M fb = E _M peak sin γ, 60 ° ≦ γ ≦ 90 ° (13-2).

Therefore, in order to obtain a more accurate electric motor terminal voltage peak value, from equations (13-1) and (13-2),
E _M peak voltage is Therefore, as shown by the dotted line in FIG. 7, the calculation of the equations (13-3) and (13-4) is performed by inserting the data corrector 36b into the output device of the data reader 36 to perform the correction calculation. The fluctuation of Ed _M fb due to the fluctuation of commutation lead angle γ is corrected, and the peak value of the terminal voltage can always be detected with high accuracy.

In addition, this method fluctuates as shown in FIG. 10 (b) even when the electric motor has 6 phases (12 pulses), and its relational expression is Therefore, if the same correction as in the case of the three-phase motor is performed by the equations (13-5) and (13-6), the peak value of the terminal voltage can always be detected accurately.

The DC voltage deviation amplifier 22 amplifies the deviation so that the DC voltage signal Efb becomes equal to the DC voltage command Eref,
The output is output via the first switch 23 to the first adder 2
In step 5, it is added to the phase command β of the vector calculator 20 and given to the phase control circuit 34.

The output signal of the DC voltage deviation amplifier 22 serves as a correction signal for the phase control angle of the second converter 2, and the second converter 2
The DC voltage of follows the DC voltage command Eref.
In this way, the commutation lead angle γ of the second converter 2 is controlled to a predetermined value as shown by the equation (13), so that commutation failure can be prevented.

The first switch 23 is closed when the speed of the electric motor 3 is a predetermined speed or higher or the armature current is a predetermined value or higher. In the extremely low speed region, the level of the terminal voltage becomes extremely low, and when the output of the PT 32 is rectified and detected by the diode bridge circuit as the terminal voltage detector 33, the drop in the diode deteriorates the detection accuracy of the terminal voltage. .

Further, when the armature current is extremely small, the pulsating component causes an intermittent phenomenon of the current, and the arm thyristors of the first and second converters 1 and 2 are turned off, and the second converter 2 The DC voltage of becomes zero at each timing of the current interruption, and the relational expression of the equation (13) does not hold. In order to prevent these phenomena, the first level discriminator 24
There is provided a first switch 23 which is opened / closed by discriminating the speed and the level of the armature current, and the output signal of the DC voltage deviation amplifier 22 is turned off when the speed is low or the armature current is small.

The terminal voltage command Vref is calculated by the following formula.
Is calculated by.

Vref = V + raIa (14) where the first term on the right side is the calculated value of the terminal voltage of the vector calculator 20, and the second term is the drop in the armature resistance ra of the motor 3. There is. The terminal voltage deviation amplifier 27 amplifies the deviation between the terminal voltage command Vref and the average value Vfb of the DC voltage of the power converter 2, and the output is a vector calculator in the second adder 30 via the second switch 28. 20 field current command
It is added to Ifp and given to the field current deviation amplifier 15.

The output signal of the terminal voltage deviation amplifier 27 functions as a correction signal for the field current command, and the terminal voltage of the electric motor 3 follows the terminal voltage command Vref. In this way, fluctuations in the terminal voltage of the electric motor 3 due to errors in the field current command Ifp can be prevented, and fluctuations in the output of the electric motor 3 can be suppressed.

The second switch 28 is closed by the second level discriminator 29 only when the speed of the electric motor 3 is equal to or higher than a predetermined speed. This is because the detection accuracy of the average value Vfb of the DC voltage of the power converter 2 is deteriorated when the speed is low as described above.

In the above embodiment, the constants Kad, Kaq, and Kc mean d-axis armature reaction reactance, q-axis armature reaction reactance, and commutation reactance, respectively, and these constants are proportional to the frequency of the electric motor 3. Although it is omitted for convenience of explanation, it may be changed by a table or the like according to the output signal of the speed generator 7. Similarly, the no-load saturation curve table 20
When the magnetizing current iμ is calculated by 3, the terminal voltage V, which is the input signal thereof, may be converted into a signal inversely proportional to the speed of the electric motor 3 and given.

In the above embodiment, the detection signal of the armature current Ia is used as the input signal of the vector calculator 20, but the output signal of the speed deviation amplifier 9 may be used. If the response characteristic of the current deviation amplifier is enhanced so that the deviation between the detection signal of the armature current Ia and the reference signal of the armature current which is the output signal of the speed deviation amplifier 9 becomes small, the same effect as in the above-described embodiment can be obtained. .

Further, in the above embodiment, the calculation of the vector calculator 20 may be digitally processed by a microcomputer or the like, and in this case, the calculation accuracy is improved as compared with the analog one. Further, in the above-described embodiment, a 6-phase rectifier circuit is shown as the second converter 2 in FIG.
Even if a plurality of the second converters are arranged in parallel or in series to form a rectifying circuit of 12 phases or more,
The same effect as that of the above embodiment is obtained.

Further, in the above-described embodiment, even if the operation of the data reader 36 and the processing of the data corrector 36b utilize the interrupt function of the microprocessor, the same effect as that of the above-described embodiment is obtained.

Further, in the above embodiment, even if the filter 35 is a sample holder operated by an interrupt signal, the same effect as that of the above embodiment can be obtained.

〔The invention's effect〕

As described above, according to the present invention, the table of the phase difference angle θ in which the vector locus of the terminal voltage changes in parallel to the no-load terminal voltage in the d-axis direction is used, depending on the basic component of the armature current. The vector calculation is performed, the no-load saturation curve is used for the calculation of the magnetizing current, the phase angle of the second converter is corrected by the DC voltage control, and the field current is corrected by the terminal voltage control. In addition, since the terminal voltage is detected at each commutation start point and the fluctuation of the commutation lead angle γ is corrected, the accuracy of the device can be improved and a stable commutation operation can be performed. There is an effect that can be obtained.

[Brief description of drawings]

FIG. 1 is a block diagram showing a control device for an AC electric motor according to an embodiment of the present invention, FIG. 2 is a detailed block diagram of a vector calculator in FIG. 1, and FIG. 3 is a diagram for explaining the operating principle of the present invention. 4 is a vector diagram of FIG. 4, FIG. 4 is a characteristic diagram of the θ arithmetic circuit, FIG. 5 is a characteristic diagram showing a no-load saturation curve, FIG. 6 is a waveform diagram of armature current, and FIG. 1 is a block diagram of FIG. 1, FIG. 8 is a waveform diagram of each part of FIG. 7, FIG. 9 is a terminal voltage waveform diagram, FIG. 10 is a terminal voltage characteristic diagram, FIG. 11 is a configuration diagram of a conventional device, and FIG. Is a vector diagram showing the relationship between the voltage and current of the electric motor, FIG. 13 is a vector diagram for explaining the operation of the device shown in FIG. 1, and FIG. 14 is a voltage waveform diagram of the thyristor. 1 is a first power converter, 2 is a second power converter, 3 is an AC motor (synchronous motor), 4 is a position detector, 18 is a power factor angle command circuit, 19 is a no-load terminal voltage command circuit, 20 is a vector calculator, 21 is a DC voltage command circuit, 22 is a DC voltage control circuit (DC voltage deviation amplifier), 26 is a terminal voltage command circuit, 27 is a terminal voltage control circuit (terminal voltage deviation amplifier), and 201 is a phase difference angle. Calculation table, 202 is a terminal voltage calculator, 203 is a no-load saturation curve table, and 204 is d
Axis component magnetizing current calculator, 205 is a q-axis armature reaction voltage calculator, 206 is a field current calculator, 207 is a field current command generating circuit (adder), 208 is a commutation overlap angle calculator,
209 is a phase command generation circuit (adder), 35 is a filter, 36 is a data reader, and 341 is an interrupt signal generator. In the drawings, the same reference numerals indicate the same or corresponding parts.

Claims (3)

[Claims] 1. A power converter for performing frequency conversion of AC, an AC motor driven by an output of the power converter, a position detector for outputting a position signal according to a rotational position of the AC motor, and DC voltage detecting means for detecting the DC voltage of the power converter, terminal voltage detecting means for detecting the terminal voltage of the AC motor, field current control circuit for controlling the field current of the AC motor, and the AC motor A no-load terminal voltage command circuit that sets the magnitude of the no-load terminal voltage, a power factor angle command circuit that commands a power factor angle of the AC motor, a command signal of the no-load terminal voltage command circuit, and the power factor A vector calculator that outputs a field current command of the AC motor and a phase command of the power converter according to the magnitude of the armature current of the AC motor based on a command signal of an angle command circuit; and the armature current. Previous A DC voltage command circuit for calculating a DC voltage by the output signal of the terminal voltage detection means and the signal of the sum of the commutation overlap angle and the power factor angle of the vector calculator, and the output signal of the DC voltage command circuit and the DC voltage. A DC voltage control circuit for controlling a phase of the power converter by adding a signal obtained by amplifying a deviation from an output signal of the voltage detecting means to a phase command of the vector calculator, a terminal voltage signal of the vector calculator, and an electric machine. A terminal voltage command circuit for calculating the terminal voltage by the child current, and a signal obtained by amplifying the deviation between the output signal of the terminal voltage command circuit and the output signal of the terminal voltage detection means is used as the field current signal of the vector calculator. A terminal voltage control circuit that adds and gives a field current command to the field current control circuit, and a data reader that reads the terminal voltage of the AC motor at a commutation start point of the power converter. The control device of the example was AC motor. 2. The control device for an AC electric motor according to claim 1, further comprising a data corrector for correcting the output of the data reader by changing the commutation lead angle. 3. A d-axis armature reaction for performing a vector operation such that a vector locus of a terminal voltage of the AC motor changes in a vertical direction with respect to the no-load terminal voltage according to a magnitude of an armature current. A phase difference angle calculation table for calculating a phase difference angle by inputting a per unit value of voltage, a terminal voltage calculator for calculating a terminal voltage from the phase difference angle and the no-load terminal voltage signal, and an alternating current for calculating a magnetizing current from the terminal voltage signal A no-load saturation curve table of the electric motor, a d-axis component magnetizing current calculator for obtaining a d-axis component of the magnetizing current from the phase difference angle, and a q-axis armature reaction voltage from the phase difference angle, the power factor angle and the armature current. A desired q-axis armature reaction voltage calculator, a field current calculator for compensating the q-axis armature reaction voltage component to obtain a field current component for canceling, and an armature reaction compensator. A field current command generation circuit that adds the field current signal and the d-axis component magnetizing current to generate the field current command, and a commutation overlap angle based on the terminal voltage signal, the armature current signal, and the power factor angle. A commutation overlap angle calculator and a phase command generation circuit that adds a commutation overlap angle signal, a power factor angle, and a phase difference angle to generate a phase command for the power converter,
The control device for an AC motor according to claim (1), characterized in that a vector arithmetic unit is configured.