JPH0641970B2 - Method for measuring ferromagnetic resonance absorption - Google Patents

Description

TECHNICAL FIELD The present invention relates to a method for measuring ferromagnetic resonance absorption of a ferromagnetic material having a small loss in microwaves.

[Conventional technology]

When measuring the ferromagnetic resonance absorption of a ferromagnetic material with a small loss in the microwave, it is roughly divided into (1) the resonance method and
(2) There is a non-resonant method. The former resonance method refers to the case where the component loaded with the sample is a microwave resonator. This is used when the ferromagnetic resonance half-value width ΔH is relatively large or when the sample size is small, that is, when the resonance signal as a whole is small. As the measuring device, a device for measuring the reflected wave from the cavity resonator 3 as shown in FIG. 14 is used.

The microwave power changes its direction from the microwave oscillator 1 via the waveguide 8a by the circulator 7 and enters the cavity resonator 3 as the input power Pi. A sample 4 is placed inside the cavity resonator 3. As another method, the sample 4 is disposed from the outside in the vicinity of the coupling hole provided on the side wall of the resonator. When the ferromagnetic resonance does not occur, the cavity resonator 3 and the measurement system are in a state of substantially critical coupling, the reflected power Pr is very small, and Pr << Pi holds. At this time, the output of the microwave detector 2 becomes almost zero. When the static magnetic field necessary for the space where the cavity resonator is arranged is generated by the pole pieces 5a and 5b of the electromagnet to cause the ferromagnetic resonance, the reflected power Pr from the cavity resonator 3 slightly increases. The microwave detector 2 detects this as a signal. In practice, the magnetic field modulation coil 6 causes the electrostatic field to fluctuate slightly at high frequencies, and the differential waveform is observed as the change. The precondition for this configuration is that Pr << Pi holds in the range where the perturbation theory holds, that is, in the ferromagnetic resonance state. Therefore, this method is not suitable when the ferromagnetic resonance signals of Pr to Pi are large.

On the other hand, in the non-resonant method, the microwave component loaded with the sample is the short-circuit end of the normal short-circuit line 9 as shown in FIG. This is used when the ferromagnetic resonance half-value width ΔH is relatively small or when the sample size is large, that is, when the resonance signal as a whole is large. However, the reflected power Pr in the state where the ferromagnetic resonance does not occur is almost equal to the incident power Pi. As the measuring device, the one having the structure shown in FIG. 15 may be used. However, normal ESR (Electron Spin Resona
nce) is not configured to measure large reflected waves, it is impossible to use this method as it is. Further, the coupling between the sample and the microwave is considerably smaller than that of the resonance method, but in the case of the short-circuit line 9, the coupling is still too large in the sample with a small loss because a standing wave is generated.

[Problems to be solved by the invention]

When the ferromagnetic resonance half-value width ΔH is relatively small or when the sample size is large, that is, when the overall resonance signal is large,
As mentioned above, the non-resonant method using the short-circuit line is a normal ES.
There was a problem that it was not circuit-compatible with the R device.

The present inventor has already filed a patent application 6 as a method for solving this problem.
No. 0-175067 proposed a method of a straight waveguide with a matched load. Although this method is a non-resonant method,
Fits very well with However, this method has a problem that the real part and the imaginary part of the magnetic susceptibility are always measured in a mixed form. Further, there is a phenomenon in which the detection sensitivity is lowered because it is necessary to lower the binding degree in order to increase the measurement accuracy of ΔH.

An object of the present invention is to provide a highly sensitive method of separately measuring an imaginary part and a real part of magnetic susceptibility by using a direct waveguide method with a matched load as a non-resonant method suitable for an ordinary ESR device. Is.

[Means for solving problems]

The present invention provides a matching load at an end of a microwave transmission line, a coupling portion at a side portion of the microwave transmission line, and a ferromagnetic sample to be measured is coupled to the coupling portion in a microwave manner. In the method of applying a static magnetic field from the outside for ferromagnetic resonance, injecting microwave power from the input part of the microwave transmission line, and measuring the reflected wave, a part of the output of the microwave oscillator is branched. As a standard wave, the reflected wave and the standard wave are caused to interfere with each other, and the interference wave is rectified into an output signal, and the phase of the standard wave is changed,
A method for measuring ferromagnetic resonance absorption, characterized in that the real part and the imaginary part of the microwave susceptibility of the output signal are obtained separately.

[Theoretical Support of the Present Invention]

The measuring apparatus of the present invention uses the detection method shown in FIG. A small reflected from the sample reflected wave _{er e r = A · cos (} ωt + φ r) (1) and micro standard wave guided branched by the directional coupler the part of the wave oscillator signal e _s e _s = B · cos (ωt + φ s) (2) both are synthesized _{e t} next, it is rectified by a detector 2.
Because in normal operating conditions it is _e r _{«e s,} i.e. A
<< B holds.

_{e t = e r + e s} = cosωt (Acosφ r + Bcosφ s) -sinωt (Asinφ r + Bsinφ s) = A'cosωt-B'sinωt (3a) _{However, A '= Acosφ r + Bcosφ} s (3b) B' = Asinφ _r + Bsinφ _s (3c).

In the case of half-wave rectification by the detector 2 such as a diode, half of the effective value is actually detected, so the output signal is Becomes Substituting equations (3a), (3b), and (3c) into equation (4), and calculating A << B To get This output signal is a function of the amplitude of the reflected wave e _r and the phase difference between the standard wave and the reflected wave.

In equation (5), A and φ _r contain the information about the ferromagnetic resonance of the sample. The output signal of a normal ESR is magnetic field modulated. Therefore, differentiating equation (5) with the magnetic field Becomes Here, σ is a dimensionless number proportional to the external magnetic field and is σ = rHo / ω as shown in the equation (10). The transformation of Eq. (6) is as follows.

When the phase φ _{s of the} standard signal changes variously, the output signal also changes according to the above equation. As a special case, the following two calculation results are shown.

1) φ _s = 0. In this case, the second term of equation (7) becomes zero.

2) φ _s = π / 2. In this case, the first term of equation (7) becomes zero.

δv / δσ∝ (δA / δσ) sinφr + Acosφr (δφr / δσ) (8
b) However, the derivative of the second item of the above equation is Is also represented.

In the present invention, the coupling state between the sample to be measured and the microwave transmission line is represented by the equivalent circuit in FIG. Zo is the characteristic impedance of the transmission line. rZo is the deviation from the characteristic impedance of the real part of the matched load. The dotted LC resonant circuit represents the reactance portion of the matched load. This LC resonant circuit can be ignored if the measurement frequency is selected as this resonant frequency. Circuit constants in blocks connected in series represent loaded ferromagnetic samples. β is a constant indicating the coupling state between the line and the sample.

As shown in FIG. 3, this equivalent circuit has a real part x ′ (dotted line) and an imaginary part x ″ (solid line) of the magnetic susceptibility x at an effective magnetic field Ho = 0, Ho = ∞ of ferromagnetic resonance. Both are considered to be zero.

Here, the real part and the imaginary part of the complex magnetic susceptibility x in the vicinity of the ferromagnetic resonance are respectively expressed by the following equations (reference: Konishi, "Recent Micro Technology Using Ferrites", The Institute of Electronics and Communication Engineers, Showa 40) Year pp.10).

However, ω _m = γ4πMs, ωi = γHo, α is Gilber
This is a t-type relaxation constant. ω is the angular frequency of the microwave, 4
πMs is the saturation magnetization of the film, Ho is the effective magnetic field obtained from Kittel's resonance condition, and γ is the gyromagnetic ratio.

At this time, the impedance Z seen from the input end at the above-mentioned specific frequency is expressed by Z = Zo (1 + r + βx ″ + jβx ′) (11). The reflection coefficient Γ at this time is Becomes

When this is divided into a real part and an imaginary part and expressed as an index, Eq. (12) can be written as Γ = Aexp (jφ _r ) (13). Further, if the amplitude of the incident wave is 1, the equation (13) is also an index representation of the reflected wave e _r = Acos (ωt + φ _r ). Here, if a = r + βx ″ and b = βx ′ Is.

<When output signal is very small> Here, as a special case, the state where the coupling is extremely weak (a, b
<< 1) and off-resonance matching is good (r
Consider << 1). Under these conditions (14)-(17)
The formula becomes simple Becomes

Using this, Eqs. (8a) and (8b) are calculated.

1) When φ _s = 0, the following is obtained.

The signal in this case is only related to the imaginary part of the magnetic susceptibility.

2) When φ _s = π / 2, the following is obtained.

The signal in this case is only related to the real part of the susceptibility.

When the output signal is very small as described above, the real part and the imaginary part of the magnetic susceptibility can be separately measured by changing the phase of the standard signal. FIG. 4 is an external magnetic field dependence diagram of the output signal calculated when ω _m = 1 and α = 0.001. φ _s =
When it is 0, it is a differential curve of x ″, but when φ _s = π / 2, it is a differential curve of x ′. ΔH can be calculated from each singular point.

That is, when φ _s = 0, the true ΔH is calculated from the magnetic field interval ΔH ′ between the peak and the valley of the curve. Becomes

When φ _s = π / 2, the magnetic field interval ΔH ″ at which the curve crosses zero and the true ΔH are the same as ΔH = ΔH ″ (21).

<General treatment> When the bond is strong, the equations (14) to (17) must be numerically calculated as they are.

1) When φ _s = 0, the output signal is as follows.

2) When φ _s = π / 2, the output signal is as follows.

FIG. 5 is an external magnetic field dependence diagram of the output signal obtained by numerical calculation of the equation (22) using the coupling coefficient β and the residual resistance r as parameters.
The curves are point symmetric, so only half are shown. Coupling coefficient β
It can be seen that the apparent ΔH 'gradually increases with increasing. The presence of residual resistance r is ΔH
Has almost no effect on the measured value of. FIG. 6 shows a curve when the reflected power calculated from the coupling coefficient β is taken on the horizontal axis and the correction coefficient of ΔH is taken on the vertical axis. It can be seen that the correction coefficient rapidly increases when the reflected power ratio becomes 10 or more.

〔Example〕

FIG. 1 is a block diagram of a measuring apparatus showing an embodiment for realizing the principle equivalent circuit of FIG. 2 of the present invention. The direct waveguide 10 serves as a measuring arm, and the pole pieces 5a and 5b of the electromagnet are used.
Enter between The matching load 11 is connected to the end of the direct waveguide 10. The sample 4 is arranged in a portion where the magnetic field distribution is as uniform as possible, and in the drawing of this embodiment, the sample is placed outside the waveguide through the coupling hole. Of course, the sample 4 may be inside the waveguide.

Further, the standard wave splits a part of the output of the microwave oscillator, and is mixed with the reflected wave in the detection unit 2 via the phase shifter 13.

As can be seen from the block diagram of the principle of the present invention in FIG. 1, when the ferromagnetic resonance does not occur, the incident power Pi is consumed by the matching load 11 as it is, and the reflected power Pr is extremely small. That is, Pr << Pi holds. Pole pieces 5a, 5b
When a magnetic field is generated by this and the sample 4 is brought into a ferromagnetic resonance state with the microwave, the impedance of the direct waveguide changes to generate a reflected wave, and a signal appears at the output of the detector. Figure 7 shows
The actual dimensions of the straight waveguide and the matching load used in the embodiment of FIG. 1 are shown. A coupling hole of about 6 mmφ is opened in the H surface of the waveguide.

As a method of holding the sample, it is possible to attach the sample to the jig and insert the sample into the straight waveguide through the coupling hole.
Alternatively, as another method, as shown in FIG.
It is also conceivable that a is brought close to or in contact with the coupling hole 12 from the outside. In a general process, it is necessary to examine the physical properties of the wafer with the LPE film as soon as possible without processing the wafer, so the method of FIG. 8 is practical.

Further, as another example, an experiment was conducted in which a similar coupling hole was formed on the E-face of the waveguide, but it was possible to find a large difference between them in comparison with the case of the H-face of the above-mentioned embodiment. could not.

The ferromagnetic resonance absorption curve was measured by this method using a Ga-substituted YIG sphere (4πMs = 400G) of about 1 mmφ. The dimension of the coupling hole of the straight waveguide is 6 mmφ, and the spherical sample 4b is placed on the slide glass 16 with the scotch tape 15 as shown in FIG.
It was pasted with a. FIG. 10 shows the measurement result. The main resonance occurs at about 3170 [Oe]. Other absorption is about 25
Only a slight signal was observed on the low magnetic field side of [Oe], and nothing else was observed.

The calculation result of the detected waveform when φ _s is changed is described in FIG. 4 described above. To confirm this, a similar experiment was conducted using this Ga-substituted YIG sphere. FIG. 11 shows the result. The phase change amount of the phase shifter could not be measured numerically due to the device, but the number of rotations of the knob of the phase shifter is shown in the figure. -1, -2, -3, and -4 are the numbers rotated counterclockwise from the reference state, respectively. As can be seen by comparing FIGS. 4 and 12, the changes in the two are very similar qualitatively. In particular, the measured ΔH ′ and ΔH ″ are , Which is almost the same as the theoretical prediction. This confirms that the computational model of the present invention is quite correct.

FIG. 12 shows a 1 inch diameter GGG (Gadolunium Gal).
Lium Garnet) Approximately 20 μm fabricated on a wafer
Thickness of YIG (Yttrium Iron Garnet) LPE (Liq
uid Phase Epitoxial) Data obtained by contacting a thick film from the outside of the coupling hole as shown in FIG. 8 at a frequency of 9.03 GHz. The diameter of the coupling hole was 6 mmφ. From this figure, the apparent ΔH 'is measured as 0.42 [Oe], and the true ΔH is It becomes 0.73 [Oe]. Also, the ferromagnetic resonance magnetic field is 5000
Since it is [Oe], 4πMs = 1756 [Oe] is obtained by calculating 4πMs from the Kittel resonance conditional expression ω = γ (Hext−4πMs). This value is almost equal to the other literature value 1800G as the saturation magnetization of YIG.

FIG. 13 shows another embodiment of the present invention in which a coaxial line is used as the microwave transmission line. It is a transmission line suitable for measurement in the frequency band below about 2 GHz.

Although only the waveguide and the coaxial line are shown as examples of the microwave transmission line, the effect of the present invention is also effective in other microwave transmission lines such as a microstrip line and a dielectric waveguide. It can be easily understood by a private house related to the present technology.

〔The invention's effect〕

According to the measuring method of the present invention, it is possible to relatively easily measure the ferromagnetic resonance absorption of a sample of low ΔH or a sample of large size, which has been difficult to measure conventionally. Also, by changing the phase of the standard wave, the imaginary part and the real part of the microwave susceptibility of the ferromagnetic material can be measured separately.

[Brief description of drawings]

FIG. 1 is a block diagram of a combination of a detection method and a direct waveguide method with a matching load, which are the basis of the present invention, FIG. 2 is an equivalent circuit diagram of the non-resonant method used in the present invention, and FIG. External magnetic field dependence diagrams of the real and imaginary parts of the ratio, FIGS. 4 and 5 are external magnetic field dependence diagrams of the output signal according to the present invention, and FIG. 6 is a relationship diagram of the reflected power ratio and the correction coefficient of ΔH, FIG. 7 is an assembly view of a direct waveguide with a matching load showing an embodiment of the present invention, FIGS. 8 and 9 are attachment views of a sample according to an embodiment of the present invention,
10, 11 and 12 are external magnetic field dependence diagrams of detection signals according to the present invention, FIG. 13 is a layout diagram of a transmission line and a sample showing another embodiment of the present invention, and FIG. 14 is a conventional technique. FIG. 15 is a block diagram of a measuring device using the resonance method according to FIG. 15, and FIG. 15 is a block diagram of a measuring device using the non-resonance method according to the related art.

Claims (1)

[Claims] 1. A matching load is provided at an end of a microwave transmission line, a coupling portion is provided at a side portion of the microwave transmission line, and a sample of a ferromagnetic material to be measured is microwave-coupled to the coupling portion. In a microwave device that applies a static magnetic field from the outside for ferromagnetic resonance, injects microwave power from the input portion of the microwave transmission line, and measures the reflected wave, a microwave device that generates microwave power is used. A part of the output of the wave oscillator is branched into a standard wave, the reflected wave and the standard wave are caused to interfere with each other, the interference wave is rectified into an output signal, and the phase of the standard wave is changed to output the output. A method for measuring ferromagnetic resonance absorption, characterized in that a real part and an imaginary part of a microwave magnetic susceptibility are separately obtained from a signal.