JPH0648905B2 - Series resonant converter - Google Patents

Description

Description: TECHNICAL FIELD The present invention relates to a series resonant converter capable of reducing loss and switching noise in a switching element and stably operating even in an overload such as a load short circuit.

[Conventional technology]

A DC-DC converter is applied to a circuit for converting a DC voltage into a DC voltage having a different polarity or a different voltage value, aiming at downsizing, weight reduction, and high efficiency. DC
-In the DC converter, by increasing the conversion frequency,
It is possible to reduce the size and weight of transformers and output filters.

Currently, the most frequently used circuit type of the DC-DC converter is a forward type or flyback type in which the current flowing through the switching element is forcibly cut off by turning off the switching element. These circuit types have the following problems when increasing the frequency.

(1) When the switching element is turned on, a spike-shaped current flows due to the recovery current of the rectifying diode, resulting in large loss and noise at turn-on.

(2) The load at the turn-off of the switching element is inductive due to the leakage inductance of the transformer and the inductance component distributed in the circuit, so the loss in the switching element and snubber circuit (spike voltage absorption circuit) is large.

Due to these problems, at present, even if a MOSFET is used as a switching element up to an output capacity of about 100 W, a conversion frequency is about 500 kHz and 100 kHz for a power of 1 kW or more.
The limit is about z. Furthermore, in order to increase the conversion frequency, there are many items to be considered, such as improving the characteristics of switching elements and rectifying elements, manufacturing transformers with less leakage inductance, mounting components with less stray inductance, and reducing the loss of snubber circuits. Therefore, a DC-DC converter that can reduce the loss in the switching element by making the waveform of the current flowing through the switching element sinusoidal by the capacitor and the inductor has been actively studied. The circuit type of this DC-DC converter is called a series resonance converter. FIG. 2 shows a circuit configuration of a series resonance converter which is generally considered. A switching element (for example, a MOS-F
ET) 2 and 3 series circuits, feedback diodes 4 and 5 series circuits and resonance capacitors 11 and 12 series circuits are connected, and the connection points of the switching elements 2 and 3 and the feedback diodes 4 and 5 are connected. One end of the primary winding of the transformer 6 is connected to the connection point, the other end is connected to the connection point of the resonance capacitors 11 and 12 through the resonance inductor 13, and both ends of the secondary winding of the transformer 6 are rectifying diodes. A smoothing capacitor 9 and a load resistor 10 are connected between the respective ends of the rectifying diodes 7 and 8 and the middle points of the secondary windings of the transformer 6, which are connected to the respective ends of the rectifying diodes 7 and 8. Next, the operation will be described. When the switching element 2 is turned on, the DC power supply 1-the switching element 2-the primary winding of the transformer 6-
Resonance inductor 13-Resonance capacitor 12-DC power supply 1 loop, resonance capacitor 11-Switching element 2-Primary winding of transformer 6-Resonance inductor 13-
A resonance current flows in the loop of the resonance capacitor 11. DC power supply 1-when switching element 3 is turned on
Resonance capacitor 11-Resonance inductor 13-Transformer 6 primary winding-Switching element 3-DC power supply 1 loop, resonance capacitor 12-Resonance inductor 13
A resonance current flows in the loop of the primary winding of the transformer 6, the switching element 3 and the resonance capacitor 12. Transformer 6
The ratio of the number of primary windings (N ₁ ) to the number of secondary windings (N ₂ ) is n
If (= N ₁ / N ₂ ), n times the resonance current flows in the secondary winding of the transformer 6, so this resonance current is rectified by the rectification diodes 7 and 8 and the smoothing capacitor 9 is set. It is charged and a DC voltage is supplied to the load resistor 10.

Assuming that the voltage of the DC power supply 1 is E, if the charging voltage of the resonance capacitor is 3E / 2 or less, no current flows in the feedback diode 4 or 5. This area will be called mode I.

The resonance current i ₀ flowing through the switching element 2 or 3 in the mode I is given by the equation (1) when t = 0 when the switching element is turned on.

i ₀ = [I ₀ sin [t / {2C ₀ (L ₁ + L ₂ )} ^1/2 ]] / 4nf ₁ {2C ₀ (L ₁ + L ₂ )} ^1/2 (1) where I ₀ Is the current flowing through the load resistance 10, L ₁ is the leakage inductance of the transformer 6, L ₂ is the inductance of the resonance inductor 13, and C ₀ is the resonance capacitor 1
The capacitance of ₁ , 12 and f ₁ are the operating frequencies.

In mode I, when the load resistance 10 is reduced and the flowing current I ₀ is increased, the peak value of the resonance current i ₀ is also increased, as can be seen from the equation (1). As a result, the voltage (output voltage) of the load resistor 10 becomes constant and given by E / (2n). Since E and n cannot be changed freely, constant voltage D
It is not used in this mode as a C-DC converter.

The charging voltage of the resonance capacitors 11 and 12 is the load resistance 10
Rises as it becomes smaller, and becomes 3E / 2 or more, the resonance capacitor 12-the resonance inductor 13-the primary winding of the transformer 6-the feedback diode 4-the DC power supply 1-the resonance capacitor 12 loop or the resonance capacitor 11
A feedback current starts to flow in the loop of the DC power supply 1-the feedback diode 5-the primary winding of the transformer 6-the resonance inductor 13-the resonance capacitor 11. This basin will be called Mode II. In mode II, the output current is constant even if the load resistance 10 is changed, and if the resonance frequency is f ₀ , then f ₀
In the case of> 2f ₁ , it is given by 8f ₁ C ₀ nE. The output voltage is given by 8f ₁ C ₀ nER when the resistance value of the load resistor 10 is R, and is obtained by changing f ₁ to keep the output voltage constant. Therefore, in this mode as a DC-DC converter, use. The resonance current i ₀ flowing through the switching element 2 or 3 in the mode II is given by the equation (2).

i ₀ = [{nV ₀ + (E / 2)} sin [t / {2C ₀ (L ₁ + L ₂ )} ^1/2 ]] / {(L ₁ + L ₂ ) / 2C ₀ } ^1/2 (2) The feedback current i ₁ flowing through the feedback diode 4 or 5 is i
_{If the} time at which ₁ starts to flow is T ₀ , equation (3) is obtained.

i ₁ =-[{-nV ₀ + (E / 2)} sin [(tT ₀ / {2C ₀ (L ₁ + L ₂ )} ^1/2 ]] / {(L ₁ + L ₂ ) / 2C ₀ } ^1/2 (3) Here, V ₀ is the output voltage (voltage of the load resistor 10).
FIG. 3 shows current waveforms flowing through the switching element 2 or 3 and the feedback diode 4 or 5 in the mode I and the mode II.

In mode II, when the load resistance 10 becomes smaller, nV ₀ also becomes smaller. Therefore, the peak value of the resonance current decreases from the formula (2), and the formula (3)
Therefore, the peak value of the feedback current increases and the sum of the average values of i ₀ and i ₁ becomes constant, so that the output current becomes constant. That is,
If the operating frequency f ₁ is fixed, good voltage drooping characteristics can be obtained. Further, since the leakage inductance L ₁ of the transformer 6 is in series with the resonance inductance L ₂ , it does not adversely affect the circuit operation.

However, in this series resonance converter, as the resonance frequency f ₀ is increased, the reverse blocking capability recovery time of the feedback diode becomes a problem. That is, if the switching element turns on before the reverse blocking capability of the feedback diode is restored, a mode in which the DC power supply is short-circuited occurs,
There is a problem that loss and noise increase, and the switching element may be destroyed in some cases. Further, when this series resonant converter is used at f ₀ <2f ₁ , the switching element is turned on when the feedback current flows through the primary winding of the transformer and the feedback diode. In addition to a rising current, a large spike-shaped current flows until the reverse blocking function of the feedback diode is restored, causing problems such as increased loss and noise, and increased stress on the switching element. Occur. For these reasons, in the conventional series resonance converter, the operating frequency f ₁ must be used at ½ or less of the resonance frequency f ₀ , and there are problems that the output capacitance is restricted and the smoothing capacitor becomes large.

In order to solve this problem, by connecting a feedback diode so that the feedback current can be fed back to the DC power supply without passing through the primary winding of the transformer, the current of the switching element rises even when used with f ₀ <2f _1. There is proposed a series resonant converter in which the spike current does not flow even when the feedback diode does not have the reverse blocking function restored. In this circuit, feedback diodes 4 and 5 are connected to the connection point between the primary winding of the transformer 6 and the resonance inductor 13 as shown in FIG. _In the case of ₀ <2f ₁ , a spike-shaped current does not flow in the switching element even if the reverse blocking function of the feedback diode is not restored.

Next, the operation of the circuit will be described. When the switching element 2 is turned on, the DC power supply 1-the switching element 2-the primary winding of the transformer 6, the resonance inductor 13-the resonance capacitor 12-the loop of the DC power supply 1 and the resonance capacitor 11-the switching element 2-transformer. 6 primary winding-
A resonance power supply flows in a loop of the resonance inductor 13 and the resonance capacitor 11. When the switching element 3 is turned on, the DC power supply 1-the resonance capacitor 11-the resonance inductor 13-the primary winding of the transformer 6-the switching element 3-the loop of the DC power supply 1 and the resonance capacitor 1
A resonance current flows in a loop of 2-resonance inductor 13-transformer 6 primary winding-switching element 3-resonance capacitor 12. If the ratio of the number of primary windings to the number of secondary windings of the transformer 6 is n, n times the resonance current flows in the secondary windings, so this resonance current is rectified by the rectifying diodes 7 and 8 and smoothed. The charging capacitor 9 is charged and a DC voltage is supplied to the load resistor 10. Assuming that the voltage of the DC power supply 1 is E, if the charging voltage of the resonance capacitor is E or less, no current flows through the feedback diode 4 or 5. This area will be called mode I.

The resonance current i ₀ flowing through the switching element 2 or 3 in the mode I is given by the equation (1), where t = 0 is the time when the switching element is turned.

In mode I, as can be seen from equation (1), when the load resistance 10 is reduced and the flowing current is increased, the peak value of the resonance current increases. As a result, the output voltage becomes constant and E /
It is given by (2n).

The charging voltage of the resonance capacitor rises as the load resistance 10 is reduced, and when it reaches E, the resonance capacitor 11-
DC power supply 1-feedback diode 5-resonance inductor 1
3-loop of resonance capacitor 11 or resonance capacitor 12-resonance inductor 13-feedback diode 4
A feedback current begins to flow in the loop of the DC power supply 1-resonance capacitor 12. This area is called Mode II.
In mode II, when the output current is constant even if the load resistance 10 is changed and f ₀ > 2f ₁ , it is given by 4f ₁ C ₀ nE.

The resonance current i ₀ flowing through the switching element 2 or 3 in the mode II is given by the equation (4).

i ₀ = [Esin [t / {2C ₀ (L ₁ + L ₂ )} ^1/2 ]] / 2 {(L ₁ + L ₂ ) / 2C ₀ } ^1/2 (4) Feedback diode 4 or 5 The feedback current i ₁ flowing through
_{If the} time when ₁ starts to flow is T ₀ , the equation (5) is obtained.

i ₁ ＝-[{-nV ₀ + (E / 2)} sin {(tT ₀ ) / (2C ₀ L ₂ ) ^1/2 }] / (L ₂ / 2C ₀ ) ^1/2 (5) Mode II Then, when the load resistance 10 becomes smaller, the peak value of the feedback current increases from the equation (5), but the peak value of the resonance current becomes constant from the equation (4). The output current is n of the average value of i ₀
It is constant because it is doubled. That is, the output voltage decreases.

FIG. 5 shows current waveforms flowing through the switching element 2 or 3 and the feedback diode 4 or 5 in the mode I and the mode II.

[Problems to be Solved by the Invention]

The circuit of FIG. 4 has many advantages as described above,
In case of overload such as load short circuit, transformer 6
Due to the leakage inductance L ₁ of, there are the following problems. The voltage V ₁ generated in the leakage inductor of the transformer 6 is given by the equations (4) to (6).

V ₁ = L ₁ (di ₀ / dt) (6) The voltage represented by the equation (6) is the voltage n of the primary winding of the transformer 6.
When nV ₀ <V ₁ (7) with respect to V ₀ , the primary winding of the transformer 6-the feedback diode 4
A circulating current flows in the loop of the switching element 2-the primary winding of the transformer 6 or the primary winding of the transformer 6-the switching element 3-the feedback diode 5-the loop of the primary winding of the transformer 6. While the circulating current is flowing, it is necessary to keep the switching element in the ON state in order to prevent the current from being forcibly cut off from the viewpoint of surge voltage and noise. The current i ₂ flowing through the primary winding of the transformer 6 is attenuated according to the following equation if I is the primary winding current of the transformer 6 when the circulating current flows out.

i ₂ = I- (nV ₀ t / L ₁ ) (8) When the output is short-circuited and V ₀ = 0, i ₂ = I and i ₂ is not attenuated.

FIG. 6 shows i ₀ , i ₁ , i ₂ when nV ₀ <V ₁ .
Shows the waveform of. No problem if t ₂ <t ₃ , but t ₂ >
When it becomes t ₃ , there is a problem that a phenomenon that the resonance current flows out again occurs, the operation becomes unstable, and a good voltage drooping characteristic cannot be obtained.

An object of the present invention is to speed up the decay of the circulating current and lengthen the period during which the feedback current is flowing, thereby eliminating the adverse effect of the leakage inductance of the transformer 6 and ensuring stable operation and series resonance with good voltage drooping characteristics. To provide a converter.

[Means for Solving the Problems]

According to the present invention, a second resonance inductor is connected between the connection point of two feedback diodes and the connection point of the primary winding of the transformer and the resonance inductor, so that even if an overload such as a load short circuit occurs, the series connection is achieved. The most important feature is that the resonant converter can be operated stably.

〔Example〕

Examples of the present invention will be described below. FIG. 1 is a connection diagram of an embodiment of the present invention, in which parts corresponding to those in FIG. 4 are designated by the same reference numerals. In the present invention, the resonance inductor 14 is connected between the connection point of the feedback diodes 4 and 5 and the connection point of the primary winding of the transformer 6 and the resonance inductor 13.

Next, the operation of the circuit will be described. When the switching element 2 is turned on, the DC power supply 1-the switching element 2-
Primary winding of transformer 6-Resonance inductor 13-Resonance capacitor 12-DC power supply 1 loop and resonance capacitor 11-Switching element 2-Primary winding of transformer 6-Resonance inductor 13-Resonance capacitor 11 Resonant current flows in the loop. When the switching element 3 is turned on, the DC power supply 1-the resonance capacitor 11-the resonance inductor 13-the primary winding of the transformer 6-the switching element 3-the loop of the DC power supply 1 and the resonance capacitor 1
A resonance current flows in a loop of 2-resonance inductor 13-transformer 6 primary winding-switching element 3-resonance capacitor 12. If the ratio of the number of primary windings to the number of secondary windings of the transformer 6 is n, n times the resonance current flows in the secondary windings, so this resonance current is rectified by the rectifying diodes 7 and 8 and smoothed. Charge the capacitor 9 and apply DC voltage to the load resistor 1
Supply to 0. When the voltage of the DC power supply 1 is E, if the charging voltage of the resonance capacitor is E or less, no current flows in the feedback diode 4 or 5 and the resonance inductor 14. This area is the same as the mode I of the operation of FIG.
The resonance current flowing through the switching element 2 or 3 is calculated by the formula (1)
Given in. In mode I, when the load resistance 10 is reduced as can be seen from the equation (4), the peak value of the resonance current increases and the output voltage becomes constant, which is given by E / (2n).

The charging voltage of the resonance capacitors 11 and 12 is the load resistance 10
Rises as the value becomes smaller and becomes equal to or higher than E.-Resonance capacitor 11-DC power supply 1-Feedback diode 5-Resonance inductor 14-Resonance inductor 13-Resonance capacitor 11 loop or resonance capacitor 12-Resonance capacitor A feedback current starts to flow in the loop of the inductor 13-resonance inductor 14-feedback diode 4-DC power supply 1-resonance capacitor 12. This area is called Mode II. In mode II, the output current is constant even if the load resistance 10 is changed, and when f ₀ > 2f ₁ , it is given by 4f ₁ C ₀ nE. The resonance current i ₀ flowing through the switching element 2 or 3 in the mode II is given by (4).

The feedback current i ₁ flowing through the feedback diode 4 or 5 is i
Is given by Eq. (9), where T ₀ is the starting time.

i ₁ = [{-nV ₀ + (E / 2)} sin [(tT ₀ ) / {2C ₀ (L ₂ + L ₃ )} ^1/2 ]] / {(L ₂ + L ₃ ) / 2C ₀ ) } ^1/2 (9) Here, L ₃ is the inductance of the resonance inductor 14.

The resonance half cycle of the feedback current in Eq. (5) is π (2C ₀ L ₂ ) ^1/2 , but the resonance half cycle of the feedback current in Eq. (9) is π (2C ₀ (L ₂ +
Since L ₃ } ^1/2 , it can be easily determined that the circuit according to the present invention is longer than the conventional circuit during the period in which the feedback current is flowing.

In mode II, when the load resistance 10 becomes smaller, the peak value of the feedback current increases from the equation (5), but the peak value of the resonance current becomes constant from the equation (4). The output current is n of the average value of i ₀
It is constant because it is doubled. That is, the output voltage decreases.

When the load resistance 10 becomes small and the condition expressed by the expression (7) is satisfied, the primary winding of the transformer 6-resonance inductor 14-feedback diode 4-switching element 2-the primary winding loop of the transformer 6 or the transformer 6 Primary winding-switching element 3-feedback diode 5-resonance inductor 14
A circulating current begins to flow in the loop of the primary winding of the transformer 6. If the current flowing through the resonance inductor 14 is i ₂ , the following equation holds.

nV ₀ = L ₁ (di ₁ / dt) + L ₃ (di ₂ / dt) (10) In the case of load short circuit, if V ₀ = 0 in the equation (10), the following equation is obtained.

di ₁ / dt = − (L ₃ / L ₁ ) (di ₂ / dt) (11) As is clear from the equation (11), (di ₁ / dt) <0 even if a load short circuit occurs. It can be seen that the current in the primary winding of the transformer 6 is attenuated. In order to increase the attenuation factor, the inductance L ₃ of the resonance inductor 14 may be increased.

〔The invention's effect〕

As described above, the series resonant converter according to the present invention can eliminate the adverse effects of the leakage inductor of the transformer and provide good voltage drooping characteristics even in the case of an overload such as a load short circuit, so it is highly efficient and stable. There is an advantage that a series resonant converter having the above starting characteristics can be obtained.

BRIEF DESCRIPTION OF THE DRAWINGS FIG. 1 is a connection diagram of a series resonance converter of the present invention, and FIG.
The figure is a connection diagram of a conventional series resonant converter, and Fig. 3 is the second diagram.
FIG. 4 is a resonance current waveform diagram, FIG. 4 is a connection diagram of a series resonance converter in which the problems of the series resonance converter of FIG. 3 are improved,
FIG. 5 is a resonance current waveform diagram of FIG. 4, and FIG. 6 is a current waveform diagram when the voltage generated in the leakage inductor of the transformer is larger than the voltage of the primary winding of the transformer in FIG. 1 ... DC power supply, 2, 3 ... Switching element, 4, 5
...... Feedback diode, 6 …… Transformer, 7,8 …… Rectification diode, 9 …… Smoothing capacitor, 10 …… Load resistance, 11,12 …… Resonance capacitor, 13,14
…… Resonance inductor.

Claims (1)

[Claims] 1. A series circuit of two switching elements, a series circuit of two resonance capacitors, and a series circuit of two feedback diodes are connected to a DC power source, and are connected to a connection point of the series circuit of the switching element. The primary winding of the transformer is connected to the primary winding of the transformer, and the primary winding of the transformer is connected to the connection point of the series circuit of the resonant capacitor. The first resonance inductor of the series circuit is connected, and the second resonance inductor is connected between the connection point of the series circuit of the feedback diode and the connection point of the primary winding of the transformer and the first resonance inductor. A series resonance converter in which an inductor is connected and a full-wave rectifier circuit and a smoothing capacitor are connected to the secondary winding of the transformer.