JPH0712170B2 - Demodulator - Google Patents

Description

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a demodulation device of a digital signal transmission device that wirelessly transmits a digital signal in a multipath transmission line such as an urban area.

2. Description of the Related Art In recent years, even in the field of mobile communication, digitization is progressing due to improvement of confidentiality, sophistication of communication, and compatibility with peripheral communication networks. However, in urban areas where such demand is considered to be most concentrated, communication quality is significantly deteriorated due to multipath due to reflection and diffraction by buildings and other structures. In the case of digital transmission, if the propagation delay time difference between the waves forming the multipath cannot be ignored with respect to the data time slot, the waveform error and the tracking failure of the synchronous system will significantly deteriorate the code error rate characteristic.

Hereinafter, an example of the above-described conventional demodulation device will be described with reference to the drawings.

FIG. 7 shows a circuit configuration diagram of a modulator of a conventional digital signal transmission device. In FIG. 7, 71 is a data input terminal, 72 is a Gaussian low-pass filter, 73 is an FM modulator,
74 is a GMSK output terminal.

The operation of the modulator of the conventional digital signal transmission device configured as described above will be described below.

A Gaussian low-pass filter 72 limits the base band of an NRZ (Non Return Zero) digital signal. The band-limited signal enters FM modulator 73. The FM modulator has a modulation index of 0.
It is set to 5 and is called GMSK (Gaussion Filtered MSK) because it is MSK (Minimum Shift Keying) band-limited by a Gaussian filter in the base band. Similar to MSK, GMSK has the characteristics of a fixed line, and is also excellent in spectral concentration and convergence. (For example, KE MROTA, KE HIRADE: "GM SK Modulation For Digital Mobile Radio Radio Telephony" EYE ETRANS ACTION COM-29, No. 7, 1044-1
Page 050 (K.MUROTA, K.HIRADE: “GMSK Modulation for Dig
ital Mobile Radio Telephony "IEEE Trans., VOL.COM-2
9, NO.7, pp.1044-1050)) Regarding demodulation of such a GMSK signal, similar to MSK,
It is possible with either a synchronous detector or a frequency discriminator. Hereinafter, an example of a conventional demodulation device by the latter method will be described with reference to the drawings.

FIG. 8 is a circuit diagram of a conventional demodulator. In FIG. 8, 81 is an input terminal, 82 is an amplitude limiter,
83 is a monostable multi-valve, 84 is a low-pass filter, and 85 is a demodulation signal output terminal.

The operation of the conventional demodulation device configured as described above will be described below.

The GMSK signal input to the input terminal 81 is converted into a rectangular wave by the amplitude limiter 82. Further, it is converted into a pulse train having a constant width by the monostable multivalve 83. GMSK
Since the signal is a kind of FM signal, the density of the pulse train of this constant width changes depending on the modulation signal. Therefore, by averaging this pulse train by the low pass filter 84, the change in frequency can be extracted. (For example, Miki, "GM
Experimental Study of SK Frequency Detection ", IEICE Technical Report, CS82-89, 1982) Problems to be solved by the invention However, in the above-mentioned configuration, waveform distortion due to multipath is significant and code error occurs as described above. Especially, when we examine the relationship between the signal S / N ratio and the error rate, there is a region where the error rate does not decrease even if the S / N ratio is improved. It is called an irreducible error, etc. Due to such a so-called irreducible error, the actual data transmission rate in urban areas is greatly limited, and high-speed transmission is impossible.

In view of the above problems, the present invention provides a demodulation device of a digital signal transmission device capable of high-speed digital transmission in a multipath transmission line such as an urban area.

Means for Solving the Problems In order to solve the above problems, the demodulator of the present invention has a delay device for delaying a received signal by a time corresponding to one time slot of data, and an output signal of the delay device. A first multiplier which produces a component of the product of the received signal and a means for obtaining a signal which is 90 ° out of phase with the output signal of this delay device;
A second multiplier that produces a component of a product of a signal having a different phase and a received signal, and a first multiplier for blocking a frequency component other than the data base signal in the output signals of the first and second multipliers to obtain a detection signal. And a second low-pass filter, means for adjusting the timing of both detection signals, and an adder for synthesizing both detection signals.

Operation The present invention has the above-described configuration, and switches one data time slot to the first and second frequencies to perform differential encoding.
Phase demodulates the signal that has been phase-modulated and sent. There is a specific relationship between both frequencies and the data transmission rate, and by performing differential detection with the delay time of the delay device set to a value determined by these three parties, the signals sent at both frequencies are respectively Separate detection is performed, and the timing is adjusted and combined to obtain a demodulated signal. Therefore, the first or second frequency wave is a burst-like intermittent wave having a narrower width than that of the data signal time slot. Waveform distortion due to the path is not easily received. Further, since two frequencies are used, there is an effect of frequency diversity. Also, a time diversity effect due to the difference in transmission time of both frequencies can be expected. Due to the effects as described above, digital transmission at a higher speed than ever can be performed on a multipath transmission line.

Embodiment A demodulator according to an embodiment of the present invention will be described below with reference to the drawings.

FIG. 1 shows a circuit configuration diagram of a demodulation device according to a first embodiment of the present invention. In FIG. 1, 1 is an input terminal, 2 and 3 are balanced modulators, 4 is a 1 time slot delay device, 5 is a 90 ° phase shifter, 6 is a half time slot delay device,
Reference numerals 7 and 8 are low-pass filters, 9 is an adder, 10 is a band limiting filter, and 11 is a demodulation signal output terminal.

The operation of the demodulation device configured as described above will be described below with reference to FIGS. 1 and 2.

FIG. 2 shows a transmission signal. The data has been differentially encoded in advance, and has two-phase phase modulation (BPSK: Binary).
Phase Shift Keying) can be applied. Since it is differentially encoded, it is accurate to say that differentially encoded BPSK (DPSK: Differentia
l Phase Shift Keying) modulation is applied. However, as shown in the transmission signal 1 in FIG. 2, the transmission signal is transmitted at the frequency f ₁ in the first half of the time slot and at the frequency f ₂ in the latter half. Then, the transmission information is, for example, in FIG.
It is expressed by the phase difference between f ₁ -A and f ₁ -B and the phase difference between f ₂ -A and f ₂ -B. However, the phase difference referred to here is 0 ° or 180 °. The transmission signal may have a duty ratio smaller than that of the transmission signal 1, as shown in the transmission signal 2 in FIG.

It is assumed that the two transmission frequencies f ₁ and f ₂ and the data transmission rate f _b have a relationship represented by the following equation, where n is an integer.

f ₁ −f ₂ = (2n−1) × f _b / 4 …… If T is the time slot length, then T = 1 / f _b ……, but if both sides of the equation are rearranged by 2πT, then It becomes like the following formula.

2πf ₁ T−2πf ₂ T = (2n−1) × π / 2 From the equation, the relationship of the equation is that when the signal is delayed by T,
It is shown that the phase relationship between the f ₁ signal and the f ₂ signal further deviates from each other by 90 °.

In FIG. 1, a signal α as shown in the following equation is input to the input terminal 1.
Is entered.

α = a _n cos ω ₁ t + b _n cos ω ₂ t ... where a _n and b _n are data strings (a _n = ± 1, b _n = ± 1)
Therefore, ω ₁ = 2πf ₁ and ω ₂ = 2πf ₂ . That is, the signal α
Shows a binary phase-modulated signal at _two frequencies f ₁ and f ₂ . Further, assuming that the delay time T ₁ of the one time slot delay unit 4 has a relationship of cos (ω ₁ t + ω ₁ T ₁ ) = cos ω ₁ t ..., The signal after passing through the one time slot delay unit 4 β becomes β = a _n-1 cos ω ₁ t ± b _n-1 sin ω ₂ t ... By using the relation of the equation. Therefore, according to the equations, the output signals α and β of the balanced modulator 2 are as follows.

_{α · β = a n cosω 1} t · a n-1 cosω 1 t ± a n cosω 1 t · b n-1 sinω 2 t + b n cosω 2 t · a n-1 cosω 1 t ± b n cosω 2 t・ B _n-1 sin ω ₂ t = 1/2 {a _n a _n-1 (1 + cos2ω ₁ t) ± a _n b _n-1 sin (ω ₁ + ω ₂ ) t ± a _n b _n-1 sin (ω ₁ −ω ₂ ) t + b _n a _n-1 cos (ω ₁ + ω ₂ ) t + b _n a _n-1 cos (ω ₁ −ω ₂ ) t ± b _n b _n-1 sin2ω ₂ t} ...... where If the frequency components of 2ω ₁ , ω ₁ ± ω ₂ , and 2ω ₂ are removed by the low-pass filter 7, the output signal γ of the low-pass filter 7 is given by the following equation.

γ = 1 / 2a _n a _n-1 ... If the data string a _n is differentially encoded, the output signal γ becomes a demodulated data string. In this way, the conditions of the formulas are set for f ₁ , f ₂ , and f _b, and the delay time of the 1 time slot delay unit 4 is
By performing differential detection having the relationship of T ₁ in the equation, it is possible to obtain a detection signal of only the modulated data sequence a _n for the carrier wave of frequency f ₁ .

The detection of the modulated data string b _{n with} respect to the carrier wave of the frequency f ₂ can be performed in the same manner. By further phase-shifting the output signal of the one-time-slot delay unit 4 by 90 °, the output signal δ of the 90 ° phase shifter 5 is expressed by the following equation.

_{δ = b n-1 cosω 2} t ± a n-1 sinω 1 t ...... However, when the sign of the expression of the second term is +, 90 ° phase shifter 5 is of 90 ° phase advance, - when, With a 90 ° lag, the formulas and compound signs are in reverse order. Therefore, according to the equations and the equations, the output signal α · δ of the balanced modulator 3 similarly becomes the following equation.

α ・ δ = 1/2 {b _n b _n-1 (1 + cos2ω ₂ t) ± b _n a _n-1 sin (ω ₁ + ω ₂ ) t ± b _n a _n-1 sin (ω ₁ −ω ₂ ) t _{+ a n b n-1 cos} (ω 1 + ω 2) t + a n b n-1 cos (ω 1 -ω 2) t ± a n a n-1 sin2ω 1 t} ...... here, 2ω _1, ω ₁ ± omega _2, the frequency component of 2 [omega ₂ is to be removed by the low-pass filter 8, the output signal epsilon of the low-pass filter _{8, ε = 1 / 2b n} b n-1 ...... , and the data sequence b _n Is differentially encoded, the frequency
It is possible to obtain the detection signal of only the modulated data string b _n for the carrier wave of f ₂ .

As shown in FIG. 2, the transmission signal corresponding to the demodulator of the present invention has a frequency f in the first half of the data time slot.
_{At 1} , the latter half transmits the same differentially encoded data sequence at frequency f ₂ . Therefore, the half time slot delay unit 6 delays the detected signal corresponding to the former by half time slot to frequency-separate and receive the same data sequence transmitted by both frequencies, and detect the detected signals corresponding to each of them at the same timing. Can be synthesized. The respective detection signals synthesized by the adder 9 are band-limited by the band-limiting filter 10 to the extent that the data signal can pass, and noise components are removed. By recovering the clock component from the demodulated signal obtained in this way and instantaneously identifying the demodulated signal, the data string is decoded.

Next, the reason why the demodulator of the present invention exhibits excellent code error rate characteristics against multipath distortion will be described below with reference to FIGS. 3 to 5.

In the demodulation process, as described above, the carrier frequencies f ₁ and f ₂ are separately detected and then combined, so that
Consider the effect of multipath distortion on a transmission system with a frequency of f ₁ . A typical two-wave model is considered as a multipath model. The wave that precedes in time is called the direct wave, and the wave that is delayed is called the delayed wave.

FIG. 3 is a diagram illustrating how a detection signal of a transmission system having a frequency of f ₁ becomes under a two-wave multipath. FIG. 3A shows an example of phase transition of the direct wave. The amplitude becomes zero in the latter half of the time slot.
On the other hand, it cannot be ignored compared to time slots,
The phase transition of the delayed wave delayed by the propagation delay time difference τ is
It becomes as shown in FIG. As described above, since the detection method is differential detection of one time slot, the detection output at a certain point is the vector inner product of the combined phase of the two waves at that time and the combined phase of the two waves one time slot before. . For example, in FIG. 3 (c), the detection output in the section B is
It is the value of the vector inner product of the two-wave composite phase for B'and that for B. However, the time delay due to the half-time slot delay device 6 in FIG. 1 and the distortion of the detection signal waveform due to the low-pass filter 7 are not considered here for the sake of simplicity.

FIG. 4 illustrates the combined phase of the direct wave and the delayed wave at each time point A ′ to E ′ and A to E. In addition,
The amplitude ratio between the direct wave and the delayed wave is ρ, and the phase difference is φ. Fourth
From the figure, the detection output at each time point A to E in FIG. 3 (c) is as follows.

A …… 0 B …… 1 C …… 1 + ρ ² + 2ρ cosφ D …… ρ ² E …… 0 Depending on the values of ρ and φ, even if the detection output may become zero in the section of C, B or D In the section of, the detection output never becomes zero. In this way, deterioration of the unpattern due to multipath is small.

The same is true for the f ₂ frequency system. However, the phase difference φ between the direct wave and the delayed wave at the frequency of f ₂
Is uncorrelated with the phase difference φ at f ₁ and is generally different. That is, the value of the detection output in the section C is generally different from that of the system of f ₁ . Therefore, as described above, the half-time slot delay unit 6 in FIG. 1 synthesizes both detection outputs at the same timing, so that the diversity effect can be expected in the section C. To be precise, this diversity effect is a frequency diversity effect.

FIG. 5 is a diagram showing a state of such frequency diversity effect. In FIGS. 5A and 5B, the detection output 1 is the detection output of the frequency f ₁ system, and the detection output 2 is the frequency f ₂ system detection output. The timings of both are matched by the half-time slot delay device 6. By combining the detection output 1 and the detection output 2, a combined waveform as shown by the solid line in FIG. 5 (c) can be obtained. Further, by passing this combined waveform through the band limiting filter 10, a demodulated signal output as shown by the dotted line in FIG. 5 (c) can be obtained.

As described above, in the demodulator of the present invention, since each wave of the frequency f ₁ f ₂ is intermittent, there is little overlapping portion with the direct wave even if a delayed wave is present, and waveform distortion due to multipath is Hard to receive. Further, there is an effect of frequency diversity due to two frequencies. Further, the frequency f ₁ system and the frequency f ₂ system are temporally shifted by a half time slot, and a time diversity effect can also be expected. With the above effects, the code error rate characteristic is significantly improved in the multipath transmission line as compared with the conventional method, and high-speed digital transmission becomes possible.

In the above description, one time slot delay unit 4
The delay detection by, firstly, the transmission system of frequency f ₁ is detected, 1 having a delay time T ₁ in place of the equation satisfy _{cos (ω 2 t + ω 2} T 1) = cos ω 2 t ...... formula If the time slot delay device 4 is used, the transmission system of the frequency f ₂ is detected. Further, by supplying a signal obtained by phase-shifting this signal by 90 ° to the balanced modulator 2, the transmission system of the frequency of f ₁ is detected. Therefore, in this case, in FIG. 1, the output signal of the 1-time-slot delay unit 4 is supplied to the balanced modulator 3,
If the output signal of the phase shifter 5 is supplied to the balanced modulator 2, the same operation is performed.

Further, in FIG. 1, the 90 ° phase shifter 5 may be constituted by a circuit network including an active element such as a resistor or a capacitor in some cases, but may be constituted by a delay line as described below. If the delay time of this delay line is T ₂ and the phase rotation amount of this delay line is 90 ° with respect to the frequency f ₁ , then 2πf ₁ T ₂ = π / 2. On the other hand, the phase rotation amount with respect to the frequency of f ₂ is, 2 [pi] f by using Equation _{2 T 2 2πf 2 T 2 -} (2n-1) πf b T 2/2 With ...... equation equation 2 [pi] f ₂ T ₂ = π / 2− (2n−1) πf _b / 8f ₁ ...... Normally, f _b << f ₁ , so the second term on the right side of the equation can be ignored, and the equation becomes as follows, 2πf ₂ The phase rotates by almost 90 ° even for the frequency of T ₂ ≈ π / 2 …… f ₂ .

A second embodiment of the present invention will be described below with reference to the drawings.

FIG. 6 is a circuit configuration diagram of a demodulation device showing a second embodiment of the present invention. In the figure, 61 is an input terminal, and 62 and 63.
Is a balanced modulator, 66 is a half time slot delay, 67 and
68 is a low pass filter, 69 is an adder, 610 is a band limiting filter, and 611 is a demodulation signal output terminal. The above is the same as the configuration of FIG. The difference from the configuration of FIG.
The point is that the delay circuit composed of the time slot delay unit 4 and the 90 ° phase shifter 5 is composed of one time slot delay unit of 64 and 65.

The delay time of the one-time-slot delay unit 64 is selected to be the same T _{1 as} that of the one-time-slot delay unit 4. On the other hand, by selecting the delay time of the one-time-slot delay device 65 as T ₁ + T ₂ , the signals supplied to the balanced modulators 62 and 63 are
This is exactly the same as the embodiment of. Subsequent operations are the same as those in the first embodiment, and will be omitted.

In the first embodiment, the positions of the half time slot delay device 6 and the low pass filter 7 may be interchanged. Similarly, in the second embodiment, the positions of the half time slot delay device 66 and the low pass filter 67 may be interchanged.

Further, in the above description, as shown in FIG.
The number of transmissions of both frequencies in the time slot is 1 each
Although the transmission signal is used every time, it is demodulated in the same manner as in the above embodiment even for a transmission signal that is transmitted a plurality of times. That is, in all the above-described embodiments, demodulation can be performed in exactly the same way by making the delay time of the half-time slot delay device 6 or 66 equal to one transmission time of one frequency.

EFFECTS OF THE INVENTION As described above, according to the present invention, a delay device that delays a received signal by a time corresponding to one time slot of data, and a first multiplier that produces a component of a product of an output signal of the delay device and the received signal. A means for obtaining a signal having a 90 ° phase difference from the output signal of the delay device, a second multiplier for producing a component of the product of the 90 ° phase difference signal and the received signal, and the first and second First and second low-pass filters that obtain a detection signal by blocking frequency components outside the data base signal in the output signal of the multiplier, a means for adjusting the timing of both detection signals, and a combination of both detection signals By providing the adder, it becomes possible to perform digital transmission at a higher speed than before in the multipath transmission line.

[Brief description of drawings]

FIG. 1 is a block diagram of a demodulator in the first embodiment of the present invention, FIG. 2 is a schematic diagram of a transmission signal, and FIGS. 3 to 5 show that the demodulator of the present invention is resistant to multipath distortion. FIG. 6 is a schematic diagram of a signal for explaining, FIG. 6 is a block diagram of a demodulator in a second embodiment of the present invention, and FIGS. 7 and 8 are blocks of a modulator and a demodulator of a conventional digital signal transmission apparatus, respectively. It is a figure. 1,61 …… Input terminal, 2,3,62,63 …… Balanced modulator, 4,64,65
...... 1 time slot delay device, 5 …… 90 ° phase shifter, 6,66
...... Half-time slot delay device, 7,8,67,68 ...... Low pass filter, 9,69 …… Adder, 10,610 …… Band limiting filter, 11,611 …… Demodulation signal output terminal.

Claims (4)

[Claims] 1. A system in which signals of first and second frequencies are differentially encoded two-phase phase modulated with the same data signal and both modulated signals are switched and transmitted within one time slot of data. A first delayer for delaying the received signal by a time corresponding to one time slot of the data, the difference between the second frequency and the second frequency being an odd multiple of a quarter of the data transmission rate; and the first delayer. A first multiplier for producing a component of the product of the output signal and the received signal, means for obtaining a signal having a 90 ° phase difference from the output signal of the first delay device, and reception of the signal having a 90 ° phase difference A second multiplier that produces a signal product component, and first and second low components that block frequency components outside the data basis signal in the output signals of the first and second multipliers to obtain a detected signal. A bandpass filter and means for adjusting the timing of both detected signals , Comprises an adder for combining the two detection signals, said first and second signals that have been transmitted by the frequency respectively separated detection, demodulation and wherein the obtaining a demodulated signal by synthesizing. 2. The first means for obtaining signals having a 90 ° phase difference
2. A demodulator according to claim 1, wherein said demodulator comprises a 90 ° phase shifter connected to the output of said delay device. 3. Means for obtaining signals having a 90 ° phase difference is firstly provided.
2. The demodulator according to claim 1, wherein the demodulator comprises a second delay device connected to the output of the delay device. 4. A third means is provided for obtaining signals having 90 ° different phases.
The demodulator according to claim 1, wherein the delay signal is directly obtained by delaying the received signal.