JPH0748672B2 - Spread spectrum receiver - Google Patents

Description

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a spread spectrum receiver, and more particularly to improvement of a phase control method in a convolver system of the receiver.

[Summary of the Invention] Spread spectrum for demodulating data by correlation between the same spread spectrum received signal given to one input of two convolvers constituting a correlator by a switcher and two reference signals given to the other input In the receiver, a phase shifter is provided on the other input side of the convolver, and the phase shifter is adapted to generate a reference signal having a predetermined phase difference. When the CW signal is applied to one input of the convolver by switching, the phase adjusting PLL loop controls the phase of the phase shifter in response to the correlation output.

[Prior Art] In spread spectrum communication, a transmission spectrum is spread (widened) by a PN code (for example, an M sequence code having a code length of 127) that is much faster than the information rate to be transmitted, and the reception side In this case, data is demodulated by correlating with the PN code in the receiver, and it has the feature that deterioration of the received signal due to frequency selective fading can be reduced.

As a spread spectrum receiver used for such communication, there is, for example, the one disclosed in Japanese Patent Laid-Open No. 59-186440. The basic structure of this receiver is as shown in FIG.
The matched filters 1 and 2, the phase shifter 3, and the phase detector 4 are provided, and the spread spectrum received signal S is given to the matched filters 1 and 2 to obtain correlation, and output signals A and B are obtained.
Then, the output signal B is phase-shifted by 90 ° by the phase shifter 3, and the phase-shifted output signal B ′ and the output signal A are given to the phase detector 4 for detection, and the data signal D is demodulated. .

[Problems to be Solved by the Invention] However, according to the configuration of the spread spectrum receiver as described above, since the phase shifter is provided on the output side of the matched filter, the output signal of the wide band is naturally phase-shifted. This phase shifter needs to be wideband, as it must. Moreover, the wider the spread band, the wider the bandwidth of the phase shifter must be, and the phase must be uniformly shifted within that band. Realization of a phase shifter that satisfies these conditions Is very difficult. In addition, in order to correct the phase shift caused by variations in the characteristics of parts, variations in the electrical length of wiring, temperature changes, etc., the phase shifter must have a phase controllable structure. It is extremely difficult to achieve in.

Further, when the phase shifter is configured to be capable of controlling the phase as described above and the phase shift is automatically adjusted, the correlator input is the spread spectrum signal, and the correlator output becomes the correlation spike waveform signal. Has a drawback that the control circuit becomes complicated because the time width is narrow.

Therefore, an object of the present invention is to provide a spread spectrum receiver which is easy to realize a phase shifter, is capable of controlling its phase, and is capable of automatically adjusting the phase shift, which is suitable for practical use.

[Means for Solving the Problems] In order to achieve the above-mentioned object, the spread spectrum receiver of the present invention includes first and second convolvers in which the same spread spectrum received signal is given to each first input, A first reference signal generated by multiplying a first CW signal by a first PN code, a first multiplier for supplying a second input of the first convolver, and a first CW signal, The second CW signal generated by multiplying the second CW signal and the second PN code by a phase shifter for generating a second CW signal having the same phase and a predetermined phase difference with respect to the first CW signal. A second multiplier for applying the reference signal of 1 to the second input of the second convolver and a third multiplier for demodulating data by multiplying the outputs of both convolvers.
To the first and second multipliers, and first switching means for selectively applying the first and second PN codes or the DC bias voltage to the first and second multipliers, respectively, and to the first inputs of both convolvers. Second switching means for selectively applying the received spread spectrum signal or the first CW signal, and phase adjusting means for controlling the phase of the phase shifter in response to the output of the third multiplier. It is characterized by having and.

In the present invention, the phase control means includes a binarization circuit that determines the polarity of the third multiplier output and a sequential loop filter that filters the output of the binarization circuit. The output may control the phase shifter.

[Operation] In the spread spectrum receiver of the present invention, since the phase shifter only shifts the phase of the first CW signal to a predetermined value to generate the second CW signal, the phase shifter does not need to have a wide band, and Realization is easy, and the phase can be controlled. When the switch is switched and the first CW signal is applied to the first input of the convolver, the phase shift is automatically adjusted by the same operation as the PLL loop.

[Examples] The present invention will be described below with reference to the examples shown in the drawings.
FIG. 1 shows a basic configuration of an embodiment of a spread spectrum receiver according to the present invention. In the figure, 5 and 6 are convolvers, 7 and 8 are multipliers, 9 is a phase shifter, 10 and 11 are amplifiers, 12 is a multiplier, and 13 is a low-pass filter. Also
Reference numeral 14 is a switch, 15 is a phase control circuit, and 18 is a holding circuit.

When the switch 14 is on the contact I ₁ side, the received spread spectrum signal S is applied to one input of the convolvers 5 and 6,
The first and second reference signals R _f1 and R _f2 are applied to the other input.

The CW signal CW ₁ having the same frequency as the RF carrier signal of the spread spectrum signal S is given to one input of the phase shifter 9 and the multiplier 7. The phase shifter 9 phase shifts the CW signal CW ₁ by 90 ° and supplies it to one input of the multiplier 8.

The other input of the multipliers 7 and 8 is the PN code required for demodulation.
₁ ▼ and ▲ ₂ ▼ are given, and the outputs of the multipliers 7 and 8 become the first and second reference signals R _f1 and R _f2 .

The convolvers 5 and 6 respectively correlate the spread spectrum signal S with the first and second reference signals R _f1 and R _f2, and the respective correlation outputs V _c1 and V _c2 are multiplied by the multiplier 12 via the amplifiers 10 and 11. And the output of the multiplier is applied to the low-pass filter 13 to obtain the data demodulation signal V _f .

In the present invention, two channels, I and Q, are provided as transmission signals, and data is not placed on one channel in order to avoid the influence of multipath fading.

That is, even if the phase of the high frequency component in the output V _c1 of the convolver 5 (data is added) changes randomly due to the multipath fading, the phase of the high frequency component in the output V _{c2 of the} convolver 6 is also similarly affected. Therefore, multiply both by
It is possible to obtain the demodulation signal Vf having a polarity corresponding to the digital data “1” and “0” by multiplying by 12 and passing through the LPF 13.

Next, the data data demodulation signal V _f is obtained from the spread spectrum signal S received by the configuration of the above embodiment.

The received spread-spectrum signal S is S = Vd (t) = P 1 (t) SIN (ω 0 t) + A · P 2 (t) COS (ω
₀ t) is represented by (1). Here, P ₁ (t) and P ₂ (t) are the first and second PA codes used for modulation on the transmission side, A is data 1 or -1, and the signal S is two convolvers. Is given equal to.

First and second reference signals input to the two convolvers
R _f1 and R _f2 are the phase shift amount of the phase shifter θ, Is expressed as Where ₁ (t) and ₂ (t) are the PN codes ▲ ₁ ▼ and ▲ ₂ ▼ of the receiving side used at the time of demodulation, respectively, and the mirror image of P ₁ (t) and P ₂ (t) of the transmitting side. (Time inversion signal).

The outputs V _c1 and V _c2 of the two convolvers are V _c1 (t) = CONV {Vd (t), V _r1 (t)} (4) V _c2 (t) = CONV {Vd (t), V _r2 (t)} (5) Where CONV {V ₁ (t), V ₂ (t)} is two inputs V
Representing the convolution of ₁ (t) and V ₂ (t), if V ₁ (t) = COS (ω ₀ t)… (6) V ₂ (t) = COS (ω ₀ t + θ)… (7) , The convolver output CONV {V ₁ (t), V ₂ (t)} is CONV {V ₁ (t), V ₂ (t)} = η ・ COS (2ω ₀ t + θ + φ)…
(8) Here, η is the efficiency of the convolver, φ is the additional phase unique to the convolver, and it can be seen that the phase change θ of one input V ₂ (t) appears at the output as it is.

Now, P ₁ (t) and ▲ ▼, P ₂ (t) and ▲ ▼
Since the cross-correlation of is small, However, there is no big error. Further solving (9) and (10), V _c1 (t) = η ₁ · R ₁ (t) SIN (2ω ₀ t + φ ₁ )… (11) V _c2 (t) = η ₂ · A · R _{2 (t) COS (2ω 0 t} + θ + φ 2) ... (1
2) Where R ₁ (t) and R ₂ (t) are P ₁ (t) and ▲
The convolution of ▼, P ₂ (t) and ▲ ▼, φ ₁ and φ ₂ are additional phases unique to each convolver.

The output Vm (t) after multiplication of V _c1 (t) and V _c2 (t) is Vm (t) = V
_c1 (t) ・ V _c2 (t) = η ₁・ η ₂・ A ・ R ₁ (t) ・ R ₂ (t) ・ COS (2ω ₀ t +
φ ₁ ) ・ COS (2ω ₀ t + θ + φ ₂ ) ... (13) In the equation (13), if θ + φ ₂ = φ ₁ −π / 2… (14), then Vm (t) = η ₁ · η ₂ · A ・ R ₁ (t) ・ R ₂ (t) ・ SIN (2ω ₀ t
+ Φ ₁ ) ・ COS (2ω ₀ t + φ ₁ −π / 2) = η ₁・ η ₂・ A ・ R ₁ (t) ・ R ₂ (t) ・ SIN ² (2ω ₀ t +
φ ₁ ) (15) Further, the demodulated signal V _f (t) obtained by passing Vm (t) through a low-pass filter is V _f (t) = η ₁ · η ₂ · A · R ₁ (t) · R ₂ (t)… (17)

Fig. 2 shows an example of V _c1 (t), V _c2 (t) and V _f (t) in the case of equation (14). It is possible to demodulate data from the equation and equation (17). Recognize.

By the way, in the equation (14), when θ + φ ₂ = φ ₁ , V _f (t) = 0
Therefore, it cannot be demodulated. As mentioned above, φ ₁ ,
φ ₂ does not always match due to differences in the characteristics of the two convolvers, temperature characteristics, and slight differences in wiring length.

Therefore, in this case, the predetermined phase shift amount θ of the phase shifter is (14)
From the equation, it can be seen that θ = φ ₁ −π / 2−φ ₂ (16)

Further, it is obvious that the multipliers 7, 8 and 12 may be non-linear circuits using transistors or diodes, for example.

Therefore, in the present invention, in order to correct the phase shift in such a case, the switch 14 is switched to the contact I ₂ side and the convolver is switched.
The CW signal CW ₁ is applied to one of the inputs 5 and 6, and the first PN code and the second PN code are switched to the DC bias voltage.

If the input signal is Vd (t), then Vd (t) = COS (ω ₀ t) (18), and since the first PN code and the second PN code are switched to the DC bias voltage, Becomes Therefore, the convolution output V ′ _c1 (t),
_V'c1 (t) is _V'c1 (t) = η ₁ · COS (2ω ₀ t + φ ₁ ) (20) _V'c2 (t) = η ₂ · COS (2ω ₀ t + φ ₂ ) ... (21) Becomes Then V _'c1 (t) and V' _c2 output V 'm (t) after multiplication of the (t) _{is, V'm (t) = η 1} · η 2 · COS (2ω 0 t + φ 1) · COS
(2ω ₀ t + φ ₂ ) (22) Here, if θ + φ ₂ = φ ₁ −π / 2 (23), then V′m (t) = η ₁ · η ₂ · COS (2ω ₀ t + φ ₁ ) · SIN
(2ω ₀ t + φ ₁ ) (24), and the LPF output V ′ _f (t) is V ′ _f (t) ∝ η ₁ · η ₂ · COS (φ ₁ −θ −φ ₂ )… (25) Becomes

When the output is a positive value, the phase shifter is controlled to the delayed phase, and when the output is negative, the phase shifter is controlled to the advanced phase, so that V ′ _f (t) = The balance is controlled by 0 (26). The equilibrium condition at this time is φ ₁ −θ−φ ₂ = π / 2 (27), and therefore θ = φ ₁ −φ ₂ −π / 2 (28).

The condition of equation (28) matches the optimum state of data demodulation.

Moreover, this control can be achieved by a well-known PLL control method.

That is, the output of the LPF 13 is given to the binarization circuit 16, the output of the binarization circuit is filtered by the sequential loop filter 17, the output of the sequential loop filter is given to the holding circuit 18, and the output of the holding circuit is given. Therefore, the phase shifter 9 may be controlled by. Here, the holding circuit 18
While the phase control is being performed, the output of the sequential loop filter 17 is given to the phase shifter 9 as it is. However, during the data demodulation, the sequential control immediately before switching from the phase control to the data demodulation is performed. Loop filter
It is used to hold the output of 17 and continue to give the value to the phase shifter 9 to keep the optimum phase state.

The phase shifter 9 may be of any type as long as the phase can be controlled, and the switcher may be operated at an appropriate periodic timing as required.

In addition to the above, the phase control circuit 15 may be an analog PLL that passes the output of the LPF 13 through the loop filter 19 and supplies it to the holding circuit 20, as shown in FIG. Binarization circuit 21, CPU2 as shown in FIG.
2, it is obvious that it may be a PLL configured by the D / A converter 23.

[Effects of the Invention] As is apparent from the above description, according to the present invention, the phase shifter is arranged at the input side of the correlator consisting of two convolvers, unlike the conventional case, and a wide band like a spread spectrum signal is obtained. It is easy to realize and suitable for practical use because it is necessary to shift the phase of the non-CW signal.

Further, for example, the phase control of the phase shifter necessary for correcting the phase shift due to the difference between φ ₁ and φ ₂ can be easily realized.

[Brief description of drawings]

FIG. 1 is a block diagram showing an embodiment of the present invention, FIG. 2 is a waveform diagram for explaining its operation, and FIG. 3 is a block diagram showing an example of a conventional spread spectrum receiver, FIGS. 4 and 5. FIG. 3 is a block diagram showing a configuration of a phase control circuit. 5,6 ... Convolver, 7,8,12 ... Multiplier, 9 ... Phase shifter.

Claims (2)

[Claims] 1. A first and a second convolver, wherein the same spread spectrum received signal is applied to each first input, and a first convolver.
The first generated by multiplying the CW signal of
A first multiplier for providing the second reference signal of the first convolver to the second input of the first convolver, the first CW signal being input, and the first CW signal having the same frequency and a predetermined phase difference with respect to the first CW signal. A phase shifter for generating two CW signals, and a second reference signal generated by multiplying the second CW signal and the second PN code by a second convolver second signal.
Second multiplier applied to the input of the above, a third multiplier for demodulating data by multiplying the outputs of both convolvers, and a first and a second PN, respectively. First switching means for selectively applying a sign or DC bias voltage, and second for selectively applying the received spread spectrum signal or first CW signal to the first inputs of both convolvers.
And a phase adjusting means for controlling the phase of the phase shifter in response to the output of the third multiplier. 2. The phase control means has a binarization circuit for determining the polarity of the output of the third multiplier, and a sequential loop filter for filtering the output of the binarization circuit, and the sequential loop filter. The spread spectrum receiver according to claim 1, wherein the phase shifter is controlled by the output of the.