JPH0785677B2 - Control method of voltage source inverter - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION [Industrial application] The present invention mainly relates to a control method for preventing occurrence of torque ripple and deterioration of control characteristics in a voltage control type PWM inverter.

[Conventional technology]

In general, vector control that enables high-speed response and high-accuracy control is known for a device that controls the speed of an induction motor using a frequency converter or an inverter. This vector control has been developed as a control system that obtains the same performance as a DC motor with a squirrel-cage induction motor, so to speak, the rectifying function of the DC machine is performed by voltage control,
It is intended to control the rotation speed by forming an orthogonal relationship between the magnetic flux vector and the current vector.

The inverter device uses a voltage-type inverter that uses a transistor or GTO, and the output current is controlled by an AC current controller so that it is proportional to the sine wave current command pattern (so-called current control type). It is used because of the fact that But recently
In order to simplify the circuit configuration by deleting the AC current regulator, and to reduce the burden of arithmetic processing when a microprocessor is applied to the current regulator and the PWM signal generator,
A method called voltage control type has been announced (Multimicropr
ocessor-Based Control System for Quick Response I
nduction Motor Drive, IEEETRANS. on INDUSTRY APPLIC
ATIONS, VoL, IA-21, No4, MAY / JUNE 1985, PP602-60
9). In this device, the output voltage command of the inverter is calculated based on the current command and the frequency command, and the output voltage is controlled by PWM control. However, as will be described later, the output voltage cannot be accurately controlled due to the voltage drop (non-linear with respect to the output current) of the inverter circuit, and therefore when the vector control is applied, the coordinate reference axis (voltage command reference phase) of the output voltage command is applied. ) Does not match the magnetic flux axis (phase of induced electromotive force) of the motor, and high-speed response and high-precision control, which is a feature of vector control, cannot be satisfactorily performed. Further, since the voltage drop of the inverter includes a harmonic component (because the voltage drop is non-linear with respect to the output current), torque ripple occurs. These phenomena are particularly remarkable at low speed rotation.

The cause of the above-mentioned voltage drop will be described below.

In the pulse width modulation inverter, the P-side and N-side switching elements that form the inverter are alternately conductively controlled to perform PWM control of the output voltage. However, since the switching element has a delay in switching due to the turn-off time, one side should be turned off and the other side should be turned on after a predetermined time (dead time) so that the P side and the N side do not turn on at the same time. ing.

Due to the influence of the dead time, the output voltage fluctuation and the distortion become large, especially when the inverter output frequency is low, which causes the above-mentioned problem.

[Problems to be solved by the invention]

Conventionally, as a countermeasure against this, as described in Japanese Patent Publication No. 59-8152 and Japanese Patent Publication No. 59-123478, the output current direction of the inverter is detected, and based on this detection output, the inverter due to the influence of dead time is detected. A method of providing a compensating means for compensating the waveform distortion of the output voltage is known. However, in these methods, the influence of the variation of the turn-off characteristics of the switching element and the variation of the turn-off time due to the magnitude of the current, that is, the dependence of the voltage drop on the output current is not compensated and cannot be sufficiently compensated.

In addition, as a method to prevent this drawback, as described in pp27-28 of "Mitsubishi Electric Technical Report" Vol. 58. No12.1984, by detecting the instantaneous output voltage of the inverter and performing feedback control to the command signal. A method of compensating for waveform distortion of an output voltage is known. However, this method requires a detector for detecting the output voltage of the inverter,
Further, since the output voltage of the inverter is almost proportional to the output frequency, the lower the frequency is, the more difficult it is to detect and the insufficient compensation cannot be performed.

[Object of the Invention]

An object of the present invention is to provide a control method for a voltage source inverter that can prevent the deterioration of control characteristics and the occurrence of torque ripple.

[Means for solving problems]

The present invention estimates the voltage drop of an inverter circuit based on an inverter output current detection signal, subtracts a predetermined amount according to the output current from the estimated voltage drop value, and corrects the voltage command based on the subtracted amount to correct the voltage command. The harmful DC component of the output current caused by the unbalanced operation of the switching elements (P side and N side) is prevented.

〔Example〕

 Examples of the present invention will be described below.

FIG. 1 shows an embodiment of the present invention. In the figure, a PWM inverter 1 converts a DC voltage into a variable frequency AC voltage. The inverter 1 is composed of a self-extinguishing element and a feedback diode connected in antiparallel to each self-extinguishing element. A switching element such as a transistor or a gate turn-off thyristor is used as the self-extinguishing element. An induction motor 2 is connected to the AC output terminals of each phase U, V, W of the inverter 1. The primary currents iu, iv, iw (output current of the inverter 1) of the U-phase, V-phase and W-phase of the induction motor 2 are detected by the current detectors 3U, 3V, 3W. The speed command signal ωr * of the speed command circuit 6 is added to the adder 7,
Speed estimation signal from 39 And the adder 7 outputs those deviations. The speed deviation amplifier 9 outputs a command signal iq * of a motor current component iq described later according to the deviation. The signal iq * is added to the adder 11 and the coefficient unit 29. The adder 11 uses the signal iq *
The current deviation amplifier 35, which outputs a deviation between a signal iq (described later in detail) extracted from the coordinate converter 32, outputs a frequency control signal Δω according to the deviation. The signal Δω is an adder
36, where the output signal ω from the first-order delay circuit 37
r ^** (a signal that is first-order delayed with respect to the speed command signal ωr *)
Is added and the frequency command signal ω ₁ * is output.

Further, the oscillator 12 outputs a sine wave signal (a phase reference of the inverter output voltage) whose amplitude is constant at a frequency proportional to the command signal ω ₁ *. The output signal is applied to coordinate converters 17, 18, 32.

The detection signals iu, iv, iw of the current detectors 3U, 3V, 3W are applied to the three-phase / two-phase converter 31, and the AC signals iu, iv, iw are converted into two-phase signals iα, iβ.

The signals iα, iβ are applied to the coordinate converter 32, and based on the sine wave signal of the oscillator 12, the signals iα, iβ are converted into a component delayed by 90 ° and an in-phase component with respect to the induced electromotive force, Each signal
Output i _d and i _q .

The signal i _d is added to the adder 33, and the signal i _q is added to the adder 11 and the function generator.
Added to 38. The function generator 38 outputs the slip frequency signal ωs based on the signal i _q . The adder 39 outputs the signal ω
Estimated value of the rotation speed of the induction motor 2 by subtracting the signal ωs from ₁ * Is output to the adder 7.

The exciting current command circuit 13 outputs an exciting current command i _d * for the motor. The signal i _d * is applied to the induced electromotive force calculator 14, the adder 33 and the coefficient unit 40. The adder 33 outputs the deviation between said i _d * and i _d , which is added to the current deviation amplifier 34. The amplifier 34 outputs a voltage command signal v _d * according to the deviation, and the output is added to the adder 41. The coefficient unit 40 outputs the signal i _d *
Is multiplied by k and output. The output signal ki _d * of the coefficient unit 40 is applied to the adder 41, where the output signal of the current deviation amplifier 34.
Add a new voltage command signal v _d ^** to v _d * and add it to adder 28.

The coefficient unit 29 multiplies the signal i _q * by k and outputs it. The output signal ki _q * of the coefficient unit 29 is added to the adder 30, where it is added to the output signal v _q * of the induced electromotive force calculator 14 to output a new voltage command signal v _q ^** , which is added. Add to machine 15. Here, the signals Δv _d * and Δv _q * applied to the adders 15 and 28 will be described later. The coordinate converter 18 converts the output signal v _d ^*** of the adder 28 and the output signal Δv _q ^*** of the adder 15 from the rotating magnetic field coordinates to the stator coordinates based on the sine wave signal of the oscillator 12. Two-phase AC signals vα * and vβ * are output. Signal vα
*, Vβ * is added to the 2-phase / 3-phase converter 19, and the voltage command signals v _u *, v _v *, v _w * whose frequency is 120 ° phase difference proportional to the frequency command signal ω ₁ * are taken out. . Signal v _u *, v
_v *, _{v w} * is applied to each switch circuit 20.

The output signals of the switch circuit 20 are comparators 24U, 24V, 24
W is added to and compared with the carrier signal from the oscillator 25 for pulse width modulation control. And PWM inverter 1
It generates pulse width modulation pulses (PWM pulses) to turn on and off the switching elements that make up the. The gate circuit 26 gives a gate signal to the switching element according to the output pulses of the comparators 24U, 24V, 24W. The above is the configuration related to vector control.

Next, the components for compensating the voltage drop of the inverter related to the present invention will be described.

The function generators 4U, 4V, 4W generate signals Δv _u *, Δv related to the voltage drop of the inverter and the resistance drop of the motor winding and wiring according to the magnitude and polarity of the output signals of the current detectors 3U, 3V, 3W.
Outputs _v * and Δω *. The signal is a 3-phase to 2-phase converter 1
It is added to 6 and converted into a two-phase signal of Δvα * and Δvβ *. The signals Δvα *, Δvβ * are applied to the coordinate converter 17, and based on the sine wave signal of the oscillator 12, the signals Δvα *,
Δvβ * is converted into a component delayed by 90 ° with respect to the induced electromotive force and an in-phase component, and signals Δv _d * and Δv _q * are output. The adder 15 adds the output signal v _q ^** of the adder 30 and the signal v _q *, and outputs a new voltage command signal v _q ^*** . The adder 28 adds the signal v _d ^** and the signal Δv _d * and outputs a new voltage command signal v _d ^*** .

As described above, FIG. 1 shows an example of application of the present invention to a vector control device that does not use a speed sensor. First, the operation of vector control will be briefly described. The output currents i _{u to} i _w detected by the current detectors 3U to 3W are converted into two-phase signals iα and iβ in the three-phase to two-phase converter 31, and further the phase reference signal from the oscillator 12 in the coordinate converter 32. And the respective current component signals i _d and i _q are detected. The relational expression is as follows.

Where cosw ₁ t and sinw ₁ t are phase reference signals,
The phase of sinw ₁ t matches the induced electromotive force of the U phase of the electric motor. i
d is the sum of the current components of each phase with a phase delay of 90 degrees with respect to the induced electromotive force of each phase, and corresponds to the exciting current of the motor. on the other hand,
i _q is a current component in phase with the induced electromotive force, and corresponds to the torque current of the electric motor (proportional to the secondary electric current of the electric motor).

The deviation between the excitation current command i _d * and the signal i _d from the command circuit 13 is amplified in the amplifier 34, and the voltage command v _d * is extracted,
In accordance with this, the d-axis voltage v _d (voltage component orthogonal to the induced electromotive force) of the electric motor is controlled, and as a result, i _d matches i _d *.

On the other hand, since the torque, the slip frequency and the current component i _q of the electric motor are proportional to each other, the slip calculator 38 estimates the slip frequency ωs based on the signal i _q . Adder 39
, The slip estimation signal ωs is subtracted from the frequency command signal ω * to obtain the velocity estimation signal Is detected. Speed command signal ω _r * and signal Of the torque current command i _q
* Is taken out. Further, the deviation between i _q * and i _q is increased in the amplifier 35, and the frequency command ω ₁ * is determined according to the output signal Δω. At this time, if i _q is smaller than i _q *, ω ₁ * increases, the slip frequency changes in the increasing direction, and i _q increases. The reverse operation is performed in the opposite way, i _q
Is controlled to match i _q *.

Further, the signal ω ₁ * and i _d * are multiplied to obtain the voltage command v _q *, and the q-axis voltage v _q (the voltage component in phase with the induced electromotive force) of the motor is controlled accordingly, resulting in induction. It is controlled so that the ratio of electromotive force and frequency is constant (constant magnetic flux).

As described above, the magnetic flux of the motor is controlled to a predetermined value according to the exciting current command i _d *, and the slip frequency and the torque are controlled according to the torque current command i _q *. As described above, in the present control device, the voltage commands v _d * and v _q * are calculated based on the current commands i _d *, i _q * and the current component detection signals i _d and i _q ,
This is converted into a stator coordinate amount (three-phase AC) by the coordinate converter 18 and the two-phase to three-phase converter 19, and the instantaneous value command v _u * to v _w * of the inverter output voltage is calculated and proportional to this. To control the inverter output voltage.

Here, there is one problem. That is, the voltage command v _U
Since the relationship between * to v _w * and the actual output voltage has a non-linear characteristic for the reason described later, the output voltage cannot be controlled according to the voltage command and vector control cannot be performed satisfactorily. Because, from the above, the voltage commands v _d *, v _q * and the actual motor voltage v _d , v _q do not match, and as a result, the induced electromotive force that should originally match and the phase of the phase reference signal of the oscillator 12 do not match, The detection of the signals i _d and i _q and the control operation described above become uncertain, and stable operation cannot be performed with the high-speed response that is a feature of vector control.

The cause of the above-mentioned non-linear characteristic will be described below.

The switching element forming the inverter 1 has a time delay in turn-off, and in order to prevent a short circuit of the positive side arm and the negative side arm due to it, the gate signal based on the PWM pulse includes a signal shown in FIG. An on-delay (dead time) of time t _d shown by the broken line in FIG. Normally, the on-delay time t _d is the turn-off time of the switching element, which is 2
It is set to 3 times. Now, when the line voltage (= E _UN −E _VN ) between the U and V phases is calculated by ignoring the on-delay time t _d , the waveform shown by the solid line in FIG. 2D is obtained. Next, determine the line voltage U-V phases considering the Onderei time t _d. The direction in which the output current of the inverter 1 flows from the inverter 1 into the electric motor 2 has a positive polarity. Assuming that the U-phase current is positive and the V-phase current is negative, the on-delay period is
Current flows through the diode provided in anti-parallel to the switching element of the negative arm. Therefore, the potential between U during the on-delay period becomes negative. On the other hand, in the V-phase in which the current flows negatively, the current flows through the diode in the positive arm during the on-delay period. Therefore, the V-phase potential during the on-delay period becomes +. From the above, the U phase and V phase considering the on-delay time are as shown in FIGS. 2 (e) and 2 (f). From this, the line voltage between the U and V phases (= V _un −V _vn )
Shows the waveform (hatched portion) shown by the broken line in FIG. 2D, and the area is smaller than the waveform of the solid line when the on-delay time is ignored.

Further, it is known that the turn-off time of the switching element changes depending on the magnitude of the current flowing through the switching element (output current of the inverter 1). Therefore, the period during which the switching elements of the positive and negative arms are simultaneously turned off during the on-delay period varies depending on the magnitude of the output current of the inverter 1. That is, the size of the hatched portion of the waveform of the broken line in consideration of the on-delay time in FIG. 2D changes according to the size of the output current of the inverter 1.

As described above, the voltage drop due to on-delay occurs at every on-off cycle of the switching element. If the PWM switching frequency is sufficiently higher than the inverter output frequency, the phase of the voltage drop (fundamental wave component) is the inverter output current. They are in phase, and are similar to the resistance drop in this respect.

On the other hand, the magnitude of the voltage drop is non-linear with respect to the output current. The characteristic can be measured as follows.

Now, the voltage command signals v _u * to v _w * for each phase are v _u * = v _dc * v _v
Fig. 3 shows the equivalent circuit of the motor when * =-v _dc *, v _w * = 0 is set and a direct current is passed from the inverter to the motor. Here, E is the voltage drop of the above-mentioned inverter, and R is the resistance of the motor winding and the wiring cable between the inverter and the motor. At this time, the relationship between the voltage command v _dc * and the inverter output current i _dc (direct current) is expressed by the following equation.

v _dc * = Ri _dc + E (i _dc ) That is, Ri _dc and E can be measured from v _dc *. here,
If R is known, Ri _dc and E can be separated. An example of the measurement result of the voltage drop is shown in FIG. As shown in the figure, the magnitude of the voltage drop is non-linear with respect to the output current, so that when the output current is an alternating current, the voltage drop includes a harmonic component.

Therefore, the voltage drop E and the resistance drop Ri are measured in advance, and the characteristics are set in the function generator (memory) 4U to 4W,
From that, E and Ri are taken out according to the output current, and the part for compensating each voltage drop is the function generators 4U-4W,
A three-phase / two-phase converter 16, a coordinate converter 17, and adders 15 and 28. That is, in the configuration described above, the voltage drop signal Δ
Convert vu * to Δvw * into two-phase signals Δvα * and Δvβ *,
Furthermore, the rotating magnetic field coordinate quantities Δv _d *, Δv q * are converted to adders.
In addition to the 15, 28, v _d * and v _q * (v _q ^**) Fix v _d ^** and v
_{q Take} out ^*** . Since each voltage drop is predicted and the voltage drop is compensated as described above, the induced electromotive force of the motor can be controlled so as to match the voltage commands v _d * and v _q * (v _q ^** ).

By the way, if the sum of the voltage drop E and the resistance drop Ri of the inverter is set as the input / output characteristic of the function generator, both voltage drops can be compensated, but due to the unbalanced operation of the positive side and the negative side of the switching element of the inverter. It is not possible to suppress the generation of the DC component of the output current due to the generated DC component of the inverter output voltage. Therefore, the quantities related to the output current are E and Ri.
It may be better to subtract from the sum of and compensate based on it. That is, the contents of the function generators 4U to 4W are set according to the following equation as an example.

Output of function generator = (R−k) i + E where ki is a quantity related to the output current described above. Note that this value does not have to be proportional to i, and a similar effect can be obtained as long as it is an amount related to i.

Further, the resistance drop component proportional to the output current of the inverter 1 is gain K (= R + k) in the coefficient unit 40 and the coefficient unit 29.
Is added and added to the adders 41 and 30. By doing so, the voltage drop due to the on-delay time of the inverter 1 and the voltage drop due to the winding resistance and the wiring resistance of the induction motor 2 can be compensated for, so that the induced electromotive force of the induction motor 2 is calculated by the induction electromotive force calculator 14. It matches well with the output signal v _q *. As a result, it is possible to prevent the deterioration of the control performance described above.

Next, a method for automatically setting the contents of the function generator will be described.

The voltage drop E of the inverter is peculiar to the inverter,
The resistance drop Ri changes depending on the resistance of the connected motor and wiring. Therefore, it is necessary to set the contents of the function generators 4U to 4W accordingly, but it is complicated. Therefore, a method of automatically measuring these voltage drops and automatically setting the contents of the function generator will be described.

As shown in FIG. 1, 21,22U to 22W, the switch circuit 20, and the meter 5 are related components.

First, E and Ri are measured as follows prior to the actual operation of the electric motor. That is, the DC voltage command v _DC * is output from the DC voltage commander, and the voltage command v _u * of each phase is output from 22U to 22W.
~ V _w * respectively, v _u * = v _dc *, v _v * =-v _dc *, v _w * = 0
Set to. At this time, the switch circuit 20 is switched to the a side. At this time, the current i _dc flows from the inverter,
If v _dc * with respect to i _{dc at} that time is measured, it becomes as shown in FIG. This characteristic is set in the function generators 4U to 4W.

During actual operation, the switch circuit is switched to the b side, and the output voltage is controlled according to the output signals v _u * to v _w * of the 2-phase to 3-phase converter 19. At this time, the signal of the function generators 4U to 4W Δv _u *
As described above, the voltage drop is compensated according to ˜Δv _w *. As described above, in order to prevent the generation of the direct current component of the inverter output current, the amount related to the output current can be subtracted from the measurement result, and the characteristic can be set in the function generator.

It should be noted that in the above-described embodiment, the invention is applied to the analog circuit in order to make the explanation of the operation easier to understand, but it is obvious that the present invention can be applied to a digital control unit using a microprocessor.

FIG. 5 shows another embodiment of the present invention. The same parts as those in FIG. 1 are designated by the same reference numerals, and the description thereof will be omitted. The difference from FIG. 1 is that the outputs of the function generators 4U, 4V, 4W are added to the AC voltage command signals v _u *, v _v *, v _w *, respectively.

The 2-phase to 3-phase converter 19 has a voltage command signal v _u *, v _v *, v _w * having a phase difference of 120 ° whose frequency is proportional to the frequency command signal ω ₁ *.
Is output. The voltage command signals v _u *, v _v *, and vω are added to the adders 40U, 40V, 40W, respectively. Adder 40U, 40V, 40W voltage command signal _{v u *, v v *,} vω * a function generator 4U, 4V, the output signal of _{4W Δv u *, Δv v *} , added by polarity Derutabuiomega * and shown , Voltage command signals v _u ^** , v _v ^** , _v ω ^** are output. The voltage command signals v _u ^** , v _v ^** , and vω ^** are applied to the switch circuit 20, respectively.

According to this embodiment, the same effects as those of the embodiment of FIG. 1 can be obtained, and the converters 16 and 1 required in the embodiment of FIG.
7 can be omitted and the configuration becomes simple.

〔The invention's effect〕

As described above, according to the present invention, the voltage drop of the primary resistance of the wiring and the motor and the internal voltage drop of the inverter due to the on-delay are compensated, so that stable control can be performed at low speed and low frequency. Further, since the generation of the direct current component of the output current of the inverter is suppressed, the effect that torque ripple does not occur can be obtained.

[Brief description of drawings]

FIG. 1 is a circuit configuration diagram of a PWM inverter device showing an embodiment of the present invention, FIG. 2 is an operation waveform diagram for explaining the operation of the PWM inverter device, and FIG. 3 is a circuit configuration for explaining the operation principle of the present invention. 4 and 5 are characteristic curve diagrams for explaining the operation principle of the present invention, and FIG. 5 is a circuit configuration diagram of a PWM inverter device showing another embodiment of the present invention. 1 ... Inverter, 2 ... Induction motor, 3 ... Current detector, 4 ... Function generator.

 ─────────────────────────────────────────────────── ─── Continuation of front page (72) Inventor Noboru Fujimoto 4026 Kuji-machi, Hitachi City, Ibaraki Prefecture Hitachi Research Laboratory, Hiritsu Manufacturing Co., Ltd. (56) References JP-A-60-118082 (JP, A)

Claims (1)

[Claims] 1. A voltage source inverter for supplying an alternating current of variable voltage variable frequency to an AC motor, wherein the output voltage of the voltage source inverter is controlled according to an output voltage command. Output current of the inverter, the voltage drop due to the on-delay of the inverter obtained based on the output current detection value, and the bridge-connected switching element of the main circuit constituting the inverter obtained according to the output current detection value A method of controlling a voltage source inverter, wherein the output voltage command is corrected according to a sum of an inverter output voltage DC component generated by an unbalanced operation of the positive side arm and the negative side arm.