JPH0812240B2 - Electronic scanning monopulse receiver and electronic scanning monopulse / transceiver using the receiver - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION (Industrial field of application) The present invention irradiates a target such as an aircraft, a flying object or a vehicle with a radio wave and determines the angle of the target by using the radio wave reflected from the target as a medium. An electronic scanning monopulse receiver and electronic scanning monopulse transmitter / receiver that can perform beam driving of an antenna by an electronic method instead of a servo system using a machine and also realize spatial stabilization in a tracking guided vehicle mounted radar It is about machines.

(Summary of the Invention) The present invention provides an electronic scanning monopulse receiver and an electronic scanning monopulse transceiver that can perform beam driving of an antenna by an electronic method instead of a mechanical servo device and can also realize spatial stabilization. Therefore, by performing phase processing and signal processing by a monopulse comparator after dividing the received signal of each antenna, the swing angle of the antenna beam can be increased even if the distance between the antennas is increased. As a result, the number of antenna elements can be reduced.

(Prior Art) The problems of the electronic scanning type monopulse antenna up to now will be described using the conventional example of FIG.

In FIG. 3, the first received signal 71 is the first antenna 74.
Received at the first antenna output 75,
In the phase shifter 20, the phase amount corresponding to the signal of the phase generator first output 29, which is the output of the phase generator 28, changes to become the first phase shifter output 21, which is input to the monopulse comparator 24. .

The second reception signal 73 is received by the second antenna 76 and becomes the second antenna output 77, and the second phase shifter 22 changes the phase amount according to the signal of the phase generator second output 31 to change the second It becomes the phase shifter output 23 and is input to the monopulse comparator 24.

In the monopulse comparator 24, the sum and difference of the first phase shifter output 21 and the second phase shifter output 23 are calculated to be a monopulse comparator sum output 25 and a monopulse comparator difference output 27.

The angular velocity detector 86 is an electronic scanning monopulse antenna
It detects the angular velocity of the motion of a flying body equipped with a seeker, and is composed of, for example, a gyro. The output of the angular velocity detector 86, which is the output of the angular velocity detector 86, is
Is integrated into the integrator output 89, and is further nonlinearly corrected by the tangent corrector 32 to be the tangent correction output 33, which is input to the phase generator 28.

Here, when the target is in the direction of the angle α as shown in FIG. 2, the first antenna output 75 (X ₇₅ ) and the second antenna output 77 (X ₇₇ ) are as follows.

X ₇₅ = sin (ωt + φ) (1) X ₇₇ = sin (ωt−φ) (2) However, the phase difference 2φ between the first antenna output 75 (X ₇₅ ) and the second antenna output 77 (X ₇₇ ) is , Caused by the direction of arrival of radio waves deviating from the center of the two antennas 2
It is the phase difference corresponding to the arrival time difference at the two antennas,
The relationship between α and φ is shown by the following equation from FIG.

φ = (πd / λ) sinα (2a) where λ is the wavelength of the angular frequency ω (radian / second) of the radio wave. In order to have a one-to-one correspondence between α and φ, | α | << π (2b). Maximum power can be received in the direction that is deviated from the center by α (radian).

The first phase shifter 20 shifts the phase by −Θ radian,
When the phase shifter 22 shifts the phase by Θ radian, the following signals are output to the first phase shifter output 21 (X ₂₁ ) and the second phase shifter output 23 (X ₂₃ ), respectively.

X ₂₁ = sin (ωt + Δθ) (3) X ₂₃ = sin (ωt−Δθ) (4) where Δθ = φ−θ (4a). The monopulse comparator sum output 25 (X ₂₅ ) and the monopulse comparator difference output 27 (X ₂₇ ) become X ₂₅ = 2cosΔθsinωt… (4b) X ₂₇ = 2sinΔθcosωt… (4c), and the first intermediate frequency amplifier 82 and the second Intermediate frequency amplifier
The first intermediate frequency amplifier output 83, which is amplified by 84 respectively
(X ₈₃ ) and the second intermediate frequency amplifier output 85 (X ₈₅ ) X ₈₃ = cos Δθ sinωt (4d) X ₈₅ = sin Δθ cosωt (4e) At this time, the error angle output 35 (X ₃₅ ) which is the output of the multiplication demodulator 34 becomes X ₃₅ = sin (2Δθ) sin2ωt… (4f), and the direction of the beam of the antenna when 2Δθ = 0… (4g) is the target. Position (angle)
Is. The case where the first phase shifter 20 shifts the phase by −Θ radian and the second phase shifter 22 shifts the phase by Θ radian has been described, but the phase shift amount is set to satisfy the expression (4g). May be given respectively.

By controlling the phase generator 28 using the error angle output 35, the error angle output 35 can be tracked to zero.

(Problems to be Solved by the Invention) By the way, the conventional angle tracking radar is roughly classified into a method of mechanically driving an antenna beam and a method of electronically driving the antenna beam.

In order to mechanically drive the antenna beam, since the antenna beam is driven by a mechanical servo device, it is difficult to reduce the size and weight and drive the antenna beam at high speed.

On the other hand, the technique of driving the antenna beam electronically is already used as shown in FIG. But the third
In the conventional electronic scanning type monopulse antenna shown in the figure, when the distance d between the antennas is larger than the wavelength λ of the radio wave used, as is clear from the equation (9) referred to in the description of the principle of the present invention described later. , The range of the swing angle of the antenna directivity angle becomes narrow. That is, in order to execute target tracking over a wide range, the number of antenna elements must be increased.

Proportional navigation of guided missiles equipped with small radar devices
In order to make a stable flight toward the target, even if there is shaking (vibration) in the bullet, only the antenna beam is always directed toward the target, which requires spatial stabilization performance.

Therefore, a. It is possible to drive the antenna beam at high speed. An electronic scanning monopulse receiver and an electronic scanning monopulse transceiver having a small number of antenna elements have been demanded.

(Means for Solving the Problems) In view of the above points, the present invention uses an electronic scanning monopulse receiver capable of driving an antenna beam at high speed and having a small number of antenna elements, and the receiver. The present invention aims to provide a conventional electronic scanning monopulse transceiver.

The present invention, in an electronic scanning monopulse receiver having a plurality of antennas, divides a signal received by one antenna by a first frequency divider into a first divided signal and a signal received by another antenna into a second divided signal. The second frequency-divided signal frequency-divided by the frequency divider is phase-adjusted by the phase shifter, added to the monopulse comparator, converted into the sum signal and the difference signal by the monopulse comparator, and the sum signal and the difference signal Is detected respectively, and a large / small discriminator discriminates between these two amplitude detection outputs, and a large signal is used as a signal for automatic gain control, and the larger one of the sum signal or the difference signal is output. There is provided means for taking out an angle error signal by performing a multiplication operation in a multiplier using the sum signal and the difference signal which are automatically gain-controlled while keeping the signal constant. It has solved the problems.

Prior to explaining the embodiment of the present invention shown in FIG.
The most basic means of the present invention will be described with reference to the drawings.

The distance between the first antenna 74 and the second antenna 76 is d (centimeter), and the wavelength of the radio wave used is λ.
(Cm) When the radio wave from the target arrives by deviating from the front (antenna center axis A) by α (radian), the first antenna output when received by two antennas (74 and 76) When the phase difference between 75 and the second antenna output 76 is 2φ, 2φ = (2πd / λ) sinα (5) or φ = (πd / λ) sinα (5a). However, π is the pi.

Similarly, assuming that the directivity angle when the two antennas are oscillated during transmission and reception is β (radian), the phase difference 2θ between the signals transmitted and received by the two antennas is 2θ = (2πd / λ) sinβ (6) or θ = (Πd / λ) sinβ (7)

If the phase difference 2θ is created between the signals of the two antennas, the antenna can be swung by β, but θ is the phase, and θ and β are in a one-to-one relationship due to the periodicity of the triangular function. Θ must be in the range of −π / 2 <θ <π / 2… (8), and the swing angle β must be −arcsin [λ / (2d)] <β <arcsin [ λ / (2d)]…
It falls within the range of (9), and according to the equation (9), when λ / (2d) is small, the swing angle β becomes a small value.

In order to solve the problem, the present invention intends to make λ effectively large so that the antenna pointing direction can be largely swung even when the distance d between the antennas is large (Equation (9) above). Expand the upper and lower limits of β by multiplying the apparent λ by an integer). Therefore, according to the present invention, an analog frequency divider (54 or
56) is used. The analog frequency divider contains at its output both amplitude and phase information of the original signal.

In the case of reception, the source 40, the first of which is indicated by the dotted line in FIG.
And the second analog multipliers 42 and 44 are not used, but the first and second analog frequency dividers 54 and 56 are used. Details of the configuration of the first and second analog frequency dividers 54 and 56 are shown in FIG.

In the analog frequency divider of FIG. 6, the second output 47b (X ₄₇ b) of the third duplexer can be expressed the same as the first antenna output 75 (X ₇₅ ) shown in FIG. 1 and FIG. Therefore, X ₇₅ = X ₄₇ b = sin (ωt + φ) (10).

The principle of the first analog frequency divider 54 and the second analog frequency divider 56 will be described. In FIG. 6, the first and π / 2 phase shifters
Of the noise components of the first and π / 2 phase shifter output 67, which is the output of 66, X ₆₇ = εcos {(ωt + φ) / 2} (10a) (where ε is the amplitude term) Then, the first mixer output 51, which is the output of the first mixer 50, passes through the low-pass filter 78 to the low-pass filter output 79 (X ₇₉ ) and is multiplied by X ₇₉ by the multiplication and filtering effect. = (Ε / 2) sin {(ωt + φ) / 2} (10b) is output. Since the phase of the first / π / 2 phase shifter output 67 is delayed by π / 2, X ₆₇ = (ε / 2) Acos {(ωt + φ) / 2} (10c). However, A is the gain (amplification degree) of the loop. Therefore, A> 2 ... (10d), the amplitude of the reference signal of the first .pi. / 2 phase shifter output 67 is made larger than that of the input signal (third transmission / reception switch second output 47b), and the first .pi. / 2 If the leakage of the phase shifter output 67 to the first mixer output 51 is minimized, the first mixer output will be output when there is no input.
When the output to 51 disappears and there is an input signal, the sixth
As shown in the figure, the output 79 of the low-pass filter is 1/2 of the input signal (the second output 47b of the third duplex switch) X ₇₉ = (1/2) sin {(ωt + φ) / 2 } (10e) is output, and the amplitude and phase of the input signal are divided into 1/2 and saved.

That is, the output of the first / π / 2 phase shifter 66 of FIG. 6, which is the output of the first / π / 2 phase shifter 67 (X ₆₇ ), is X ₆₇ = (A / 2) cos {(ωt + φ) / 2} (11), and the first output together with the third output switch second output 47b
Mixed in the mixer 50 to become the first mixer output 51,
The high-pass component is filtered by the low-pass filter 78 to become the low-pass filter output 79. A first power divider output 61 and a first analog divider output 55 which are outputs of the first power divider 60.
Can be expressed in the same way as the low-pass filter output 79. X ₇₉ = X ₆₁ = X ₅₅ = E {X ₄₇ bX ₆₇ } = (1/2) sin {(ωt + φ) / 2}… (12 (However, E {} represents an average). In this way, the analog frequency divider holds the amplitude information correctly while having a function as a frequency divider.

The above α (radian) is the direction of the incoming radio wave, but the above β (radian) wants to end up in the direction of α, so that what has been described above regarding α is also applicable to β. Therefore, in order to obtain the output of the analog frequency divider, the frequency is halved, the wavelength λ is doubled, arcsin [λ / (2d)] is increased from the equation (9), and the swing angle of β is increased. Can be increased. With this, the most basic channel of the receiving system can be explained.

Next, the case of transmission will be described. First in transmission
The transmission source 40 and the first and second analog multipliers 42 and 44, which are indicated by dotted lines in the figure, are used, and the first and second analog frequency dividers 54 and
56 is not used (see FIG. 5, which is a block diagram during transmission). Details of the configuration of the first and second analog multipliers 42 and 44 are shown in FIG.

The transmission source output 41 of FIG. 5 is X ₄₁ = sin [(ω / 2) t] (13a), which is shifted by −θ / 2 in the first phase shifter 20. and Figure 7 a first duplexer first output 37a (X ₃₇ a) and the second is the output of the second power divider 62
The first output 63 (X ₆₃ ) of the power distributor is X ₃₇ a = X ₆₃ = sin [(ωt−θ / 2] (13)) The second output π of the second output 65 of the second power distributor / 2
The second π / 2 phase shifter output 69 (X ₆₉ ) which is the output of the phase shifter 68
As compared with the first duplexer first output 37a (X ₃₇ a) π /
Since the phase is delayed by 2, X ₆₉ = cos [(ωt−θ) / 2] (14), and the first analog multiplier output 43 (X ₄₃ ) is switched to the first mixer in the second mixer 52. Device first output 37a and second
・ Since it is made by the product of π / 2 phase shifter output 69, X ₄₃ = sin (ωt−θ) (15). Thus, the first and second analog multipliers 42,4
In 4, the analog multiplier output that accurately holds the relationship between the frequency and the phase of the input and the output is obtained as shown in the equations (13), (114) and (15). With this, it was found that the transmission source of the angular frequency ω / 2 can transmit the angular frequency ω, and the same device portion can be shared for the transmission and the reception, and the most basic channel of the transmission system can be explained.

(Operation) The operation described above shows how the configuration can increase the swing angle of the antenna.

The first analog frequency divider output 55 (X ₅₅ a), which is the output of the first analog frequency divider 54 shown in the embodiment of FIG. 1, is X ₅₅ a = sin [(ωt + φ) / 2] (18) ). The second analog frequency divider output 57 (X ₅₇ a), which is the output of the second analog frequency divider 56, becomes X ₅₇ a = sin [(ωt−φ) / 2] (19).

The output of the first phase shifter 20, the first phase shifter second output 21b
(X ₂₁ ba) is phase-shifted by -θ / 2 to X ₂₁ ba = sin [(ωt + φ) / 2-θ / 2] (20), and the second phase shifter second output 23b (X ₂₃ ba) ) Is phase-shifted by θ / 2, resulting in X ₂₃ ba = sin [(ωt−φ) / 2 + θ / 2] (21).

In this case, the monopulse comparator sum output 25, which is one output of the monopulse comparator 24, is the first phase shifter second output 21b (X ₂₁
ba) and second phase shifter second output 23b (X ₂₃ ba) sum (addition)
Since it made of _{X '25 = X 21 ba +} X 23 ba = 2cos (φ / 2-θ / 2) sin [(ω / 2) t] ... and becomes (22), monopulse comparator sum output 25 (X ₂₅ a) is delayed by π / 2 in the monopulse comparator 24 (it is delayed by π / 2 due to the nature of the monopulse comparator) X ₂₅ a = cos (φ / 2−θ / 2) cos [(ω / 2) t] (23) The other output of the monopulse comparator 24, the monopulse comparator difference output 27 (X ₂₇ a), is in this case the first phase shifter second output 21b (X ₂₁ ba) and the second phase shifter second output 23b. (X ₂₃
Since made the difference ba) _{(subtraction) X 27 a = X 23 ba} -X 21 ba = sin (φ / 2-θ / 2) cos [ω / 2) t] ... a (24).

Since the output of the multiplication demodulator 34 in FIG. 4 is made by averaging the products of the monopulse comparator sum output 25 (X ₂₅ a) and the monopulse comparator difference output 27 (X ₂₇ a), the error angle output 35 (X _{35 a} a) is approximately X ₃₅ a ≈ sin (φ−θ) (25) using a low-pass filter. To make equation (25) zero, φ should be equal to θ. By doing so, the incoming radio waves can be received by matching the incoming direction of the radio waves with the antenna directivity angle. At this time, the phase shifters 20 and 22 need to have a phase shift amount of θ / 2 as described above.

Since the monopulse sum output 25 is delayed in phase by π / 2 radians, synchronous detection is possible, and the error angle output 35 (X ₃₅ ) of equation (25) is obtained.

In the conventional example, the relationship between β and θ is shown in FIG. 2, and since θ = (πd / λ) sinβ (36a) as shown in equation (6) later, the error angle output 35 Becomes X ₃₅ = sin2θ (36b) when the amount of phase shift is zero in the first phase shifter 20 and the second phase shifter 20. In order to have a one-to-one correspondence between β and θ, −π / 2 <2θ <π / 2… (36c) −π / 2 <2 (πd / λ) sin β <π / 2… (36d) is necessary. On the other hand, in the present invention, X ₃₅ = sin θ (36e). In order to have a one-to-one correspondence between β and θ, it becomes −π / 2 <θ <π / 2… (36f) −π / 2 <(πd / λ) sinβ <π / 2… (36g). , Θ is twice as wide, the swing angle β
Is twice as wide. Alternatively, it is shown that the spacing between the antennas can be doubled if the swing angles of the antennas are equal.

From the above description, if the frequency used is halved by using the analog frequency divider and the conventional synchronous detector is made into a circuit including the multiplier / adjuster 34, apparently λ in equation (2) is 2 It is shown that an electronic scanning type monopulse receiver in which the oscillating angle β of the antenna is doubled can be configured.

Next, the transmission state will be described with reference to FIGS. 5 and 7. The transmission source output 41 (X ₄₁ ) which is the output of the transmission source 40 can be assumed to be X ₄₁ = sin [(ω / 2) t] (37), and the first phase shifter first output 21a (X ₂₁ a) and the first output switch 37a (X
₃₇ a) becomes X ₂₁ a = X ₃₇ a = sin [ωt−θ) / 2] (38).

In FIG. 7 showing the analog multiplier, the output 69 (X ₆₉ ) of the second π / 2 phase shifter 68, which is the output of the second π / 2 phase shifter 68, is X ₆₉ = cos [(ωt−θ) / 2] (39) The first analog multiplier output 43 (X ₄₃ ) is the first output / reception switch first output 37a (X ₃₇ ) of the formula ( ₃₈ ) and the second of the formula (39).
・ Because it is made by the product of the π / 2 phase shifter output 69 (X ₆₉ ), X ₄₃ = sin (ωt−θ) (40) and the first transmission output 71a (X ₇₁ a) from the first antenna 74. Is radiated toward the target.

X ₇₁ a = sin (ωt−θ) (41) Similarly, the second transmission output 73a is radiated from the second antenna 76 toward the target.

_{X 73 a = sin (ωt +} θ) ... (42) Here, the equation first transmission output 71a (X _71a) and a second transmission output 73a (X _73a) of formula (42) in (41), the synthesis of the transmission signal _(Z 71a73a) becomes _{X 71a73a = X 71a + X 73a} = cos2θsinωt = cos {2 (πd / λ) sinβ} sin (ωt) ... (43). However, the relationship between β and θ is 1 in FIG.
In order to correspond by pair 1, θ = (πd / λ) sinβ (44) | β | << π (45). The maximum power t is radiated in the direction offset by β (radian) from the center.

From the above description, if an analog multiplier is used on the transmitting side and an analog frequency divider is used on the receiver, even if the spacing between the antenna elements is widened, the antenna beam is continuously transmitted during transmission and reception compared to the conventional case. You can see that you can shake it greatly.

(Example) Hereinafter, the Example of this invention is described according to drawing.

FIG. 1 shows a receiving state of a high frequency section in the embodiment, and FIG. 5 shows a transmitting state. The apparatus of the embodiment includes a first antenna 74, a first analog frequency divider 54, a first analog multiplier 42 and the like. It includes a first transmitting / receiving module including the second antenna 76, a second antenna 76, a second analog frequency divider 56, a second analog multiplier 44, and the like.

First, the detailed description will be started from the block diagram showing the reception state in FIG. However, the portion used only in the transmission state in FIG. 1 is shown by a dotted line.

The first reception signal 71, which is a reflected wave from the target, is received by the first antenna 74 and input to the third transmission / reception switcher 46. Since the switcher 46 is now in the reception state, the third transmission / reception switch The output of the switch 46 becomes the third output switch second output 47b, is input to the first analog frequency divider 54, and becomes the first analog frequency divider output 55. Since the first transmission / reception switching device 36 is in the receiving state, the output of the first transmission / reception switching device 36 becomes the first transmission / reception switching device second output 37b, which is input to the first phase shifter 20 and the phase generator first output 29. The signal is subjected to phase modulation by the signal of 1), becomes the first phase shifter second output 21b, and is input to the monopulse comparator 24. The phase generator first
As the structure for obtaining the output 29, the same structure as the conventional case shown in FIG. 3 can be used.

Similarly, the second received signal 73 that is a reflected wave from the target is the second
The signal is received by the antenna 76 and input to the fourth transmission / reception switching device 48. Since the switching device 48 is now in the reception state, the output of the fourth transmission / reception switching device 48 is the fourth transmission / reception switching device second output 49b. Is input to the second analog frequency divider 56 and becomes the second analog frequency divider output 57. Since the second transmission / reception switch 38 is in the reception state, the second analog frequency divider 56
Output becomes the second output / switch second output 39b.
It is input to the phase shifter 22, undergoes phase modulation by the signal of the phase generator second output 31, and becomes the second phase shifter second output 23b,
It is input to the monopulse comparator 24.

Next, the signal processing unit of FIG. 4 will be described. The monopulse comparator sum output 25 is input to and amplified by the first intermediate frequency amplifier 82 whose amplification degree is controlled by the AGC amplifier output 59, and becomes the first intermediate frequency amplifier output 83. One of the outputs of the first intermediate frequency amplifier output 83 is input to the first amplitude detector 90 and becomes the first amplitude detector output 91.

The monopulse comparator difference output 27 is input to and amplified by the second intermediate frequency amplifier 84 whose amplification degree is controlled by the AGC amplifier output 59, and becomes the second intermediate frequency amplifier output 85. Second
One output of the intermediate frequency amplifier output 85 is the second amplitude detector.
It is input to 92 and becomes the output 93 of the second amplitude detector.

In the large / small discriminator 80, the first amplitude detector output 91 and the second amplitude detector output 93 are input, and the larger one of them is output to become the large / small discriminator output 81, which is input to the AGC amplifier 58. Becomes the output 59 of the AGC amplifier and is input to the first intermediate frequency amplifier 82 and the second intermediate frequency amplifier 84. As a result, the larger one of the first intermediate frequency amplifier output 83 and the second intermediate frequency amplifier output 85 is controlled to be constant.

The other output of the first intermediate frequency amplifier output 83 is the second output.
Multiply demodulator with the other output of the intermediate frequency amplifier output 85
Entered in 34. The multiplying demodulator 34 has a function equivalent to synchronously detecting the output having the smaller amplitude among the outputs 83 and 85 with the larger amplitude as the reference signal. In the multiplication demodulator 34, the multiplication operation corresponding to the equation (35) or the equation (36) is performed, and the output is proportional to the angular difference between the target position (angle) and the antenna pointing direction (angle). An error angle output 35, which is an angle error signal of elevation angle or azimuth angle, is output and is used as a signal for evaluating and tracking a target position.

Next, FIG. 5, which is an explanatory diagram of the transmission state, will be described in detail. However, the portion used only in the receiving state in FIG. 5 is shown by a dotted line.

Source output 41 produced by source 40 is a monopulse comparator
It is input to 24, and becomes the transmission source 1st output 26 and the transmission source 2nd output 30 through the path | route reverse to a receiving state. The transmission source first output 26 is phase-modulated in the first phase shifter 20 using the signal of the phase generator first output 29 to become the first phase shifter first output 21a,
Since it is now in the transmitting state, the first phase shifter first output 21a input to the first transmission / reception switching device 36 becomes the first transmission / reception switching device first output 37a, and the first analog multiplier 42 (frequency divider). 54
The first analog multiplier output 43 is passed through the third analog transmission / reception switching device 46 to become the third analog transmission / reception switching device first output 47a. It is transmitted from the one antenna 74 to the target as the first transmission output 71a.

Similarly, the second source output 30 is at the second phase shifter 22,
The second phase shifter, which is phase-modulated using the signal of the second output 31 of the phase generator, becomes the first output 23a of the second phase shifter, and the second phase shifter is input to the second transmission / reception switch 38 because it is in the transmitting state. The first output 23a becomes the second output / switching switch first output 39a, is input to the second analog multiplier 44 (having a multiplication ratio which is the reciprocal of the division ratio of the frequency divider 56), and is multiplied to produce the second analog multiplication. Output 45
Then, after passing through the fourth transmission / reception switching device 48, the fourth transmission / reception switching device has a first output 49a, which is transmitted from the second antenna 76 to the target as the second transmission output 73a.

 A supplementary explanation of the embodiment will be given.

(I) Although the division ratio is set to 1/2 in FIG. 6, other division ratios, for example, 1/4 may be used.

(B) Although the multiplication ratio is set to 2 in FIG. 7, other multiplication ratios,
For example, it may be 4.

(A) Although an analog multiplier has been described in FIG. 7 for the sake of explanation, it does not have to be an analog system because it is a transmission system.

(Effects of the Invention) According to the electronic scanning monopulse receiver and the electronic scanning monopulse transceiver using the receiver according to the present invention, the following effects can be obtained.

(Ii) The head swing of the antenna can be greatly moved during transmission and reception of the electronic scanning antenna.

(B) Even with four transmission / reception modules, a small monopulse radar that tracks the elevation angle and the azimuth angle can be configured, which is extremely effective as a radar for mounting a missile that requires a small size and light weight.

(Ha) By using the angular velocity detector attached to the flying object, even if the flying object fluctuates as shown in Fig. 3, it is possible to cancel the fluctuation with a small number of antenna elements and direct the antenna beam to the target. be able to.

[Brief description of drawings]

FIG. 1 is a block diagram showing a receiving state in a high frequency section in an embodiment of an electronic scanning monopulse receiver and an electronic scanning monopulse transceiver according to the present invention, and FIG. 2 is for explaining the embodiment of FIG. 3 is an explanatory view showing the relationship between the signal from the target and the directivity angle of the antenna, FIG. 3 is an explanatory view showing a conventional example of the electronic scanning monopulse antenna, and FIG. 4 is a block diagram of the signal processing unit of the embodiment. FIG. 5 is a block diagram showing a transmitting state in a high frequency section in the embodiment, FIG. 6 is a block diagram of an analog frequency divider used in the embodiment, and FIG.
The figure is a block diagram of an analog multiplier used in the embodiment. 20 …… First phase shifter, 21 …… First phase shifter output (X ₂₁ ), 21a
...... First phase shifter first output, 21b ...... First phase shifter second output (X ₂₁ ba, X ₂₁ bb), 22 ...... Second phase shifter, 23 ...... Second phase shifter output (X ₂₃ ), 23a …… Second phase shifter first output, 23b ……
2nd phase shifter 2nd output (X ₂₃ ba, X ₂₃ bb), 24 …… monopulse comparator, 25 …… monopulse comparator sum output (X ₂₅ ), 26
...... Transmission source first output (X ₂₆ ), 27 …… Monopulse comparator difference output (X ₂₇ ), 28 …… Phase generator, 29 …… Phase generator first output, 30 …… Transmission source second output , 31 …… Phase generator 2nd output, 32 …… Tangent corrector, 33 …… Tangent corrector output, 34
...... Multiply demodulator, 35 ・ ・ ・ Error angle output (X ₃₅ ), 36 ...... First transmission / reception switching device, 37a ...... First transmission / reception switching device 1st output (X ₃₇ a), 37b ...... First transmission / reception switching Second output (X ₃₇
a), 38 ... second transmission / reception switching device, 39a ... second transmission / reception switching device first output, 39b ... second transmission / reception switching device second output, 40 ... transmission source, 41 ... transmission source output (X ₄₁ ), 42 ……
1st analog multiplier, 43 ... 1st analog multiplier output (X ₄₃ ), 44 ... 2nd analog multiplier, 45 ... Analog multiplier output, 46 ... 3rd transmission / reception switch, 47a ... Three
Duplexer first output, 47b ...... third duplexer second output (X ₄₇ b), 48 ...... fourth duplexer, 49a ...
... 4th transmission / reception switcher first output, 49b ... 4th transmission / reception switcher 2nd output, 50 ... first mixer, 51 ... first mixer output, 52 ... second mixer, 54 ... First analog divider,
55 …… First analog frequency divider output (X ₅₅ ), 56 …… Second analog frequency divider, 57 …… Second analog frequency divider output (X ₅₇ ), 58 …… AGC amplifier, 59 …… AGC Amplifier output, 60
...... First power distributor, 61 …… First power distributor output (X ₆₁ ), 62 …… Second power distributor, 63 …… Second power distributor first output, 65 …… Second power distributor 2nd output, 66 …… 1st π / 2 phase shifter, 67 …… 1st π / 2 phase shifter output (X ₆₇ ), 68 …… 2nd π / 2 phase shifter, 69 …… Second, π / 2
Phase shifter output (X ₆₉ ), 71 ... First received signal, 71a ... First
Transmission output (X _71a ), 73 ... Second received signal, 73a ... Second
Transmission output (X ₇₃ a), 74 ... First antenna, 75 ... First antenna output (X ₇₅ ), 76 ... Second antenna, 77 ... Second
Antenna output (X ₇₇ ), 78 …… Low pass filter, 79 ……
Low pass filter output (X ₇₉ ), 80 ... Large / small discriminator, 81 ...
… Large / small discriminator output, 82 …… first intermediate frequency amplifier, 83 ……
First intermediate frequency amplifier output, 84 ... Second intermediate frequency amplifier, 85
...... Second intermediate frequency amplifier output, 86 …… Angular velocity detector, 87
...... Angular velocity detector output, 88 …… Integrator, 89 …… Integrator output, 90 …… First amplitude detector, 91 …… First amplitude detector output (X ₉₁ ), 92 …… Second amplitude detector Device, 93 ... Output of second amplitude detector (X ₉₃ ).

Claims (2)

[Claims] 1. An electronic scanning monopulse receiver having a plurality of antennas, wherein a first divided signal obtained by dividing a signal received by one antenna by a first divider and a signal received by another antenna are The second frequency-divided signal frequency-divided by the frequency divider 2 is phase-adjusted by the phase shifter, added to the monopulse comparator, and converted into the sum signal and the difference signal by the monopulse comparator. Amplitude detection is performed on each of the signals, and the large and small discriminators discriminate between the two amplitude detection outputs, and the large signal is used as a signal for automatic gain control. An electronic scanning monopal, wherein an output signal is made constant and an angle error signal is taken out by performing a multiplication operation in a multiplier using the sum signal and the difference signal which are automatically gain controlled. Receiver. 2. An electronic scanning monopulse transceiver having a plurality of antennas, wherein a signal received by one antenna is frequency-divided by a first frequency divider and a signal received by another antenna. And a second frequency-divided signal obtained by frequency division by a second frequency divider in a phase shifter for phase adjustment and addition to a monopulse comparator, and conversion into a sum signal and a difference signal by the monopulse comparator. And the difference signal are respectively amplitude-detected, and the large and small discriminators discriminate between the two amplitude detection outputs, and the large signal is used as a signal for automatic gain control. An electronic scanning monopulse capable of extracting an angle error signal by performing a multiplication operation in a multiplier by using the sum signal and the difference signal whose automatic gain control is made constant while keeping the output signal of the other one constant. In the above arrangement, the transmission system commonly uses the phase shifter also in the transmission system, and further divides the frequency division of the first and second frequency dividers in parallel with the first and second frequency dividers via the transmission / reception switch. An electronic scanning monopulse transmitter / receiver characterized in that a multiplier having a multiplication ratio of the reciprocal of the ratio is connected.