JP4295183B2 - Antenna direction adjusting method and OFDM receiving apparatus - Google Patents
Antenna direction adjusting method and OFDM receiving apparatus Download PDFInfo
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- JP4295183B2 JP4295183B2 JP2004269995A JP2004269995A JP4295183B2 JP 4295183 B2 JP4295183 B2 JP 4295183B2 JP 2004269995 A JP2004269995 A JP 2004269995A JP 2004269995 A JP2004269995 A JP 2004269995A JP 4295183 B2 JP4295183 B2 JP 4295183B2
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- H—ELECTRICITY
- H01—ELECTRIC ELEMENTS
- H01Q—ANTENNAS, i.e. RADIO AERIALS
- H01Q3/00—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system
- H01Q3/02—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system using mechanical movement of antenna or antenna system as a whole
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- H—ELECTRICITY
- H01—ELECTRIC ELEMENTS
- H01Q—ANTENNAS, i.e. RADIO AERIALS
- H01Q3/00—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system
- H01Q3/22—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system varying the orientation in accordance with variation of frequency of radiated wave
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/26—Systems using multi-frequency codes
- H04L27/2601—Multicarrier modulation systems
- H04L27/2647—Arrangements specific to the receiver only
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Description
本発明は直交周波数分割多重変調方式(Orthogonal Frequency Division Multiplexing:以下、ＯＦＤＭ方式と記す)を用いた伝送装置における受信アンテナの方向調整に関するものである。 The present invention relates to direction adjustment of a receiving antenna in a transmission apparatus using an orthogonal frequency division multiplexing (hereinafter referred to as OFDM).
近年、無線装置の分野では、マルチパスフェージングに強い変調方式として、ＯＦＤＭ方式が脚光を集め、欧州や日本を初めとする各国で多くの応用研究が進められている。 この内、ＵＨＦ帯の地上デジタル放送の開発動向と方式については、例えば、非特許文献１に詳しく記されている。
このＯＦＤＭ方式は中継現場の映像をテレビ局まで無線伝送するＦＰＵ(Field Pickup Unit)の伝送方式として採用され、固定や移動無線中継に使用されている。
従来のアンテナの方向調整方法では、受信側の検波器により受信電界レベルを検出し、受信電界レベルが最大になる方向を探索してアンテナの方向を調整する方法が取られてきた。しかし、受信電界が低くなると、受信信号は雑音に埋もれてしまい、精度の良い検出が困難であった。即ち、アンテナの方向調整の初期段階では受信電界は非常に低いため、ＯＦＤＭ信号が僅かながら到達していたとしても、雑音に埋もれた受信信号を捕らえることができず、手探りで方向調整を行わざるを得なかった。
In recent years, in the field of wireless devices, the OFDM method has attracted attention as a modulation method that is resistant to multipath fading, and many applied researches are being promoted in countries such as Europe and Japan. Among these, the development trend and method of terrestrial digital broadcasting in the UHF band are described in detail in Non-Patent Document 1, for example.
This OFDM system is adopted as a transmission system of FPU (Field Pickup Unit) that wirelessly transmits video at a relay site to a television station, and is used for fixed or mobile wireless relay.
In the conventional antenna direction adjustment method, a method of adjusting the antenna direction by detecting the received electric field level with a detector on the receiving side and searching for the direction in which the received electric field level becomes maximum has been taken. However, when the received electric field is lowered, the received signal is buried in noise, and it is difficult to detect with high accuracy. In other words, since the received electric field is very low at the initial stage of antenna direction adjustment, even if the OFDM signal arrives a little, the received signal buried in noise cannot be captured, and direction adjustment must be carried out by groping. Did not get.
そこで本出願人は、これらの欠点を改善したアンテナ方向調整方法（例えば、特許文献１）を考案した。以下、図３を用いて簡単に説明する。
送信装置１２の送信前処理回路１３ａに入力された情報符号は、誤り訂正符号への変換、６４ＱＡＭへの変調等の前処理により、各搬送波の信号を表す周波数分布イメージの信号列に変換され、逆フーリエ変換(ＩＦＦＴ)回路１３ｂで時間波形を表す信号列に変換される。そしてガードインターバル挿入回路１３ｃで、伝送路での遅延波に起因する受信側での符号間干渉の影響を少なくするため、送信されるＯＦＤＭ信号にガードインターバルが付加される。このガードインターバルを挿入された信号は、送信後処理回路１３ｄにおいて更に直交変調、Ｄ／Ａ変換、アップコンバート等の後処理を施され、送信アンテナ１１から送信される。
この送信されたＯＦＤＭ伝送信号は、受信アンテナ２１で受信され、ケーブルを通して復調装置２２に送られる。 復調装置２２に入力された受信信号は、ダウンコンバータ２３ａ、Ａ／Ｄ変換回路２３ｂでデジタルの複素ベクトル信号に変換された後、伝送された情報符号を復調する信号処理を実施するＦＦＴ(フーリエ変換)回路２３ｃ、伝送路応答等化回路２３ｄ、復調＆復号化回路２３ｅ等からなる本線系の経路に入力されるとともに、別経路にあるガード相関算出回路２４に入力される。
そして、ガード相関算出回路２４で、受信信号のガード期間における相関演算を行い、ガード相関信号を受信レベル算出回路２５に出力する。受信レベル算出回路２５は、ガード相関信号から受信信号のレベルを算出して方向調整信号発生回路２６に出力する。
方向調整信号発生回路２６は、算出した受信信号レベルに基づき、受信アンテナ２１の方向調整用信号を生成し、生成した方向調整用信号を用いて受信アンテナ２１の方向調整を行うようにしたものである。
The information code input to the transmission preprocessing circuit 13a of the transmission device 12 is converted into a signal sequence of a frequency distribution image representing a signal of each carrier by preprocessing such as conversion to an error correction code and modulation to 64QAM, An inverse Fourier transform (IFFT) circuit 13b converts the signal into a signal sequence representing a time waveform. The guard interval insertion circuit 13c adds a guard interval to the transmitted OFDM signal in order to reduce the influence of intersymbol interference on the receiving side caused by the delayed wave in the transmission path. The signal with the guard interval inserted is further subjected to post-processing such as quadrature modulation, D / A conversion, and up-conversion in the post-transmission processing circuit 13d, and transmitted from the transmission antenna 11.
The transmitted OFDM transmission signal is received by the receiving antenna 21 and sent to the demodulator 22 through the cable. The received signal input to the demodulator 22 is converted into a digital complex vector signal by the down converter 23a and the A / D conversion circuit 23b, and then subjected to signal processing for demodulating the transmitted information code (FFT) ) Is input to a main line path including a circuit 23c, a transmission path response equalization circuit 23d, a demodulation & decoding circuit 23e, and the like, and is input to a guard correlation calculation circuit 24 in another path.
Then, the guard correlation calculation circuit 24 performs correlation calculation in the guard period of the received signal, and outputs the guard correlation signal to the reception level calculation circuit 25. The reception level calculation circuit 25 calculates the level of the reception signal from the guard correlation signal and outputs it to the direction adjustment signal generation circuit 26.
The direction adjustment signal generation circuit 26 generates a direction adjustment signal for the reception antenna 21 based on the calculated reception signal level, and adjusts the direction of the reception antenna 21 using the generated direction adjustment signal. is there.
従来の受信側の検波器により受信電界レベルを検出して、受信電界レベルが最大になる方向を探索するアンテナの方向調整方法では、受信電界が低くなると、受信信号は雑音に埋もれてしまい、精度の良い検出が困難であった。 例えば、受信電界が−９０ｄＢｍ、伝送帯域幅が１７ＭＨｚ、初段増幅器の雑音指数が４ｄＢの時にはＣＮ比が−７ｄＢ程度となり、受信電界が−９０ｄＢｍを下回る様なレベルの受信信号を検出することは困難であった。また、図３のアンテナ方向調整方法を用いたとしても、受信電界が−９０ｄＢｍを大きく下回るようなレベルの受信信号を検出することは困難であった。
即ち、アンテナの方向調整の初期段階では受信電界は非常に低いため、ＯＦＤＭ信号が僅かながら到達していたとしても、雑音に埋もれた受信信号を捕らえることができず、手探りで方向調整を行わざるを得なかった。
In the conventional antenna direction adjustment method in which the received electric field level is detected by a detector on the receiving side and the direction in which the received electric field level is maximized is detected, the received signal is buried in noise when the received electric field is lowered. The good detection of was difficult. For example, when the received electric field is -90 dBm, the transmission bandwidth is 17 MHz, and the noise figure of the first stage amplifier is 4 dB, the CN ratio is about -7 dB, and it is difficult to detect a received signal at a level such that the received electric field is less than -90 dBm. Met. Further, even when the antenna direction adjustment method of FIG. 3 is used, it is difficult to detect a received signal at a level such that the received electric field is significantly below −90 dBm.
In other words, since the received electric field is very low at the initial stage of antenna direction adjustment, even if the OFDM signal arrives a little, the received signal buried in noise cannot be captured, and direction adjustment must be carried out by groping. Did not get.
本発明は上記課題を解決するために、ガードインターバル期間を含んだＯＦＤＭ信号を伝送する伝送装置の受信装置において、受信した上記ＯＦＤＭ信号と該ＯＦＤＭ信号を有効シンボル期間長の遅延を行った信号との複素乗算を行い、当該複素乗算の結果得られる複素乗算信号の少なくともガードインターバル期間の信号に対して平均化処理を行い、受信電界レベルに対応する信号を生成し、該生成した信号に基づき受信アンテナの方向調整を行うようにしたものである。
また、平均化処理により得られた受信電界レベルに対応する信号に対して絶対値演算処理を行い、当該絶対値演算の結果得られた信号レベルに基づき受信アンテナの方向調整用信号を生成し、該生成した方向調整用信号を用いて受信アンテナの方向調整を行うようにしたものである。
また、生成される方向調整用信号を、受信信号のレベルに応じて音質、音量、音の断続間隔の少なくとも１つが変化する方向調整用の音の信号、あるいは色、輝度、点滅間隔の少なくとも１つが変化する方向調整用の光の信号、あるいは数値、文字、グラフの少なくとも１つを表示する信号の内の少なくとも何れか１つの信号としたものである。
また、ガードインターバル期間を含んだＯＦＤＭ信号を伝送する伝送装置の受信装置において、受信した上記ＯＦＤＭ信号と該ＯＦＤＭ信号を有効シンボル期間長の遅延を行った信号との複素乗算を行う複素乗算演算手段と、当該複素乗算の結果得られる複素乗算信号の少なくともガードインターバル期間の信号に対して低域通過フィルタ演算を行うフィルタ演算手段を具備することにより、受信電界レベルに対応する信号を算出するＯＦＤＭ受信装置である。
また、フィルタ演算により得られた信号に対して絶対値演算処理を行い、該絶対値演算の結果得られた信号レベルに基づき受信アンテナの方向調整用信号を生成する方向調整信号生成手段を有し、該生成した方向調整用信号を用いて受信アンテナの方向調整を行うＯＦＤＭ受信装置である。
In order to solve the above-described problems, the present invention provides a receiving apparatus of a transmission apparatus that transmits an OFDM signal including a guard interval period, and the received OFDM signal and a signal obtained by delaying the OFDM signal by an effective symbol period length; The complex multiplication signal obtained as a result of the complex multiplication is averaged for at least the guard interval period signal to generate a signal corresponding to the received electric field level, and reception is performed based on the generated signal. The antenna direction is adjusted.
Further, an absolute value calculation process is performed on the signal corresponding to the received electric field level obtained by the averaging process, and a signal for adjusting the direction of the receiving antenna is generated based on the signal level obtained as a result of the absolute value calculation, The direction adjustment of the receiving antenna is performed using the generated direction adjustment signal.
Further, the generated direction adjustment signal is a direction adjustment sound signal in which at least one of sound quality, volume, and sound intermittent interval changes according to the level of the received signal, or at least one of color, luminance, and blinking interval. This is a light signal for adjusting the direction in which one of the signals changes, or at least one of signals indicating at least one of a numerical value, a character, and a graph.
Further, in a receiving apparatus of a transmission apparatus that transmits an OFDM signal including a guard interval period, complex multiplication operation means for performing complex multiplication of the received OFDM signal and a signal obtained by delaying the OFDM signal by an effective symbol period length And OFDM reception for calculating a signal corresponding to the received electric field level by providing a filter operation means for performing a low-pass filter operation on the signal of at least the guard interval period of the complex multiplication signal obtained as a result of the complex multiplication Device.
Also, there is a direction adjustment signal generating means for performing absolute value calculation processing on the signal obtained by the filter calculation and generating a signal for adjusting the direction of the receiving antenna based on the signal level obtained as a result of the absolute value calculation. , An OFDM receiving apparatus that performs direction adjustment of a receiving antenna using the generated direction adjustment signal.
さらには、生成した方向調整用信号を、受信信号のレベルに応じて、音質、音量、音の断続間隔の少なくとも１つが変化する音の信号に変換する手段と、色、輝度、点滅間隔の少なくとも１つが変化する光の信号に変換する手段と、数値、文字、グラフの少なくとも１つを表示する信号に変換する手段の内の少なくとも何れか１つの手段を具備するものである。
また、平均化処理、フィルタ演算を行う期間においては、受信局部発振器の発振周波数が一定になるよう制御を行うものである。
また、方向調整用信号を受信信号の実効値にほぼ比例する信号、あるいは受信信号の電力値にほぼ比例する信号、あるいは受信信号のｄＢ値(対数値)にほぼ比例する信号、あるいは受信アンテナの方向を調整する雲台の方向を制御する制御信号の内のいずれか１つの信号に変換するようにしたものである。
Furthermore, means for converting the generated direction adjustment signal into a sound signal in which at least one of sound quality, volume, and sound intermittent interval changes according to the level of the received signal, and at least color, brightness, and blinking interval It comprises at least one of means for converting to a signal of light, one of which changes, and a means for converting at least one of a numerical value, a character, and a graph.
Further, during the period in which the averaging process and the filter operation are performed, control is performed so that the oscillation frequency of the reception local oscillator becomes constant.
Further, the direction adjustment signal is a signal that is approximately proportional to the effective value of the received signal, a signal that is approximately proportional to the power value of the received signal, a signal that is approximately proportional to the dB value (logarithmic value) of the received signal, or the signal of the receiving antenna. This is converted to any one of the control signals for controlling the direction of the pan head for adjusting the direction.
本発明による受信アンテナの方向調整方法を用いると、受信アンテナの方向調整の初期段階において、受信されるＯＦＤＭ信号のＣＮ比が−７ｄＢ程度であっても、すなわち、例えば、伝送帯域幅が１７ＭＨｚ、初段増幅器の雑音指数が４ｄＢの時には、受信電界が−１０４ｄＢｍ程度であっても、ＯＦＤＭ信号の存在を的確に検出できる様になり、受信アンテナの方向を変えながら、受信されるＯＦＤＭ信号のレベルが最大になる方向を探すことができる。
また、算出した受信レベル信号を用いて、容易に受信アンテナの方向調整ができるシステムを構築することができるようにもなる。
When the direction adjustment method of the receiving antenna according to the present invention is used, even if the CN ratio of the received OFDM signal is about −7 dB in the initial stage of the direction adjustment of the receiving antenna, for example, the transmission bandwidth is 17 MHz, When the noise figure of the first stage amplifier is 4 dB, even if the reception electric field is about −104 dBm, the presence of the OFDM signal can be accurately detected, and the level of the received OFDM signal is changed while changing the direction of the reception antenna. You can find the direction that maximizes.
It is also possible to construct a system that can easily adjust the direction of the receiving antenna using the calculated reception level signal.
以下、本発明による実施例について、図示の実施形態により詳細に説明する。
図１は本発明の第一の実施例について示した図である。前述のように、受信装置に到達したＯＦＤＭ信号は受信アンテナ１で受信され、受信高周波部２で周波数変換してベースバンド信号に変換される。 このベースバンド信号は、Ａ／Ｄ変換回路３に入力され受信サンプリング系列Ｚin(ｍ)を得る(ｍはサンプル番号)。 その後、伝送された情報符号を復調するためのＦＦＴ(高速フーリエ変換)回路６、復調部７等からなる本線系の経路を経由して復調信号Ｄを外部装置に出力する。
また、受信サンプリング系列Ｚin(ｍ)は、クロックタイミングやキャリア周波数を送信信号に同期させる同期処理部８に入力され、同期信号ＳＹＮＣは受信タイミングを制御する信号としてＯＦＤＭ受信部全体に配信される。
さらに、受信サンプリング系列Ｚin(ｍ)は、これらの接続と共に、受信電界を算出する受信電界算出部４に接続され、受信電界算出部４で得られた受信電界信号Ｒは、方向調整信号発生部５に接続され、受信アンテナ１の方向調整用の信号Ｃを生成する。
Hereinafter, examples according to the present invention will be described in detail with reference to the illustrated embodiments.
FIG. 1 is a diagram showing a first embodiment of the present invention. As described above, the OFDM signal that has arrived at the receiving apparatus is received by the receiving antenna 1, and is converted into a baseband signal by frequency conversion by the receiving high-frequency unit 2. This baseband signal is input to the A / D conversion circuit 3 to obtain a received sampling sequence Zin (m) (m is a sample number). Thereafter, the demodulated signal D is output to an external device via a main line path including an FFT (Fast Fourier Transform) circuit 6 and a demodulator 7 for demodulating the transmitted information code.
The reception sampling sequence Zin (m) is input to the synchronization processing unit 8 that synchronizes the clock timing and the carrier frequency with the transmission signal, and the synchronization signal SYNC is distributed to the entire OFDM reception unit as a signal for controlling the reception timing.
Further, the received sampling sequence Zin (m) is connected to the received electric field calculating unit 4 for calculating the received electric field together with these connections, and the received electric field signal R obtained by the received electric field calculating unit 4 is the direction adjustment signal generating unit. 5 to generate a signal C for adjusting the direction of the receiving antenna 1.
次に、本発明の目的となる受信電界算出部４の構成について更に詳しく述べる。
図２は受信電界算出部４の構成について示した図である。Ａ／Ｄ変換回路３からの受信サンプル系列Ｚin(ｍ)は、有効シンボル遅延器４１に接続され、有効シンボル遅延された受信サンプル系列Ｚin(ｍ−Ｍ)出力は、複素乗算器４２に接続される。複素乗算器４２のもう一方の入力端子には、Ａ／Ｄ変換回路３からの受信サンプル系列Ｚin(ｍ)信号が接続される。 後述のように複素乗算器４２の出力信号は、積分器４３に接続され、絶対値器４４を経由して受信電界信号Ｒとして出力する。
この受信電界算出部４では、ＯＦＤＭのガードインターバル信号の相関性を利用した処理を施しているため、受信電界算出部４の動作を説明する前に、ガードインターバル信号を含むＯＦＤＭ信号について図４を用いて説明する。
ＯＦＤＭ方式は、一定の周波数間隔で配置された数百〜数千本の搬送波(キャリア)を、それぞれ一定のシンボル期間Ｔｓ’でデジタル変調して伝送する方式である。 ＯＦＤＭ信号への変調には、通常、ポイント数Ｍ(例えば、Ｍ＝１０２４)のＩＦＦＴ（逆高速フーリエ変換）が用いられる。 送信側から送出される伝送信号の１シンボルは、ＩＦＦＴで変調されたＭポイントのＯＦＤＭ信号からなる有効シンボル期間Ｔｓの信号(Ａ＋ａ)と、１シンボルの最後のＭｇ(例えばＭｇ＝１２８)ポイント期間Ｔｇの信号ａを有効シンボル期間Ｔｓ前のガード期間Ｔｇに複写したＭｇポイントのガードインターバル信号ａ’で構成される。なお、次のシンボルのｂとｂ’の部分についても同様である。
Next, the configuration of the received electric field calculation unit 4 that is an object of the present invention will be described in more detail.
FIG. 2 is a diagram showing the configuration of the received electric field calculation unit 4. The received sample sequence Zin (m) from the A / D conversion circuit 3 is connected to the effective symbol delay unit 41, and the output of the received sample sequence Zin (m−M) delayed by the effective symbol is connected to the complex multiplier 42. The The reception sample sequence Zin (m) signal from the A / D conversion circuit 3 is connected to the other input terminal of the complex multiplier 42. As will be described later, the output signal of the complex multiplier 42 is connected to the integrator 43 and is output as the received electric field signal R via the absolute value unit 44.
Since the reception electric field calculation unit 4 performs processing using the correlation of the OFDM guard interval signal, before explaining the operation of the reception electric field calculation unit 4, FIG. 4 shows an OFDM signal including the guard interval signal. It explains using.
The OFDM scheme is a scheme in which hundreds to thousands of carriers (carriers) arranged at a certain frequency interval are digitally modulated and transmitted at a certain symbol period Ts ′. For modulation to an OFDM signal, IFFT (Inverse Fast Fourier Transform) with M points (for example, M = 1024) is usually used. One symbol of a transmission signal transmitted from the transmission side includes a signal (A + a) of an effective symbol period Ts composed of an M-point OFDM signal modulated by IFFT, and the last Mg (for example, Mg = 128) point period of one symbol. It is composed of an Mg point guard interval signal a ′ obtained by copying the Tg signal a in the guard period Tg before the effective symbol period Ts. The same applies to the parts b and b ′ of the next symbol.
以上の知識を基に、図４の受信電界算出部４で実施される処理について、図５を用いて説明する。 Ａ／Ｄ変換回路３でサンプリングされ、受信電界算出部４に入力された図５（ａ）の受信サンプリング系列Ｚin(ｍ)は、有効シンボル遅延器４１にて、図５（ｂ）のように、有効シンボル期間Ｔｓに相当するサンプリング数Ｍ(例えば、Ｍ＝１０２４)だけ遅延される。 この有効シンボル期間Ｔｓだけ遅延された信号Ｚin(ｍ−Ｍ)と、遅延前の信号Ｚin(ｍ)は、複素乗算器４２でサンプル点毎に複素乗算され、
Ｚmul(ｍ)＝Ｚin(ｍ)×Ｚin(ｍ−Ｍ)* ・・・・・・・・・・・（１）
が算出される。この複素乗算信号の波形を、図５（ｃ）、（ｄ）に示す。
図５（ｃ）は、Ｃ／Ｎが無限大の時の様子を示した図である。ここで、ガードインターバル信号であるａとａ’、ｂとｂ’（図中の点線）の信号は歪みの無い理想的な条件下では同じ信号成分であるため、このガードインターバル期間では相関性を有することになり、他の期間に対して相関レベルが大きくなる。
Based on the above knowledge, the process performed by the received electric field calculation unit 4 in FIG. 4 will be described with reference to FIG. The reception sampling sequence Zin (m) of FIG. 5A sampled by the A / D conversion circuit 3 and input to the reception electric field calculation unit 4 is output by the effective symbol delay unit 41 as shown in FIG. 5B. The sampling number M corresponding to the effective symbol period Ts is delayed by M (for example, M = 1024). The signal Zin (m−M) delayed by the effective symbol period Ts and the signal Zin (m) before the delay are complex-multiplied at each sample point by the complex multiplier 42.
Zmul (m) = Zin (m) × Zin (m−M) * (1)
Is calculated. The waveforms of this complex multiplication signal are shown in FIGS.
FIG. 5C is a diagram showing a state when C / N is infinite. Here, the guard interval signals a and a ′ and b and b ′ (dotted lines in the figure) are the same signal components under ideal conditions without distortion. Therefore, the correlation level increases with respect to other periods.
受信アンテナ１の方向調整の初期段階では、受信装置のＬｏ周波数(局部発振周波数)の同期が確立されていないことが多く、図５（ｃ）のＩ成分とＱ成分で構成される複素ベクトル信号は、任意の方向に回転された信号になる。Ｌｏ周波数が確立すると、複素ベクトル信号の位相は０°となり、Ｉ成分のみに相関成分が現れる。
ここで、この相関レベルは、受信サンプリング系列Ｚin(ｍ)のガードインターバル期間(ａとａ’、ｂとｂ’)に含有する雑音成分が多くなると、雑音成分の無相関性のために、レベルが小さくなる。 従って、ガードインターバル期間の相関性が低ければ、受信サンプリング系列Ｚin(ｍ)に含有する雑音成分が多いこと、即ち、受信電界レベルが低いことを意味している。逆に、相関レベルが大きくなると、受信電界レベルが大きいことを意味する。
In the initial stage of the direction adjustment of the receiving antenna 1, the synchronization of the Lo frequency (local oscillation frequency) of the receiving apparatus is often not established, and the complex vector signal composed of the I component and the Q component in FIG. Becomes a signal rotated in an arbitrary direction. When the Lo frequency is established, the phase of the complex vector signal becomes 0 °, and the correlation component appears only in the I component.
Here, when the noise component contained in the guard interval period (a and a ′, b and b ′) of the received sampling sequence Zin (m) increases, the correlation level is increased due to the non-correlation of the noise component. Becomes smaller. Therefore, if the correlation of the guard interval period is low, it means that there are many noise components contained in the received sampling sequence Zin (m), that is, the received electric field level is low. Conversely, when the correlation level increases, it means that the received electric field level is high.
しかし、例えばＣ／Ｎが−５ｄＢを下回るような環境では含有する雑音量も多く、図５の（ｄ）のように、波形からガードインターバル期間とそれ以外の期間の相関レベル差を判断できるような相関波形を得ることは困難である。また、その様な環境下では、シンボルタイミングを検出するシンボル同期も確立していないため、ガードインターバルタイミングを判断することは困難である。
そのため本発明では、複素乗算器４２の出力信号Ｚmul(ｍ)のＩ成分、Ｑ成分それぞれに対して、積分器４３で平均化処理を施し、擾乱成分を抑圧して、有効な相関成分を抽出する。 即ち、含有する雑音量が多くなると、図５の（ｄ）のように、複素乗算器４２の出力信号Ｚmul(ｍ)自体では、ガードインターバル期間とそれ以外の期間の相関レベル差を判断できるような相関波形を得ることがむずかしくなるが、複素乗算器４２の出力信号Ｚmul(ｍ)を、積分器４３で積分(平均化処理)することにより、ガードインターバル期間以外の期間は無相関のため、ガードインターバル期間以外の積分結果は０に収束する。
一方、ガードインターバル期間については、相関性を有するため、積分処理を施すことで、Ｃ／Ｎに依存した相関レベルを得ることができる。
However, for example, in an environment where C / N is less than −5 dB, the amount of noise contained is large, and as shown in FIG. 5D, the correlation level difference between the guard interval period and other periods can be determined from the waveform. It is difficult to obtain a simple correlation waveform. In such an environment, since symbol synchronization for detecting symbol timing has not been established, it is difficult to determine the guard interval timing.
Therefore, in the present invention, the integrator 43 performs averaging processing on each of the I component and Q component of the output signal Zmul (m) of the complex multiplier 42, suppresses the disturbance component, and extracts an effective correlation component. To do. That is, when the amount of noise contained increases, as shown in FIG. 5D, the output level Zmul (m) itself of the complex multiplier 42 can determine the correlation level difference between the guard interval period and other periods. However, it is difficult to obtain a correlated waveform, but by integrating (averaging) the output signal Zmul (m) of the complex multiplier 42 by the integrator 43, the period other than the guard interval period is uncorrelated. Integration results outside the guard interval period converge to zero.
On the other hand, since the guard interval period has a correlation, a correlation level depending on C / N can be obtained by performing an integration process.
更に、ＧＰＳ等を利用し、受信装置側で絶対的な時間を有し、ガードインターバル期間を把握できるようなシステムにおいては、積分期間をガードインターバル期間に限定することで、擾乱成分の抑圧効果を向上させることができる。
なお、積分器４３の構成については、ＦＩＲフィルタ等により構成される移動平均処理、即ち低域通過フィルタ(以下ＬＰＦ：Low Pass Filter)や、ＩＩＲフィルタにより構成されるＬＰＦにより実現することができる。このＬＰＦの時定数に関しては、アンテナの方向調整制御に素早く追従できる時定数として、数百ｍｓｅｃ以内が望ましい。
また、積分処理を施す際には、受信Ｌｏ周波数は一定に保つ必要がある。前述したように、同期が確立していない場合には、複素乗算器４２の出力Ｚmul(ｍ)の位相は不確定となる。 ここで、受信Ｌｏ周波数が時間的に変動すると、図６に示すようにＺmul(ｍ)の位相も変動してしまう。 このような信号に対して積分処理を行った場合、Ｚmul(ｍ)の有益な相関成分さえも平均化され、０に収束させてしまう。そのため、受信Ｌｏ周波数を一定に保持し、Ｚmul(ｍ)の位相を一定にする制御が必要となる。
Furthermore, in a system that uses GPS or the like and has an absolute time on the receiving device side and can grasp the guard interval period, by limiting the integration period to the guard interval period, the effect of suppressing disturbance components can be reduced. Can be improved.
The configuration of the integrator 43 can be realized by a moving average process configured by an FIR filter or the like, that is, by an LPF configured by a low pass filter (hereinafter referred to as LPF) or an IIR filter. With respect to the time constant of this LPF, it is desirable that it is within several hundreds msec as a time constant that can quickly follow the antenna direction adjustment control.
In addition, when the integration process is performed, it is necessary to keep the reception Lo frequency constant. As described above, when synchronization is not established, the phase of the output Zmul (m) of the complex multiplier 42 is indeterminate. Here, when the reception Lo frequency varies with time, the phase of Zmul (m) also varies as shown in FIG. When integration processing is performed on such a signal, even the useful correlation component of Zmul (m) is averaged and converged to zero. For this reason, it is necessary to perform control to keep the reception Lo frequency constant and make the phase of Zmul (m) constant.
この様にして、積分器４３で得られたＺmul(ｍ)の平均値は絶対値器４４に入力され、Ｉ成分とＱ成分の絶対値を算出する。絶対値の算出方法としては、式（２）に示すようにＩ成分とＱ成分の二乗和を演算することで、受信電力の二乗に比例する受信電界信号Ｒを算出することができる。
Ｉ2＋Ｑ2 ・・・・・・・・・・・（２）
また、式（３）に示すようにＩ成分とＱ成分の二乗和の平方根を演算することで、受信電力に比例する受信電界信号Ｒを算出することもできる。
(Ｉ2＋Ｑ2)0.5 ・・・・・・・・・・・（３）
ところで、通常、受信装置では、受信条件で大きく変化する受信信号のレベルを、自動利得制御(ＡＧＣ：Automatic Gain Control)回路にてほぼ一定のレベルになるような制御を行った後に、各種の信号処理を実施する方式が用いられている。 そのため、受信電界算出部４に入力される信号Ｚin(ｍ)の電力も、常にほぼ一定に保たれる。
In this way, the average value of Zmul (m) obtained by the integrator 43 is input to the absolute value unit 44, and the absolute values of the I component and the Q component are calculated. As an absolute value calculation method, the received electric field signal R proportional to the square of the received power can be calculated by calculating the sum of squares of the I component and the Q component as shown in the equation (2).
I 2 + Q 2 (2)
In addition, the received electric field signal R proportional to the received power can be calculated by calculating the square root of the square sum of the I component and the Q component as shown in the equation (3).
(I 2 + Q 2 ) 0.5 (3)
By the way, normally, in a receiving apparatus, after performing control so that the level of a received signal that varies greatly depending on reception conditions becomes an almost constant level in an automatic gain control (AGC) circuit, A method of performing processing is used. Therefore, the power of the signal Zin (m) input to the received electric field calculation unit 4 is always kept substantially constant.
このような制御が行われている場合には、受信電界算出部４からの受信電界信号Ｒは、受信電界に比例する値にはならない。 特に、図７に示すように、Ｃ／Ｎが高くなると、即ち受信電界レベルが大きくなると、受信電界信号Ｒのレベルは、ある一定値で飽和してしまう。 しかし、受信電界信号Ｒのレベルが飽和領域にある場合には、同期も確立しており、所定レベル以上の受信電界が得られている状況であるため、アンテナの方向調整が完了したと判断して良い。 When such control is performed, the received electric field signal R from the received electric field calculation unit 4 does not become a value proportional to the received electric field. In particular, as shown in FIG. 7, when the C / N is increased, that is, when the received electric field level is increased, the level of the received electric field signal R is saturated at a certain value. However, when the level of the received electric field signal R is in the saturation region, synchronization is established and a received electric field of a predetermined level or higher is obtained. Good.
前述のようにして、受信電界算出部４にて算出された受信電界信号Ｒは、方向調整信号発生部５に入力され、受信アンテナ１の調整に適した信号に変換され、方向調整信号Ｃとして出力される。 この方向調整信号Ｃとしては、例えば、受信アンテナ１の調整を実施するときは、受信アンテナの調整者が調整し易いように、受信レベルが高いほど高音になる、あるいは受信レベルが高いほど間隔が狭くなる断続音等のビープ音を発生させる方向調整信号を出力するようにする。 あるいは、オシロスコープ等で発生した方向調整信号Ｃの電圧変化を観察しながら、このレベルが最大になるように調整する等、受信レベルを測定しながら受信アンテナの方向を調整する場合は、受信レベル信号の値を、調整が容易になる単位系の値、例えばｄＢ値等の値に変換して得た信号を、方向調整信号として出力するようにする。 あるいは、受信レベルを表示するメータを用い、受信レベルを測定しながら受信アンテナの方向を調整する場合は、受信レベルに応じた明るさ、色または表示バーの長さ等のメータ表示値を制御する信号を、方向調整信号Ｃとして出力するようにする。 あるいは、受信アンテナの雲台を小さく動かした時に、受信レベルが増加するときは、更に同じ方向に移動し、減少するときは反対方向に動かすことにより、最適な方向に受信アンテナを制御する方法を用いる場合は、受信レベル信号の値を雲台の制御が容易になる単位系の値に変換して得た信号を、方向調整信号Ｃとして出力するようにする。 As described above, the reception electric field signal R calculated by the reception electric field calculation unit 4 is input to the direction adjustment signal generation unit 5 and converted into a signal suitable for adjustment of the reception antenna 1, and is used as the direction adjustment signal C. Is output. As the direction adjustment signal C, for example, when the reception antenna 1 is adjusted, the higher the reception level, the higher the sound, or the higher the reception level, the easier the adjustment of the reception antenna. A direction adjustment signal for generating a beep sound such as an intermittent sound that is narrowed is output. Alternatively, when adjusting the direction of the reception antenna while measuring the reception level, such as adjusting the level of the direction adjustment signal C generated by an oscilloscope or the like while observing the voltage change of the direction adjustment signal C, the reception level signal A signal obtained by converting the value of 1 into a unit system value that facilitates adjustment, for example, a value such as a dB value, is output as a direction adjustment signal. Alternatively, when adjusting the direction of the receiving antenna while measuring the reception level using a meter that displays the reception level, control the meter display value such as brightness, color, or display bar length according to the reception level. The signal is output as the direction adjustment signal C. Alternatively, when the receiving antenna head is moved small, if the reception level increases, it moves further in the same direction, and when it decreases, it moves in the opposite direction to control the receiving antenna in the optimal direction. When used, a signal obtained by converting the value of the reception level signal into a unit system value that makes it easy to control the camera platform is output as the direction adjustment signal C.
このように、本実施例によるＯＦＤＭ方式の受信装置を用いると、受信されるＯＦＤＭ信号のＣＮ比が−５ｄＢ以下になり、従来と同じ方法、即ち受信検波レベルを用いる方法とか図３のアンテナ方向調整方法では、ＯＦＤＭ信号の存在すら検出できないような受信アンテナの方向調整の初期段階においても、ＯＦＤＭ信号の存在を検出できるようになるだけでなく、受信されたＯＦＤＭ信号の電力レベルとその変化量を、高いＳＮ比で測定することができるようになる。
そのため、受信アンテナの方向を変えながら、受信されるＯＦＤＭ信号レベルが最大になる方向を探すことができるようになり、算出した方向調整信号を用いて、容易に受信アンテナの方向調整ができるシステムを構築することができるようになる。
As described above, when the OFDM receiver according to the present embodiment is used, the CN ratio of the received OFDM signal becomes -5 dB or less, and the same method as before, that is, the method using the reception detection level, or the antenna direction of FIG. In the adjustment method, not only the presence of the OFDM signal can be detected even in the initial stage of the direction adjustment of the receiving antenna where even the presence of the OFDM signal cannot be detected, the power level of the received OFDM signal and the amount of change thereof. Can be measured with a high S / N ratio.
Therefore, it is possible to find a direction in which the received OFDM signal level becomes maximum while changing the direction of the receiving antenna, and a system that can easily adjust the direction of the receiving antenna using the calculated direction adjustment signal. Will be able to build.
１：受信アンテナ、２：受信高周波部、３：Ａ／Ｄ変換回路、４：受信電界算出部、５：方向調整信号発生部、６：ＦＦＴ、７：復調部、８：同期処理部、４１：有効シンボル遅延器、４２：複素乗算器、４３：積分器、４４：絶対値器。 1: reception antenna, 2: reception high frequency unit, 3: A / D conversion circuit, 4: reception electric field calculation unit, 5: direction adjustment signal generation unit, 6: FFT, 7: demodulation unit, 8: synchronization processing unit, 41 : Effective symbol delay unit, 42: complex multiplier, 43: integrator, 44: absolute value unit.
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