JP4748558B2 - Absolute encoder - Google Patents

Description

【００４１】
演算部１７は、上記の関係を利用してλｍアドレスＡｊを特定するとともに、上記φｍをもとに生成された分解能１μｍのパルス列ＳＡＰｍとにより、１μｍの分解能で１６λｍ区間の絶対位置を生成する。例えば、図１１に示すようにカウンタ機能（タイマー）を備えるマイクロコンピュータ（ＣＰＵ）７３を用いてソフトウエア処理で実現することができる。
【００４２】
ここで、演算部１７における主要な機能であるλｍアドレスＡｊの検出にかかる基本原理について説明する。
【００４３】
この演算部１７は、上記第１のゲート回路１５から出力される１μｍ分解能のパルス列ＳＡＰｍをカウントするパルスカウンタ７１と、第２のゲート回路２５から出力されるパルス列ＳＡＰａをカウントするパルスカウンタ７２を備え、該パルスカウンタ７１，７２の計測値をＣＰＵ７３に取り込むためのトリガ信号として、パルス幅変調信号ＰＷＳｍ，ＰＷＳａのセット入力となる基準信号ＭＯＤＳがタイミング信号生成部１８からＣＰＵ７３への割り込み端子に供給されている。
【００４４】
サンプリング時刻ｔｉにおいて、上記（２）式に示す位相量φｍ（ｔｉ）のリセットタイミングに対応した基準信号ＭＯＤＳが入力されると、ＣＰＵ７３が割り込み動作を開始し、パルスカウンタ７１で計測された第１のスケール１１の波長λｍ内の絶対位置に対応するデータＤｍ（ｔｉ）と、パルスカウンタ７２で計測された上記２つの位相変調信号の位相差Δφに対応したパルス列ＳＡＰｍの計測値ΔＤ（ｔｉ）がＣＰＵ７３に取り込まれ、ＲＡＭ７４内の所定のメモリ領域に格納される。
【００４５】
次に、ＣＰＵ７３は、上記位相差Δφに対応したこれらの計数値の差ΔＤをＲＡＭ７４内の所定の領域に格納するとともに、上記区間パルス数Ｍ／１６で除算し、その商としてλｍアドレスＡｊを特定し、該求められたＡｊをλｍ倍し、さらに、タイミングｔｉにおけるλｍ内の絶対値Ｄｍ（ｔｉ）とを加算し、１６λｍ区間全域に亘る分解能１μｍの絶対位置を生成する。
【００４６】
このようにして生成された絶対位置は、ＲＡＭ７４内の所定のメモリ領域内に格納するとともに、例えば、測定データとして外部に出力する。
【００４７】
ところで、上記の演算部１７の動作は理想的な状態で述べたものであり、実際のシステムにおいては検出された信号の不完全さ、例えば、各々の検出ヘッド１２，２２に重畳するＤＣのずれ、さらには両チャンネルのヘッドから出力される平衡変調信号の９０°位相からのずれ、すなわち、出力の位相ずれ等に起因する検出誤差、すなわち、波長λｍ内を高分解能で検出する際に発生する内挿誤差を内包しており、これらの内挿誤差がλｍアドレスＡｊの切替え部に重畳するため、λｍアドレスのＡｊ判定に誤動作を起こすことがある。
【００４８】
ここで、上述のずれと内挿誤差との関係について数式を用いて説明する。
【００４９】
本発明の具体的な実施の形態として検出ヘッドとしてＭＲセンサを用いた場合について説明してきた。周知のごとく、ＭＲセンサにおいてはリソグラフィーを基本としてマスクを製作するため、両チャンネルのヘッドの間隔を波長λに対して正確に９０°（λ／４）に設定することができる。
【００５０】
したがって、上記に示す誤差要因のうち、「出力の位相ずれ」の影響は無視することができる。
【００５１】
以下、説明を簡単にするため、検出ヘッドに重畳するＤＣのずれと出力レベルのずれによって生ずる内挿誤差について説明する。
【００５２】
上述の（１）式及び（２）式に示した位相変調信号は、２チャンネルの検出ヘッドから理想的な信号が得られることを前提にした理論式である。
【００５３】
しかしながら、実際のシステムでは各の検出ヘッドに重畳するＤＣのずれや出力レベルのずれにより、（１）式や（２）式に示すような完全な位相変調信号とのずれを生じ、結果として内挿誤差を生ずる。
【００５４】
ＤＣのずれと出力レベルのずれが一般式として、波長λのスケールにおけるＣＨ１検出ヘッド及びＣＨ２検出ヘッドから得られる平衡変調信号を各々ｅ１，ｅ２とすると、平衡変調信号ｅ１，ｅ２は、
ｅ１＝｛Ａ＋（１＋ａ）×Ｓｉｎ（２πｘ／λ）｝×Ｃｏｓωｔ （６）
ｅ２＝｛Ｂ＋（１＋ｂ）×Ｃｏｓ（２πｘ／λ）｝×Ｓｉｎωｔ （７）
ただし、
ａ：ＣＨ１出力の基準値からのずれ、ｂ：ＣＨ２出力の基準値からのずれ
Ａ：ＣＨ１ヘッドへのＤＣの重畳、 Ｂ：ＣＨ２ヘッドへのＤＣの重畳
となり、式を簡単にするため、２πＸ／λ＝Ｘ、ωｔ＝Ｔとおけば、（６）式及び（７）式は
ｅ１＝｛Ａ＋（１＋ａ）×ＳｉｎＸ｝×ＣｏｓＴ （６）’
ｅ２＝｛Ｂ＋（１＋ｂ）×ＣｏｓＸ｝×ＳｉｎＴ （７）’
と表すことができる。
【００５５】
次に、単一の誤差要因による場合の内挿誤差パターンについて説明する。
【００５６】
ここでは、説明を簡単にするため、ＤＣのみにずれがあった場合と出力レベルのみにずれがあった場合に分けて説明する。
（１）ＤＣのみにずれがあった時（出力レベルのずれはない）
ＤＣのみにずれがあったときの位相変調信号をｅｐｍ（Ｄ）は、（６）’式及び（７）’式において、ａ＝ｂ＝０とおいて加算した信号であり、次のように表すことができる。
【００５７】

Ａ及びＢが１より十分小さいとして近似式を求めると、

ただし、
δ＝ｔａｎ^−１（Ａ／Ｂ）
θ＝ｔａｎ^−１｛（Ａ＋ＳｉｎＸ）／（Ｂ＋ＣｏｓＸ）｝ （９）
ここで、（８）式の｛ ｝の中は、位相変調信号ｅｐｍ（Ｄ）のエンベロープを表しており、基準振幅１に対しＸの基準位置から位相がδだけ遅れた振幅が√（Ａ^２＋Ｂ^２）のリップルが重畳した、Ｘに対して一次の成分を有する正弦波状の信号であることが分かる。
【００５８】
また、（９）式のθは変位量を表しているので、誤差がないときの理論値｛θ＝ｔａｎ^−１（ＳｉｎＸ／ＣｏｓＸ）｝から減算すると、位相変調信号が理想からずれた、すなわち、ＤＣのずれがあった時の誤差を表すことになる。
【００５９】
今、Ｘに対して０から２πにわたる１波長内の内挿誤差をΔＸ（Ｄ）とおくと、内挿誤差をΔＸ（Ｄ）は、
ΔＸ（Ｄ）＝ｔａｎ^−１｛（Ａ＋ＳｉｎＸ）／（Ｂ＋ＣｏｓＸ）｝
−ｔａｎ^−１（ＳｉｎＸ／ＣｏｓＸ） （１０）
であるから、ここで、
α＝｛（Ａ＋ＳｉｎＸ）／（Ｂ＋ＣｏｓＸ）｝、
β＝ＳｉｎＸ／ＣｏｓＸ
とおけば
ｔａｎ^−１α−ｔａｎ^−１β＝ｔａｎ^−１｛（α−β）／（１＋α×β）｝ゆえ、
ΔＸ（Ｄ）＝ｔａｎ^−１｛（Ａ×ＣｏｓＸ−Ｂ×ＳｉｎＸ）／（１＋Ａ×ＳｉｎＸ＋Ｂ×ＣｏｓＸ）｝
となる。さらに、（Ａ×ＳｉｎＸ＋Ｂ×ＣｏｓＸ）が１より十分小さいことを考慮してΔＸ（Ｄ）の近似式を求めると、
ΔＸ（Ｄ）＝ｔａｎ^−１（Ａ×ＣｏｓＸ−Ｂ×ＳｉｎＸ） （１１）
となる。
【００６０】
さらに、微小角ではｔａｎ｛ΔＸ（Ｄ）｝＝ΔＸ（Ｄ）と近似できる。
【００６１】
したがって、ＤＣのみにずれが生じたときの誤差の近似式は次のよう表すことができる。
【００６２】

ただし、
γ＝ｔａｎ^−１（Ａ／Ｂ） （１３）
（１２）式からも明らかなように、ＤＣのみがずれた時の内挿誤差は、振幅がずれの大きさに比例し、かつ振幅の最大（最小）となる位置が（以下、位相）がずれによって変化するＸの１周期、すなわち波長λに対して１周期の正弦波となることがわかる。
【００６３】
（２）出力レベルのみにずれがあったとき（ＤＣのずれはない）
出力レベルのみにずれがあったときの位相変調信号ｅｐｍ（Ｇ）は、（６）’式及び（７）’において、Ａ＝Ｂ＝０とおいて加算した信号であり、簡単のためａ＝０とおき、ｂをＣＨ１側出力に対する変化分とすれば、次のように表すことができる。
【００６４】

ここで、ｂが１より十分小さいとして近似値を求めると、

ただし、
θ’＝ｔａｎ^−１｛ＳｉｎＸ／（１＋ｂ）×ＣｏｓＸ｝ （１５）
ここで、（１４）式の｛ ｝の中は、位相変調信号ｅｐｍ（Ｇ）のエンベロープを表しており、基準振幅（１＋ｂ／２）に対し、ｂの符号、すなわち、ＣＨ１とＣＨ２の出力の大小関係によって、極性が異なる振幅が（ｂ／２）で、Ｘに対して２次の成分を有する正弦波状の信号が重畳している事が分かる。
【００６５】
また、（１５）式のθ’は変位量を表しているので、誤差がないときの理論値｛θ＝ｔａｎ^−１（ＳｉｎＸ／ＣｏｓＸ）｝から減算すると、位相変調信号の理想からずれた、すなわち、両チャンネルの出力にずれがあった時の内挿誤差を表すことになる、今、Ｘに対して０から２πにわたる１波長内の内挿誤差をΔＸ（Ｇ）とおくと、内挿誤差ΔＸ（Ｇ）は、
ΔＸ（Ｇ）＝ｔａｎ^−１｛ＳｉｎＸ／（１＋ｂ）×ＣｏｓＸ｝−ｔａｎ^−１（ＳｉｎＸ／ＣｏｓＸ）
であり、ここで、
α’＝｛ＳｉｎＸ／（１＋ｂ）×ＣｏｓＸ｝、
β’＝（ＳｉｎＸ／ＣｏｓＸ）
とおけば、
ΔＸ（Ｇ）＝ｔａｎ^−１（α’−β’）／（１＋α’×β’）
ゆえ、

となる。
【００６６】
さらに、ｂが１に対して十分小さいときは、ｔａｎ｛ΔＸ（Ｄ）｝＝ΔＸ（Ｄ）に、分母は１と近似できる。
【００６７】
したがって、出力レベルにずれが生じたときの誤差の近似式は次のように表すことができる。
【００６８】
ΔＸ（Ｇ）＝（−ｂ／２×Ｃｏｓ２Ｘ） （１７）
（１７）式より明らかなように、出力レベルのみにずれが生じた時の内挿誤差ΔＸ（Ｇ）は、その振幅とその極性（出力の大小関係）が、ｂの絶対値と極性（両チャンネルの出力の大小関係）に比例し、位相が一定で、波長λの１周期に対して２周期の成分を有する正弦波状に変化する信号であることが分かる。
【００６９】
単一誤差要因の場合の波長λの１周期内における振動の周期と極大（山）と極小（谷）の現れる位置を次の表２に示してある。この表２から各々の要因に対して独立なことが分かる。
【００７０】
【表２】

【００７１】
次に、２つ以上の誤差要因が重畳した場合について説明する。
【００７２】
同様に、２つ以上の誤差要因が重畳した場合としては、２つのヘッドにＤＣのずれが発生した場合と、いずれかのヘッドに対するＤＣのずれと出力レベルのずれが生じた場合である。
【００７３】
（１）ＣＨ１ヘッドの正負のＤＣずれ（Ａ＞０，Ａ＜０）とＣＨ２ヘッドの正負のＤＣずれ（Ｂ＞０，Ｂ＜０）とが重畳する４通りであり、これらが重畳したときの内挿誤差は、波長λに対して１周期の成分を持ち、各々の組合せにより極大（山）と極小（谷）の位置が変化する。簡単のために係数を１とおいた時の内挿誤差の例を図１２〜図１５に示す。
【００７４】
（２）ＣＨ１ヘッドの正負のＤＣずれと出力レベルのずれが重畳する４通りであり、上記同様に係数を１とおいた時の内挿誤差の一例を図１６に示す。波長λに対して２周期成分を持つが、一組の極大（山）と極小（谷）の位置が各々の組合せによって変化する。
【００７５】
（３）ＣＨ２ヘッドの正負のＤＣずれと出力レベルのずれが重畳する４通りであり、上記同様に係数を１とおいた時の内挿誤差の一例を図１７に示す。（２）と同様、波長λに対して２周期の成分を持っており、組合せによって極大（山）と極小（谷）の位置が変化し、かつ、（２）とは独立である。
【００７６】
次に、３つ以上の要因が重畳した場合について説明する。
【００７７】
３つ以上の誤差要因の重畳としては、出力レベルのずれと２つのヘッドのＤＣずれが同時に発生した場合である。
【００７８】
（１）ＣＨ１の出力がＣＨ２の出力より小さい状態で、ＣＨ１ヘッド及びＣＨ２ヘッドの正負のＤＣずれが重畳する４通りであり、図１８に内挿誤差の一例を示す。波長λ内にただ１つの極大（山）又は極小（谷）を持ち、その位置が各々の組合せにより変化する。
【００７９】
（２）ＣＨ１の出力がＣＨ２の出力より大きい状態で、ＣＨ１ヘッド及びＣＨ２ヘッドの正負のＤＣずれが重畳した場合の４通りにおける内挿誤差の一例を図１９に示す。上記同様、波λ内にただひとつの極大（山）又は極小（谷）をもち、かつ、上記（２）とは独立である。
【００８０】
次の表３は、２つ以上の誤差要因による内挿誤差の周期と、極大（山）と（谷）の位置を一覧表に示したものである。
【００８１】
【表３】

【００８２】
以上より、位相変調信号が理想状態からずれたときには、位相変調信号のエンベロープに対し、ずれの要因に応じて固有の周期と位相を持つリップル成分が重畳し、かつ、ずれの要因に固有な周期と位相をもつ内挿誤差が発生することが分かる。
【００８３】
これらからも明らかなように、λｍアドレスＡｊの切替え部は第１のスケール１１の波長λｍの整数倍に対応した位置であるが、第２のスケール２１の波長λａに対しては、λｍアドレスＡｊごとに、波長λａの１／１６の整数倍ずつずれた位置に対応しているため、λｍアドレス切替え部において検出されるΔφには、第２のスケール２１の波長λａの内挿誤差が重畳するため位相差Δφの逆転現象が生じ、上記表１に示した関係をもとに、単純に検出されたΔφを区間位相量φＺ（＝２π／１６）に対する商を求めるだけでは、λｍアドレスＡｊの検出に誤動作を生じることになる。
【００８４】
図２０は、第２のスケール２１のＣＨ１検出ヘッド及びＣＨ２検出ヘッドに正方向のＤＣのずれが生じた時、すなわち、（１２）式に示す内挿誤差を波長λａの１周期にわたってプロットしたものであり、次の表４は、内挿誤差が重畳するλｍアドレスＡｊとの対応関係を示したものである。
【００８５】
【表４】

【００８６】
次に、本発明において適用される実際のシステム、すなわち、内挿誤差が発生した場合でも正しくλｍアドレスＡｊを検出可能とする演算部１７の構成及び動作について詳細に説明する。
【００８７】
先ず、図２１を参考にしながら、本発明におけるλｍアドレスＡｊの確定にかかる基本的な原理を説明する。
【００８８】
（１）λｍアドレスＡｊの切替え部、すなわち、λｍの両端部において検出される位相差Δφは、第２のスケール２１の波長λａの内挿誤差Ｉｅａに対応して位相量φｉｅａだけ変動する。したがって、実際にはλｍアドレスＡｊがｊとして検出されるべきものが、その前後、すなわち、ｊ−１又はｊ＋１として検出される可能性がある。
【００８９】
（２）しかしながら、上記誤検出される可能性を有する領域に対し、さらに第１のスケールにおける内挿誤差Ｉｅｍに対応する位相量φｉｅｍだけ内側の領域においては、内挿誤差の影響を受けず、λｍアドレスＡｊを一義的に決定できる領域がある。
【００９０】
（３）さらに、上記（１）においても、第１のスケールにおいて検出された位相φｍが、波長λｍ内のどの位置に属するかを、例えば、λｍの中央値λｍ／２に対して左側（ｘ≦λｍ／２）にあるか、又は右側（ｘ＞λｍ／２）に位置するかの判定条件を加え、該判定結果をもとに補正することにより、正しくλｍアドレス（Ａｊ）を判定できる。
【００９１】
すなわち、λｍアドレスＡｊの切替え部において、λｍアドレス（Ａｊ）の判定領域にヒステリシスを持たせ、該ヒステリシス領域内においては、第１のスケールλｍの位置による判定条件を付加することにより、内挿誤差の影響を受けず誤動作のないλｍアドレスＡｊの検出が実現できる。
【００９２】
次に、本発明におけるλｍアドレスＡｊの切替え部に付与すべきヒステリシス及びその他の判定条件の設定について、具体的な数値を適用し説明する。
【００９３】
上述のように、位相差Δφはパルス数ΔＤとして検出され、区間位相量φＺ（＝２π／１６）は区間パルス数ＮＤ（＝６４）に対応している。
【００９４】
第１のスケール及び第２のスケールの波長が略等しいので、システムにおいて想定される内挿誤差が略等しいとして、その振幅をＩｅとすれば、内挿誤差に対応する位相量φｉｅ及びパルス数Ｄｉｅは次のように表すことができる。
【００９５】
φｉｅ＝２π×（Ｉｅ／λｍ） （１７）
Ｄｉｅ＝Ｍ×φｉ／２π＝１，０２４×（Ｉｅ／λｍ） （１８）
すなわち、λｍアドレス（Ａｊ）の切替え部においては、パルス数ΔＤはＤｉｅずつ変動する可能性がある。
【００９６】
ここで、波長λに対する内挿誤差率Ｉｅ／λｍは通常１％程度であり、そのときのＤｉｅは、（１８）式から約１０パルスであり、λｍアドレスＡｊの切替え部においては、第２のスケールλａの内挿誤差により、区間パルス数ＮＤ（＝６４）に対して±１０パルスの変動が生ずると考えることができる。さらに、第１のスケール１１における内挿誤差の影響に伴うパルス数Ｄｉｅも１０パルスとすれば、この値を加えた区間パルス数ＮＤ（＝６４）に対して±２０パルス分内側の領域、すなわち、パルス数ΔＤが２０以上（＝０＋２０）から４４未満（＝６４−２０）の区間は、内挿誤差の影響を受けず一義的にλｍアドレスＡｊを決定できる領域（以下、無条件判別領域）である。
【００９７】
また、上記無条件判別領域の外側、すなわち、ΔＤが０以上２０未満の下側領域及びΔＤが４４以上６４未満の上端領域においては、第１のスケールにおいて検出される位相量φｍが、λｍの中間値、すなわち、Ｄｍが５１２パルス（１，０２４／２）位置を基準にして大小を判別する条件を付加し、該判定結果に応じて仮決定されたλｍアドレスＡｊを補正すれば良い。
【００９８】
次に、図２２に示す処理フローに従い、本発明におけるλｍアドレス（Ａｊ）の判定手順を説明する。
【００９９】
先ず、最初のステップＳ１では、サンプリング時刻ｔｉにおいて、基準信号ＭＯＤＳが入力されると、ＣＰＵ７３が割り込み動作を開始し、上記第１のパルスカウンタ７１の計数値Ｄｍ（ｔｉ）及び第２のパルスカウンタ７２の計数値ΔＤ（ｔｉ）を取り込み、ＲＡＭ７４の所定のメモリ領域に格納する。
【０１００】
次のステップＳ２において、ＣＰＵ７３は、上記ΔＤを区間パルス数ＮＺ（＝６４）で除算し、商ｊをλｍアドレスの候補として、また、剰余Ｒｅｓをλｍアドレス確定用のデータとしてＲＡＭ７４の所定のメモリ領域に格納する。
【０１０１】
次のステップＳ３において、ＣＰＵ７３は、剰余Ｒｅｓの値に応じて３つの領域に分別し、次の判定処理によりλｍアドレスＡｊを確定する。
【０１０２】
すなわち、０≦Ｒｅｓ＜２０のときには（ステップＳ４１）、Ｄｍ（ｔｉ）が中央値５１２を超えているか否かについての判定を行う（ステップＳ５１）。
【０１０３】
そして、このステップＳ５１における判定結果が”Ｙｅｓ”すなわちＤｍ（ｔｉ）≧５１２である場合は、上記ステップＳ２において算出された商ｊから１を減算し、この値をλｍアドレスＡｊとし（ステップＳ６１）、また、判定結果が”Ｎｏ”すなわちＤｍ（ｔｉ）＜５１２である場合には、上記ステップＳ２で検出されたＡｊをそのままλｍアドレスＡｊとする（ステップＳ６２）。
【０１０４】
また、２０≦Ｒｅｓ＜４４のときには（ステップＳ４２）、ステップＳ２で検出された商ｊをそのままλｍアドレスＡｊとする（ステップＳ６２）。
【０１０５】
さらに、４４≦Ｒｅｓ＜６４のときには（ステップＳ４３）、Ｄｍ（ｔｉ）が中央値５１２より小さいか否かについて判定を行う。
【０１０６】
そして、このステップＳ５３における判定結果が”Ｙｅｓ”すなわちＤｍ（ｔｉ）＜５１２である場合には、上記ステップＳ２において算出された商ｊに１を加算し、この値をλｍアドレスＡｊとし（ステップＳ６３）、また、判定結果が”Ｎｏ”すなわちＤｍ（ｔｉ）≧５１２である場合は、上記ステップＳ２で検出されたＡｊをそのままλｍアドレスＡｊとする（ステップＳ６２）。
【０１０７】
次のステップＳ７では、上記の手順によってλｍアドレスＡｊを確定した後、ＣＰＵ７３はλｍアドレスＡｊに波長λｍを乗じたのち、上記Ｄｍ（ｔｉ）と加算し、時刻ｔｉにおける１６λｍ区間内における絶対位置Ｘ（ｔｉ）を生成する。
【０１０８】
以下、サンプリング時刻ごとに同様の手順（Ｓ１〜Ｓ７）を繰り返し、１６λｍ区間内における絶対位置を求める。また、生成された絶対位置Ｘ（ｔｉ）は、例えば表示として利用する、又は測定データとして外部のシステムに出力するなど、必要に応じて利用できる。
【０１０９】
ところで、本発明に係る位相検出型のアブソリュートエンコーダ１００では、λｍアドレス切替え部にあらかじめシステムに想定される内挿誤差を見越した判定ゾーンを設け、上記λｍアドレスＡｊの検出に伴う誤動作を防いでいる。
【０１１０】
しかしながら、出荷時においては高精度であっても、例えば、構成部品の劣化や調整状態の変化などにより内挿誤差が悪化し、あらかじめ想定した内挿誤差の見積量を超え、結果としてλｍアドレスＡｊの検出に誤差を生じる。
【０１１１】
さらには、これらの内挿誤差を多めに見積もると上記Ｎを大きくできず、結果としてアブソリュートで計測区間が狭まる等の問題がある。
【０１１２】
そこで、このアブソリュートエンコーダ１００では、内挿誤差の劣化等を監視し、必要に応じて自動調整機能を起動することによって内挿誤差を常に適正なレベル以下に保つことを保証し、上記Ｎとして大きな値を設定可能とし、もって高精度かつ広いアブソリュート計測区間の実現を可能としている。
【０１１３】
ここで、このアブソリュートエンコーダ１００における内挿誤差の監視及び自動調整ついて説明する。
【０１１４】
すなわち、このアブソリュートエンコーダ１００は、例えば、図２３に示すような構成のλａパルス生成部１６を備える。このλａパルス生成部１６は、上記第２の変位量検出部２３から位相変調信号ｅｐａが波形整形回路６１を介して供給されるＤ型フリップフロップ６２と、このＤ型フリップフロップ６２の出力が入力される同期微分回路６３とを備え、上記タイミング信号生成部１８から基準信号ＭＯＤＳがＤ型フリップフロップ６２に供給されるとともに、クロックパルスＭ×ｆが同期微分回路６３に供給されている。そして、このλａパルス生成部１６から出力されるλａパルスが、図１１に示した演算部１７のＣＰＵ７３に入力されており、必要に応じてＣＰＵ７３に対する割り込み信号として機能する。また、上記演算部１７のＲＡＭ７４の一部は、不揮発メモリで構成されている。
【０１１５】
次に、上記に示す内挿誤差の監視と自動調整の実現にかかる基本的な原理について説明する。
【０１１６】
一般に、第１のスケール１１及び第２のスケール２１の目盛り（波長）は光波干渉計を基準として形成（記録）するため極めて高精度である。また、上記１２式あるいは１７式からも明らかなように、Ｘの周期、すなわち、波長λａの移動ごとに発生するλａパルスのピッチは、たとえＤＣのずれや出力レベルのずれに拘わらず常に一定の値となる。
【０１１７】
２）図１に示す第１のλパルス生成部１６は、アブソリュート計測区間において、第２のスケール２１の波長λａの移動毎にλａパルスを生成し、アブソリュート計測区間における移動に伴って１７ヶのλａパルスを発生する。ここで、第２のスケール２１の波長λａの各アドレス毎に発生するλａパルスを特定するため、λａアドレス（Ａｋ）がｋからｋ＋１に変化するとき発生するλａパルスをλａ（ｋ）と表すことにすれば、λａパルス（λａ（ｋ））の発生位置に対応する第１のスケール１１の波長λｍ内の位置は（１６−ｋ）×λｍ／１７となる。
【０１１８】
次に、各スケール１１，２１の記録条件が一定であり、スケール素材が一定の品質を保っていれば、比較的狭い範囲、例えば、アブソリュート計測区間における各波長内の内挿誤差は類似性を有すると考えられる。
【０１１９】
したがって、上記アブソリュート計測区間に亘る移動に際して正確なピッチで発生する１７ヶのλａパルス（λａ（ｋ））は、第１のスケール１１の１波長内をλｍ／１７ピッチでサンプルしたことに相当する。
【０１２０】
ここで、次の表５はλａパルス（λａ（ｋ））によってサンプリングされる波長λｍ内の位置（１６−ｋ）×λｍ／１７と該位置に対応するパルス数の理論値Ｄｍｔ（ｋ）を示したものであり、実際には各々のスケールにおける内挿誤差によりずれを生ずる。
【０１２１】
【表５】

【０１２２】
したがって、実際に計測された値と、上記表５に示す理論値との差異を検出することにより、各スケール１１，２１における内挿誤差を測定したことになる。
【０１２３】
ところで、この実施の形態においては、第１のスケール１１の波長λｍ内を１７回サンプリングしており、上述のようにＤＣのずれや出力レベルのずれによって発生する内挿誤差の周期は、波長λに対して最大でも２周期成分しか含まない。したがって、サンプリング定理からも明らかなように、これらの要因に伴う内挿誤差を完全に評価できる。
【０１２４】
次に、このアブソリュートエンコーダ１００における内挿誤差の監視と自動調整にかかる具体的な方法について説明する。
【０１２５】
内挿誤差の自動調整を行う場合、適切なタイミングで内挿誤差の監視を行い、該監視結果を所定の判定値と比較して自動調整を起動し、当該製品として許容する内挿誤差を維持するように構成する必要がある。
【０１２６】
１）「内挿誤差の規格値」の設定
既に述べたとおり、波長λに対する内挿誤差の比率（Ｉｅ／λ）は通常１％程度である。したがって、上記の値より大きく、かつ製品として許容し得る内挿誤差の規格値Ｉｒとして、例えば２％に設定すれば内挿誤差の規格値Ｉｒは０．００２×λとなり、この実施の形態では例えば２０μｍと設定することができる。
【０１２７】
この内挿誤差の規格値Ｉｒを演算部１７のＲＡＭ７４内の所定の不揮発領域に格納し、内挿誤差の監視時等に参照できるようにすればよいが、演算処理の面からは分解能単位のパルス数Ｄｒとして格納するのが好都合である。
【０１２８】
２）出荷時における内挿誤差の取得
上述のように、製品に許容する「内挿誤差の規格値Ｉｒ」を自動調整起動に際しての判定値とすることができる。
【０１２９】
しかしながら、同一の製品であっても個々の製品毎に内挿誤差の様子が異なること、さらには、経時的な変化を監視し、該監視結果に基づいて自動調整機能を起動することによって常に最良の精度を維持するという観点からは、当該製品の最良状態と想定される出荷時における内挿誤差を取得し、該取得された内挿誤差に所定のマージンを上乗せした値を判定値と設定する方が好都合な場合もある。
【０１３０】
以下、生産工程あるいは出荷時において実施される内挿誤差の取得について説明する。
【０１３１】
手順１１ 第１のスケール１１における最適な電気調整が終了した時点で、演算部１７の制御入力に所定の信号を与え「内挿誤差保存モード」を起動する。この内挿誤差保存モードにおいては、例えば、λａパルスλａ（ｋ）によってＣＰＵ７３に割り込みをかけ、パルスカウンタ７２の計数値、すなわち、上記λａパルスλａ（ｋ）に対応する波長λｍ内の絶対値を読込むように構成するのが好都合である。
【０１３２】
手順１２ 上記λａパルスλａ（ｋ）に対応する第１のスケール１１の波長λｍ内の理論的な絶対位置、すなわち、（１６−ｋ）×λｍ／１７に対応するパルスカウンタ７１の計数値Ｄｍ（ｋ）を取り込む。
【０１３３】
手順１３ 次に、パルスカウンタ７２の計数値ΔＤ（ｋ）を取り込み、上述のλｍアドレス特定手順に準じて、上記λａ（ｋ）が発生したλａアドレスｋを特定し、該特定されたｋを用いて上記λａ（ｋ）に対応する波長λｍ内の位置、すなわち、（１６−ｋ）×λｍ／１７位置における理論値Ｄｍｔ（ｋ）との差として該位置における内挿誤差Ｉｅｓ（ｋ）を計測し、アドレスｋをインデックスとして、不揮発メモリの所定の領域（以下、出荷時内挿誤差テーブルＴｉｓ（ｋ））に格納する。
【０１３４】
手順１４ 上記の手順１１〜手順１３を、全てのλａ（ｋ）、すなわち、ｋ＝０〜１６まで繰り返し、波長λｍ内の全域に亘って、λｍ／１７単位の内挿誤差を取得し、出荷時内挿誤差テーブルＴｉｓ（ｋ）に格納する。
【０１３５】
もし、上記によって計測された内挿誤差が製品に許容すべき内挿誤差Ｉｒより大きいときは、再度電気調整をやり直した上で上記手順１１〜手順１４を繰り返す。
【０１３６】
ここで、λａアドレスの特定は、λｍとλａとの関係を入れ換え、かつλａアドレスの１区間に対応するパルス数をＭ／１７（約６０）とし、さらに、上記Ｄｍ（ｋ）とΔＤ（ｋ）とを加算することによって、波長λａ内の絶対値Ｄａ（ｋ）を求めることにより、上述のλｍアドレスと同様の手順でλａアドレスを特定することができるが、該λａ（ｋ）パルスが入力されたときの波長λｍのＮ倍区間における絶対位置Ｘ（ｋ）を求め、該Ｘ（ｋ）をλａで除算する等の方法で実現できる。
【０１３７】
３）内挿誤差の監視
内挿誤差の監視は、通常ユーザ側において実施され、例えば、ユーザがキーボード等を介し、又は、電源投入時に自動的に「内挿誤差監視モード」を起動し、あらかじめ、当該製品の内挿誤差の規格値Ｉｒもしくは出荷時に取得され出荷時内挿誤差テーブルＴｉｓ（ｋ）に格納された値Ｉｅｓ（ｋ）に所定のマージンを持たせた値Ｉｅｃ（ｋ）を、ＲＡＭ７４内の所定の領域に設けた内挿誤差判定値テーブルＴｉｃ（ｋ）に格納し、波長λｍ内の所定の位置（λｍ／１７ピッチ）毎の判定値とするように構成するのが好都合である。また、このモードにおいては、規準信号ＭＯＤＳでの割り込みによる通常の絶対位置計測と、λａパルスでの割り込みによる内挿誤差監視とが同時並行的に実行するように構成するのが好都合であり、本発明の主旨を逸脱しない範囲で自由に選択することができる。
【０１３８】
ここでは、説明を簡単にするため内挿誤差の監視機能についてのみ説明する。
【０１３９】
手順２１ 演算部１７にλａ（ｋ）が入力されると、ＣＰＵ７３は割り込み動作を開始し、パルスカウンタ７１の計数値Ｄｍ（ｋ）を取り込み、ＲＡＭ７４の一時記憶領域に保存する。
【０１４０】
手順２２ 同時に、パルスカウンタ７２の計数値ΔＤ（ｋ）を取り込み、λａアドレスｋを特定し、該特定されたｋを用いて上記λａ（ｋ）に対応する波長λｍ内の位置、すなわち、（１６−ｋ）×λｍ／１７位置における理論値Ｄｔ（ｋ）との差として該位置における内挿誤差Ｉｅ（ｋ）を検出し、アドレスｋをインデックスとして所定の領域（以下、監視時内挿誤差テーブルＴｉ（ｋ））に格納する。
【０１４１】
手順２３ 次に、該計測された内挿誤差Ｉｅ（ｋ）を、ＲＡＭ７４の内挿誤差判定値テーブルＴｉｃ（ｋ）内の所定の領域に格納された内挿誤差の判定値Ｉｅｃ（ｋ）と比較し、該規格値を超えた時は、上記監視時内挿誤差テーブルＴｉ（ｋ）の対応する位置に内挿誤差エラーＥｒ（ｋ）を書き込む。
【０１４２】
手順２４ 上記の手順２１〜手順２３を、全（すべ）てのλａ（ｋ）に対して実行し、波長λｉｎ内におけるλｍ／１７単位のすべての位置における内挿誤差Ｉｅ（ｋ）及び内挿誤差エラーＥｒ（ｋ）を計測し、それらの結果を監視時内挿誤差テーブルＴｉ（ｋ）に格納する。
【０１４３】
４）内挿誤差エラーの判定
上記３）の内挿誤差の監視において計測されたＩｅ（ｋ）を内挿誤差判定値Ｉｅｃ（ｋ）と比較し、最初の内挿誤差エラーＥｒ（ｋ）が検出された時点で直ちに内挿誤差が判定値を超過したものと判定し、直ちに警報信号を出力することもできる。しかしながら、ただ１回の内挿誤エラーＥｒ（ｋ）のみで判定すると、例えばノイズ等による一時的な反応と混同し、過剰な警報信号となり得る。また、後述のように、通常、内挿誤差は波長λ内において１周期若しくは２周期で振動する信号である。
【０１４４】
したがって、次のように判定することにより、誤動作のない確実な内挿誤差エラーの判定が可能となる。
【０１４５】
波長λｍの１周期内で発生した内挿誤差エラーＥｒ（ｋ）の個数を計数し、その個数に応じて次の処理を行う。
【０１４６】
処理１ １箇所のみで発生した時は、次のサイクルにおいても同じ位置、すなわち、同一のλａ（ｋ）で内挿誤差エラーＥｒ（ｋ）された場合のみ、内挿誤差エラーが発生したものと判定する。
【０１４７】
処理２ 波長λｍ内の２箇所以上で内挿誤差エラーＥｒ（ｋ）発生した時は、無条件に内挿誤差エラーが発生したものと判定する。
５）内挿誤差要因の特定
内挿誤差エラーと判定されると、ＣＰＵ７３は、該誤差要因を補正する電子ボリュームＤＣ１，ＤＣ２，ＧＡＪに適切な調整信号ＡＤＪを与えるための前準備としての誤差要因の特定処理を開始する。
【０１４８】
誤差要因の特定方法としては、波長λａの１周期にわたって検出され、監視時内挿誤差テーブルＴｉ（ｋ）に格納された内挿誤差Ｉｅ（ｋ）をフーリエ解析によって分析し、波長λｍに対する１周期若しくは２周期成分のレベルと位相とに分解することにより誤差要因を特定することもできる。
【０１４９】
しかしながら、誤差の要因が極めて限られていること、また、表２にも示すごとく、波長λ内における内挿誤差の周期と位相は各々の誤差要因に対して独立なパターンを有することを利用し、波長λ内において検出された内挿誤差の周期と、該内挿誤差の極大（山）となる位置と極小（谷）となる位置のとの関係を利用し、所定の判定ルールに照らして誤差要因を特定するのがコスト的な面や処理速度の面からも好都合である。
【０１５０】
図２４は、誤差要因の特定に関する手順を示したものであり、以下詳細に説明する。
【０１５１】
先ず、Ｅｒ（ｋ）の発生した位置に対し、ｋの降順、すなわち、波長λｍ内の０側から内挿誤差Ｉｅ（ｋ）を検査し、符号を考慮して最大の内挿誤差Ｉｅ（ｋ）ｍａｘが検出されたλａ（ｋ）ｍａｘ及び最小の内挿誤差Ｉｅ（ｋ）ｍｉｎが検出されたλａ（ｋ）ｍｉｎを探す（ステップＳ１１）。
【０１５２】
次に、最大の内挿誤差Ｉｅ（ｋ）ｍａｘが検出されたλａ（ｋ）ｍａｘ及び最小の内挿誤差Ｉｅ（ｋ）ｍｉｎが検出されたλａ（ｋ）ｍｉｎから波長λｋ内の位置を求め、第１の極大値（山）及び第１の極小値（谷）としてＲＡＭ７４の所定の記憶領域に記憶する（ステップＳ１２）。ここで、λａ（ｋ）に対応する第１のスケール１１の波長λｍ内の位置は（１６−ｋ）×λｍ／１７として求まることは既に述べたとおりである。
【０１５３】
次に、上記ステップＳ１２で検出された第１の極大値（山）と第１の極小値（谷）の数を判定する（ステップＳ１３）。
【０１５４】
そして、第１の極大（山）と第１の極小値（谷）が各々１箇ずつ検出された時はステップＳ１４に移り、また、第１の極大（山）すなわちＩｅ（ｋ）ｍａｘ若しくは第１の極小（谷）すなわちＩｅ（ｋ）ｍｉｎのいずれかしか検出されなかった時は、３要因によるものと判定してステップＳ２２に移る。
【０１５５】
次のステップＳ１４では、上記Ｉｅ（ｋ）ｍａｘ及びＩｅ（ｋ）ｍｉｎを除く位置で、２番目に大きな内挿誤差Ｉｅ（ｋ）ｍａｘ２が検出された第２の極大値（山）と２番目に小さい内挿誤差Ｉｅ（ｋ）ｍｉｎ２が検出された第２の極小値（谷）を探す。そして、第２の極大値（山）及び第２の極小値（谷）の位置を求め（ステップＳ１５）、その有無と数を判定し（ステップＳ１６）、次の処理を行う。
【０１５６】
すなわち、上記Ｉｅ（ｋ）ｍａｘ２とＩｅ（ｋ）ｍｉｎ２のいずれも検出されなかった時は、１周期成分、すなわち、ＤＣのずれのみと判定し、上記ＲＡＭ７４に格納されている第１の極大値（山）と第１の極小値（谷）の値を表２及び表３のルールを用いて分析し、ＤＣの重畳しているヘッドの数（単独若しくは両方）と帯磁の方向（Ａ及びＢ符号）を判定する（ステップＳ１７）。
【０１５７】
また、Ｉｅ（ｋ）ｍａｘ２とＩｅ（ｋ）ｍｉｎ２が検出された時、すなわち、２周期成分が検出された時は、該検出された第２の極大値（山）及び第２の極小値（谷）をＲＡＭ７４の所定の記憶領域に格納し、上記ＲＡＭ７４に格納されているこれら２組の極大値と極小値の大小関係を判定し（ステップＳ１８）、その判定結果に基づいて、単独要因によるものか、２つの要因によるものかを判定する（ステップＳ１９）。
【０１５８】
ステップＳ２０では、Ｉｅ（ｋ）ｍａｘとＩｅ（ｋ）ｍａｘ２及びＩｅ（ｋ）ｍｉｎとＩｅ（ｋ）ｍｉｎ２のレベルに明らかな差が認められるときに、片側のヘッドのＤＣずれと出力レベルのずれが重畳しているものと判断し、上記ＲＡＭ７４に格納されている第１の極大値（山）と極小値（谷）の位置を表３に示すルールを用いて分析し、帯磁しているヘッド（Ａ又はＢ）とずれの方向（正負）及び出力レベルのずれの方向（ｂの符号）を特定する。
【０１５９】
ステップＳ２１では、Ｉｅ（ｋ）ｍａｘとＩｅ（ｋ）ｍａｘ２及びＩｅ（ｋ）ｍｉｎとＩｅ（ｋ）ｍｉｎ２のレベル、すなわち内挿誤が略等しいときに、ＣＨ１出力レベルとＣＨ２出力レベルのずれによる要因と判定し、上記２組の極大値（山）及び極小値（谷）を表２のルールを用いて分析し、出力レベルの大小関係、すなわち、誤差要因ｂ符号を特定する。
【０１６０】
ステップＳ２２では、Ｉｅ（ｋ）ｍａｘ（極大（山））若しくはＩｅ（ｋ）ｍｉｎ（極小（谷））のいずれかしか検出できなかった時に、両チャンネルのヘッドのＤＣずれと出力レベルのずれとその方向（Ａ，Ｂ，ｂの極性）を特定する。
【０１６１】
そして、以上の手順により特定された誤差要因（Ａ，Ｂ，ｂ）と、該ずれの方向（正負）とをＲＡＭ７４の所定の記憶領域に格納した後（ステップＳ２３）、警報信号を出力する（ステップＳ２４）。
【０１６２】
ところで、今までの説明においては、理解を容易にするため、２チャンネルのヘッドのＤＣずれに関してはずれの絶対値を等しいものとして説明した。
【０１６３】
しかしながら、実際にはずれの絶対値が等しくならないことの方が多く、その場合においては、上述の（１３）式からも明らかなように、極大値（山）と極小値（谷）の位置がλ／８（π／４）単位の位置とずれることがあり、その前後、例えば、本事例におけるごとく波長λ／１６単位でサンプルしている場合についてはその前後において極大（山）と極小（谷）が検出されることを考慮する必要がある。
【０１６４】
また、この実施の形態では、表３を波長λ間を１６分割したテーブルで示したが、１７分割し、端数を丸めた値として計算し、該丸めた位置における極大（山）又は極小（谷）を理論的に算出し、その値を比較のためのテーブルとすれば良い。
【０１６５】
５）自動調整
警報信号が出力されると、システムは「自動調整モード」を起動し、上記の特定された誤差要因の情報をもとに自動調整を開始することができる。
【０１６６】
「自動調整モード」への移行方法としては、例えば、警報信号の出力と連動して自動的に移行する方法や、ユーザや他の機器等が警報信号の出力されたことを認識し、キーボードや外部からの指令信号等を介して起動する等の方法があり、必要に応じて選択すればよい。
【０１６７】
基本的には、上述のごとく誤差要因とそのずれの方向が明らかとなっているため、その要因を特定し、ＣＰＵ７３から調整信号ＡＤＪを与え、該要因を打ち消す方向に電子ボリュームＤＣ１、ＤＣ２、ＧＡＪを制御すれば良い。
【０１６８】
以下、「自動調整モード」が起動された時の自動調整の処理について説明する。
【０１６９】
なお、実際の調整動作においては、調整時における内挿誤差を格納する一次記憶用のテーブル領域を確保しておく必要があるが、説明を簡単にするため省略する。
【０１７０】
自動調整の処理は、原理的に次の手順３１〜手順３５にしたがって実行される。
【０１７１】
手順３１ 「自動調整モード」が起動されると、ＣＰＵ７３は上記誤差要因を格納しているＲＡＭ７４を参照し、該誤差要因から調整対象とすべき電子ボリュームＤＣ１，ＤＣ２，ＧＡＪを特定し、調整信号ＡＤＪの選択信号ＳＬＴにより選択指定される該電子ボリュームに選択信号ＣＳ１，ＣＳ、ＣＳ３を供給する。
【０１７２】
手順３２ 次に、上記ＲＡＭ７４からずれの方向を参照し、該ずれの方向を補整する向きに電子ボリュームＤＣ１，ＤＣ２，ＧＡＪのワイパー位置を可変するワイパー制御信号（Ｕ／Ｄ，ＩＮＣ）を与える。
【０１７３】
手順３３ しかる後、該誤差要因において特定される極大（山）位置における内挿誤差Ｉｅ（ｋ）ｍａｘと極小（谷）位置における内挿誤差Ｉｅ（ｋ）ｍｉｎを計測する。そして、該位置における内挿誤差が低減する方向に変化していれば、該検出された内挿誤差を出荷時の内挿誤差テーブルＴｉｓ（ｋ）に格納されている値と比較し、その差異が最小になるまで上記手順３２を繰り返す。また、検出された内挿誤差が増加する方向へ変化している時は、上記電子ボリュームのワイパーを反対方向へ移動する向きにワイパー制御信号を与え、該検出された内挿誤差を出荷時における内挿誤差テーブルＴｉｓ（ｋ）と比較し、その差異が最小になるまで上記手順３２を繰り返す。
【０１７４】
手順３４ 上記の手順３１〜手順３３により、最良の調整状態における内挿誤差が出荷時の内挿誤差Ｉｅｓ（ｋ）を再現できたら終了信号を出力し、「自動調整モード」を終了する。
【０１７５】
以上、自動調整の基本的な処理について説明した。しかしながら、実際には、複数の誤差要因が重畳する場合もあり、調整の順序等も考慮する必要がある。この場合は、誤差パターンの特性を顧慮して、次のような順序で調整するのが好都合である。
【０１７６】
（ａ）ＤＣのずれと出力レベルのずれか重畳している場合は、最初にＤＣのずれを調整し、第１の極大値（山）における内挿誤差Ｉｅ（ｋ）ｍａｘと第２の極大値（山）における内挿誤差Ｉｅ（ｋ）ｍａｘ２（又は、第１の極小値（谷）における内挿誤差Ｉｅ（ｋ）ｍｉｎと第２の極小値における内挿誤差Ｉｅ（ｋ）ｍｉｎ２における差異を最小にするように調整した後、出力レベルの調整に移るのが好都合である。
【０１７７】
（ｂ）２チャンネルのＤＣずれと出力レベルのずれが重畳している場合においては、先ず、出力レベルの調整を行い、ＤＧのずれによる内挿誤差パターンを再現した後、波長λに対する１周期の成分とし、しかる後、極大点の位相からＤＣずれの大きい側のヘッドを特定して調整し、最後にＤＣずれの小さいほるのヘッドのＤＣずれを調整するのが好都合である。
【０１７８】
また、これら複数の要因が重畳している場合には、最良状態に調整でき、かつ、調整された結果が規格値を満足できた場合においても必ずしも出荷時における内挿誤差Ｉｅｓ（ｋ）のパターン、例えば、誤差の周期や（山）と極小（谷）の位置等が一致しない場合もあり得る。
【０１７９】
この場合は、最良状態に調整した時点での内挿誤差を次回に自動調整が起動された時の判定基準とする、例えば、出荷時内挿誤差テーブルＴｉｓ（ｋ）の内容を、上記最良調整状態での内挿誤差テーブルＴｉ（ｋ）の値で書き換え、次回はこの値を基準として調整するなど、本発明の主旨を逸脱しない範囲で自由に対応できる。
【０１８０】
本発明の具体的な実施の形態として、所望の分解能Ｒを１μｍ得るために、第１のスケール１１の波長λｍを所望の分解能１μｍの１，０２４倍とし、位相変調信号のキャリア周波数ｆの１，０２４（Ｍ＝１，０２４）倍のクロックパルスを用いて内挿するとともに、第１スケール１１の波長λｍの１６倍区間（Ｎ＝１６）の絶対位置を検出するため、第１のスケール１１の波長λｍと第２のスケール２１の波長λａとの関係を１６×λｍ＝１７×入ａなる２進系列の数値を設定した事例について説明した。
【０１８１】
しかしながら、上述のごとく、所望の分解能λｍ／Ｍを得るため第１のスケールの波長λｍ及び設計条件Ｍ、Ｎは１０進系列の値、例えば、λｍ＝ｌ，０００、Ｍ＝１０００、Ｎ＝２０のよう設定し得ることは言うまでもない。
【０１８２】
また、本実施例において、第１のスケールと第２のスケールの位相差Δφを第２の位相比較部２４を用いて直接算出している。
【０１８３】
しかしながら、第１の位相比較部１４におけるごとく、第２の位相比較部２４においても、位相変調信号ｅｐａと上記基準信号との位相比較により第２のスケール２１の波長λａ内の絶対位置として検出し、該位相差をφａ（ｔｉ）として検出した後、第２のゲート回路２５及びパルスカウンタ７２を介してパルス数Ｄａ（ｔｉ）として検出し、演算部１７により演算処理ΔＤ＝Ｄａ（ｔｉ）−Ｄｍ（ｔｉ）として検出できることは言うまでもない。
【０１８４】
また、本発明においては、測尺の基準となる第１のスケール１１、すなわち、第１の変位量検出部１３のみを調整する例を説明したが、図１において破線で示したごとく、第１の変位量検出部１３からの出力と基準信号を供給して第２のλパルス生成部２６から出力されるλｍパルスを用い、第２の変位量検出部２３の自動調整を行うように構成することもできる。
【０１８５】
さらに、内挿誤差の監視においても、システムに電源が投入された時点、あるいはタイマー等により定期的なタイミングで監視を起動するなど、本発明の主旨を逸脱しない範囲で様々な方法が適用できる。
【０１８６】
また、自動調整の開始においても、警報信号の出力とともに自動的に調整モードを起動するだけでなく、例えば、ユーザは外部からの制御信号を入力した時点で自動調整を起動するなど、様々な態様に適用可能である。
【０１８７】
さらに、上述の実施の形態においては磁気式について説明したが、本発明は、磁気式に制約されることなく、例えば光学方式のごとくスケールとヘッドとの相対移動に対応した２チャンネルの直交する正弦波信号が出力されるようなシステムにおいても、該検出された２相の正弦波信号を、キャリア周波数ｆの信号で直交変調した後加算することにより、位相変調信号を取り出すようにすれば同様に適用できる。
【０１８８】
さらに、本発明においてはリニアエンコーダヘの適用について説明したが、繰り返し区間を円周長と一致させることにより回転量のアブソリュート検出にも適用可能である。
【０１８９】
【発明の効果】
以上のように、本発明によれば、トラック数が少ないため、システムの構造が簡単になり、小型・低コストで絶対位置検出システムを実現できる。
【０１９０】
また、位相変調信号を用いた位置検出システムのため、該位相変調信号のキャリア周波数ｆに対し、クロックパルスの周波数Ｍ×ｆを高く、すなわち、Ｍを大きくするだけで容易に高分 解能を実現することができる。また、交流結合方式のため、例えば、差動トランス方式におけるようなＤＣドリフトの影響を受けないため、ウオームアップなしで直ちに測定できる。さらに、差動トランス方式においては出力レベルの変動が直ちに誤差の要因となるが、本方式においては、たとえ２チャンネルの検出ヘッドからの出力レベルが変化しても、出力の比が変動しない限り誤差の要因とならず、安定かつ高精度なリニアエンコーダを実現できる。
【０１９１】
また、他の計測システムの助けを借りることなく、本システム単独で内挿誤差を監視することができるだけでなく、計測された内挿誤差を分析し、該内挿誤差の発生要因を特定することができる。
【０１９２】
したがって、内挿誤差が所定のレベルを超えた時点で警報信号を出力することができるだけでなく、警報信号を検出して時点で自立的若しくは外部の制御信号の指令により自動調整を実行することができるため、取付け状態の変化や部品の劣化等に起因する内挿誤差の悪化を事前に検出し補正することができ、長期間にわたって高精度な検出が実現できる。
【０１９３】
さらに、内挿誤差を所定のレベルに維持することが可能なため、内挿誤差に起因するλアドレスの誤動作を防ぐために設定したλアドレスの切替え領域における判定ゾーンを最小に設定でき、結果としてＮを大きく選ぶことができ、アブソリュート計測範囲を拡大することができる。
【図面の簡単な説明】
【図１】本発明に係るアブソリュートエンコーダの構成を示すブロック図である。
【図２】上記アブソリュートエンコーダにおける第１の変位量検出部の構成を示すブロック図である。
【図３】上記アブソリュートエンコーダにおける検出ヘッドを構成するＭＲセンサの構成例を示す図である。
【図４】上記第１の変位量検出部に備えられる加算増幅器の構成例を示す図である。
【図５】上記第１の変位量検出部に備えられる電子ボリュームの構成例を示す図である。
【図６】上記第１の変位量検出部に備えられる調整信号制御回路の構成例を示す図である。
【図７】上記アブソリュートエンコーダにおける第１の位相比較部の構成を示すブロック図である。
【図８】上記第１の位相比較部の動作タイミングを示す図である。
【図９】アブソリュート計測区間において第１の変位量検出回路で検出された位相量φｍと、第２の変位量検出回路で検出された位相量φａとの関係を示す図である。
【図１０】第１の変位量検出部で検出された位相量φｍと、φａとφｍとの位相差Δφ（＝φａ−φｍ）との関係を示す図である。
【図１１】演算部の構成例を示すブロック図である。
【図１２】ＣＨ１ヘッド及びＣＨ２ヘッドの正負のＤＣずれが重畳した内挿誤差の一例を示す図である。
【図１３】ＣＨ１ヘッド及びＣＨ２ヘッドの正負のＤＣずれが重畳した内挿誤差の一例を示す図である。
【図１４】ＣＨ１ヘッド及びＣＨ２ヘッドの正負のＤＣずれが重畳した内挿誤差の一例を示す図である。
【図１５】ＣＨ１ヘッド及びＣＨ２ヘッドの正負のＤＣずれが重畳した内挿誤差の一例を示す図である。
【図１６】ＣＨ１ヘッド及びＣＨ２ヘッドの正負のＤＣずれが重畳した内挿誤差の一例を示す図である。
【図１７】ＣＨ１ヘッド及びＣＨ２ヘッドの正負のＤＣずれが重畳した内挿誤差の一例を示す図である。
【図１８】ＣＨ１ヘッド及びＣＨ２ヘッドの正負のＤＣずれが重畳した内挿誤差の一例を示す図である。
【図１９】ＣＨ１ヘッド及びＣＨ２ヘッドの正負のＤＣずれが重畳した内挿誤差の一例を示す図である。
【図２０】ＤＣ偏倚が乗じたときの第１のスケールの内挿誤差ΔＸの例を示す図である。
【図２１】λｍアドレスＡｊの確定にかかる基本的な考え方を説明する図である。
【図２２】λｍアドレスＡｊの判定手順を示すフローチャートである。
【図２３】上記アブソリュートエンコーダに備えれるλパルス生成部の構成例を示すブロック図である。
【図２４】誤差要因の特定に関する手順を示すフローチャートである。
【符号の説明】
１１ 第１のスケール、１２Ａ，１２Ｂ ＭＲセンサ、１２ 第１の検出ヘッド、１２Ａ_１，１２Ａ_２，１２Ａ_３，１２Ａ_４，１２Ｂ_１，１２Ｂ_２，１２Ｂ_３，１２Ｂ_４ ＭＲ素子、１３ 第１の変位量検出部、１４ 第１の位相比較部、１５ 第１のゲート回路、１７ 演算部、１８ タイミング信号生成部、２１ 第２のスケール、２２ 第２の検出ヘッド、２３ 第２の変位量検出部、２４ 第２の位相比較部、２５ 第２のゲート回路、３１，３２ 乗算器、３３ 加算増幅器、３４ 調整信号制御回路、３４Ａ デコーダ、４１ 波形整形回路、４２，４３ 第１及び第２の同期微分回路、４４ ＪＫフリップフロップ、６１ 波形整形回路、６２ Ｄ型フリップフロップ、６３ 同期微分回路、７１，７２パルスカウンタ、７３ ＣＰＵ、７４ ＲＡＭ、９１ アップダウンカウンタ、９２ ポジションデコーダ、９３ 可変抵抗器、ＤＣ１，ＤＣ２，ＧＡＪ 電子ボリューム、１００ アブソリュートエンコーダ[0001]
BACKGROUND OF THE INVENTION
The present invention is used for dimensional measurement in the industrial machinery field, for example, measurement of adhesion and parallelism of mold press dies, measurement and control of rolling amount in a roll rolling mill, and measurement of part dimensions in production / assembly factories, etc. It is related to the absolute encoder that has been used favorably.
[0002]
[Prior art]
A first modulation circuit that outputs a phase modulation signal whose phase repeatedly changes in response to a relative displacement between a first scale having a predetermined wavelength λm and a first detection head that detects a signal of the first scale. The phase repeatedly changes in response to the relative displacement between the second detecting head for detecting the second scale and the second scale having a wavelength λa different from the first scale and the second scale head for detecting the signal of the second scale. A second detection unit that outputs a phase modulation signal; and a phase comparison circuit that compares a phase difference between signals obtained from the first detection unit and the second detection unit. A signal whose phase changes corresponding to a signal having a wavelength sufficiently longer than the wavelengths of the first scale and the second scale is obtained, the first scale and the second scale, the first detection head, Relative position with the second detection head Japanese Patent Publication No. 50-23618 discloses a measuring device for detecting the above.
[0003]
In this measuring device, the phase modulation signal epm obtained from the first detection means and the phase modulation signal epa obtained from the second detection means can be expressed by the following equations.
[0004]
epm = Ep1 × Sin (2πft + 2πX / λm)
epa = Ep2 × Sin (2πft + 2πX / λa)
Further, when the relationship between the wavelengths λm and λa is selected as, for example, a relationship of N × λm = (N−1) × λa, the phases θm (= 2πX / λm) and θa (= 2πX / λa) of these signals The difference Δθ (= θm−θa) is a signal that repeats at a period N times the wavelength λm as shown in the following equation.
[0005]
Δθ = 2πX / (N × λm)
Therefore, the measuring device can detect the position in units of λm in the N × λm section by dividing the phase difference Δθ by a predetermined phase difference, that is, 2π / N.
[0006]
[Problems to be solved by the invention]
However, in the above measuring device, when an ideal signal cannot be obtained from each scale, for example, when the phase modulation signal is incomplete, a periodic error occurs in the wavelength λ, and these errors are It is also superimposed on the switching position in λm units. Therefore, when the phase difference Δθ is divided by 2π / N as described above, a deviation occurs due to the error, and as a result, an error of λm may occur.
[0007]
In order to avoid these errors, a method of directly detecting the inside of Nλm using the phase difference Δθ obtained from the above two scales can be considered, but in this method, a virtually long scale in which the phase difference Δθ is Nλm is considered. When the phase is 2π, that is, the signal periodically changes in phase, the clock frequency M × f M times the carrier frequency f is interpolated into the carrier frequency f to obtain high resolution. As the resolution λm / (M × N) increases the wavelength λm or N, it becomes more difficult to obtain a high resolution. In addition, there is a problem that detection accuracy deteriorates due to a change with time.
[0008]
Therefore, in view of the conventional problems as described above, the object of the present invention is to monitor the detection error associated with the time-dependent change of the mounting state of the absolute encoder, the component parts, etc., and perform automatic adjustment as necessary. It is always to realize highly accurate detection.
[0009]
[Means for Solving the Problems]
The present invention relates to a first scale having a scale of wavelength λm, a first detection head for detecting a signal of the first scale, and the relative relationship between the first scale and the first detection head. A first displacement amount detecting means configured to extract the displacement amount as a phase modulation signal having a carrier frequency f; a second scale having a scale having a wavelength λa different from the first scale; A second detection head for detecting a signal of a second scale, and a second displacement of the second scale and the second detection head is extracted as a phase modulation signal of the carrier frequency f. Detecting the phase difference between the displacement amount detection means, the phase modulation signal obtained from the first displacement amount detection means, and the phase modulation signal obtained from the second displacement amount detection means, and using the phase difference, First scale wavelength λm A carrier frequency obtained from the first displacement amount detection means in an absolute encoder configured to obtain a periodic signal repeated at an N-fold wavelength and to absolutely detect an N-fold section of the wavelength of the first scale. first phase comparison means for generating a pulse width modulation signal corresponding to the absolute position of the first detection head within the wavelength λm of the first scale by comparing the phase modulation signal of f with the reference signal of frequency f The pulse width modulation signal generated by the first phase comparison means is interpolated with a clock pulse M times the carrier frequency f to correspond to the absolute position of the first detection head within the wavelength λm of the first scale. A first gate circuit for converting to a pulse train of resolution λm / M, a phase modulation signal output from the first displacement amount detection means, and a second displacement amount detection means A second phase comparison means for generating a pulse width modulation signal corresponding to the phase difference of the output phase modulation signal, and the pulse width modulation signal generated by the second phase comparison means is M times the carrier frequency f. The second gate circuit that interpolates the clock pulse and converts it into a pulse train corresponding to the phase difference, the second displacement amount detecting means, and the reference signal are compared to correspond to the wavelength λa of the second scale. The pulse generation means for generating a pulse for each displacement amount and the output of the second gate circuit are used to specify the address of the wavelength λm of the first scale, and are output from the first gate circuit. The absolute position within the wavelength λm of the first scale is obtained by counting the number of pulse trains, and the above-mentioned identified λm address is combined to decompose over the N-fold section of the wavelength λm of the first scale. The absolute position within the wavelength λm of the first scale corresponding to the generation position of the λa pulse output from the second pulse generation means is measured and detected from the theoretical value of the measurement value. The deviation amount is detected as an interpolation error, the detected interpolation error and a preset standard value are stored in a storage means, and the detected interpolation error is analyzed to detect the first displacement amount. Arithmetic means for supplying an adjustment signal to an electronic adjustment means provided in the means to perform an electrical adjustment so as to minimize an interpolation error in the first scale.
[0010]
DETAILED DESCRIPTION OF THE INVENTION
Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.
[0011]
An embodiment of the present invention described below uses a phase modulation signal output from two tracks having different wavelengths and uses a phase modulation signal to detect an absolute range longer than the recording wavelength of the track, for example, a special measurement system. This is an improvement of the measuring device disclosed in Japanese Patent Publication No. 50-23618. A magnetic system to which the present invention can be most effectively applied will be described in detail with reference to the drawings. Also, in the magnetic type, there are a method using a supersaturated core in the detection head and a method using a magnetoresistive effect element (hereinafter referred to as an MR sensor). The case where an MR sensor is used as the detection head will be described with reference to the reproduction wavelength.
[0012]
In the present invention, the relationship between the wavelength λm of the first scale and the wavelength λa of the second scale is selected as N × λm = (N ± 1) × λa, and the carrier frequency f output from the first scale is By interpolating the phase-modulated signal using a clock pulse M times the carrier frequency f, measurement can be performed with a resolution λm / M over a section N times the wavelength λm of the first scale. N and M can be arbitrarily set without departing from the gist of the present invention.
[0013]
Here, in order to simplify the explanation, specific numerical values are applied to the above-mentioned λm, N, and M, the wavelength λm is 1,024 μm, and the interpolation number M is 1,024 (= 2). ¹⁰ ) And N is 16 and the relationship between the wavelengths λm and λa is 16 × λm = (16 + 1) × λa = 17 × λa, so that one wavelength of λm is divided by 1,024 to obtain a resolution of 1 μm. This is explained based on a specific configuration example that can be detected in absolute.
[0014]
FIG. 1 is a block diagram showing a configuration of an absolute encoder 100 according to the present invention.
[0015]
The absolute encoder 100 has a first scale 11 formed by recording a scale having a wavelength λm (= 1,024 μm) on a scale member (not shown), and is opposed to the first scale 11 and has a wavelength λm. On the other hand, a first detection head 12 composed of two-channel MR sensors 12A and 12B arranged with a 90 ° phase difference (λm / 4) is provided. The first detection head 12 has two systems of sine wave signals having a relative movement period of 2πx / λm (where x is a displacement) and having a 90 ° phase difference due to relative movement with the first scale 11. Sin (2πx / λm) and Cos (2πx / λm) are output.
[0016]
These two-phase sine wave signals Sin (2πx / λm) and Cos (2πx / λm) are guided to, for example, the first displacement amount detection unit 13 configured as shown in FIG.
[0017]
The first displacement detector 13 is a two-phase sine wave signal Sin (2πx / λm), Cos (2πx) obtained by the two-channel MR sensors 12A and 12B constituting the first detection head 12. / Λm) is provided by the timing signal generator 18 to perform balanced modulation with the two-phase signals MODC and MODS of the carrier frequency f having a phase difference of 90 °, and balanced modulation by the multipliers 31 and 32 is performed. A summing amplifier 33 for adding the signals Sin (2πx / λm) × Cosωt and Cos (2πx / λm) Sinωt is provided.
[0018]
As shown in FIG. 3, the two-channel MR sensors 12A and 12B constituting the first detection head 12 are each an MR element 12A. ₁ , 12A ₂ , 12A ₃ , 12A ₄ (12B ₁ , 12B ₂ , 12B ₃ , 12B ₄ DC2) is bridge-connected, and an electronic volume DC1 (DC2) for correcting a DC bias shift (hereinafter referred to as DC shift) is provided at the other terminal of the MR sensor connected in a bridge.
[0019]
The addition amplifier 33 adds the balanced modulation signals Sin (2πx / λm) × Cosωt and Cos (2πx / λm) Sinωt obtained from each channel to generate a phase modulation signal Ep1 × Sin (ωt + 2πx / λm). As shown in FIG. 4, an electronic volume GAJ for equalizing the output levels of balanced modulation signals Sin (2πx / λm) × Cosωt, Cos (2πx / λm) Sinωt between two channels is provided. Yes.
[0020]
As shown in FIG. 5, each of the electronic volumes DC1, DC2, and GAJ has an up / down counter 91 to which a selection signal CS and wiper control signals U / D and INC are supplied. The position decoder 92 decodes the count output, and a variable resistor 93 whose resistance value is switched by the decode output of the position decoder 92.
[0021]
The first displacement detector 13 includes an adjustment signal control circuit 34 for selecting the three electronic volumes DC1, DC2, and GAJ and supplying an adjustment signal. The adjustment signal control circuit 34 includes a decoder 34A that decodes the selection signal SLT of the adjustment signal ADJ supplied from the arithmetic unit 17, as shown in a specific configuration example in FIG. Selection signals CS1, CS2, CS3 and wiper control signals U / D, INC are supplied to the electronic volumes DC1, DC2, GAJ, and the wiper positions of the electronic volumes DC1, DC2, GAJ are electronically variably controlled.
[0022]
The first displacement amount detection unit 13 is a two-phase sine wave signal Sin (2πx / λm), Cos (obtained by the two-channel MR sensors 12A and 12B constituting the first detection head 12. 2πx / λm) is balanced-modulated by the two-phase signals MODC and MODS having a carrier frequency f having a phase difference of 90 ° given by the timing signal generator 18 in the multipliers 31 and 32, and balanced by the multipliers 31 and 32. The modulation signal Sin (2πx / λm) × Cosωt, Cos (2πx / λm) Sinωt is added with the signal level matched by the addition amplifier 33, and a phase modulation signal epm as shown in the following equation (1) is output. .
[0023]
epm = Ep1 × Sin (ωt + 2πx / λm) (1)
Where ω = 2πf, x = relative displacement
For example, the phase modulation signal epm is guided to the first phase comparison unit 14 configured as shown in FIG.
[0024]
The first phase comparison unit 14 includes first and second synchronous differentiation circuits 42 and 43 and a JK flip-flop 44 to which a clock pulse having a frequency M × f given by the timing signal generation unit 18 is supplied. The differential output by the first synchronous differentiation circuit 42 to which the phase modulation signal epm is supplied from the first displacement amount detection unit 13 via the waveform shaping circuit 41 is supplied to the K input terminal of the JK flip-flop 44. The differential output from the second synchronous differentiation circuit 43 to which the reference signal MODS is supplied is supplied to the J input terminal of the JK flip-flop 44, whereby the phase amount φm corresponding to the absolute position within the wavelength λm at the time ti. A pulse width modulation signal PWSm having (ti) is output from the JK flip-flop 44.
[0025]
Here, as shown in FIG. 8, the operation timing of the first phase comparison unit 14 is a phase modulation signal output from the first displacement amount detection unit 13 every sampling period T (= 1 / f). The phase amount φm (ti) corresponding to the absolute position x (ti) within the wavelength λm at the time ti of the pulse width modulation signal PWSm set by epm and reset by the reference signal MODS is expressed by the following equation (2). Can be expressed as
[0026]
φm (ti) = 2π × x (ti) / λm (2)
The pulse width modulation signal PWSm obtained by the first phase comparison unit 14 has a first gate with a clock pulse M × f M (= 1,024) times the carrier frequency f provided by the timing signal generation unit 18. Input to the circuit 15.
[0027]
Here, the pulse width modulation signal PWSm having the phase amount φm (ti) is a signal whose pulse width changes corresponding to the position x within the wavelength λm, and has reached the maximum phase amount 2π, that is, the maximum displacement amount λm. Since the pulse width is equal to the sampling period T (= 1 / f), the first gate circuit 15 uses the M × f clock frequency to reduce the phase amount φm (ti) to 1 / M resolution, that is, , Displacement x (ti) can be detected as a pulse train SAPm with a resolution of 1 μm (= λm / M = 1,024 / 1,024), and the pulse train SAPm converted by the first gate circuit 15 is sent to the arithmetic unit 17. Entered.
[0028]
The absolute encoder 100 includes a second scale 21 formed by recording a scale having a wavelength λa (= 963.8 μm) in parallel with the first scale 11 on a scale member (not shown), and the second scale 21. A second detection head 22 composed of a two-channel MR sensor facing the scale 21 and arranged with a 90 ° phase difference (λa / 4) with respect to the wavelength λa is provided. The second detection head 22 is formed integrally with the first scale detection head 12, and due to relative movement with the second scale 21, the relative movement period is about 90 ° with 2πx / λa. Two systems of sinusoidal signals 2πx / λa having a phase difference (where x is a displacement amount), and two systems of sinusoidal signals Sin (2πx / λa) and Cos (2πx / λa) having a phase difference of 90 ° are output. .
[0029]
The two sine wave signals Sin (2πx / λa) and Cos (2πx / λa) obtained by the second detection head 22 are the second displacement having the same configuration as the first displacement amount detection unit 13. Guided to the quantity detector 23.
[0030]
The second displacement amount detection unit 23 outputs a phase modulation signal epa whose phase changes by 2π for each displacement amount λa, as shown in the following equation (3).
[0031]
epa = Ep2 × Sin (ωt + 2πx / λa) (3)
Where ω = 2πf, x = position within wavelength λa
This phase modulation signal epi is compared in phase with the phase modulation signal epm output from the first displacement amount detection unit 13 in the second phase comparison unit 24 having the same configuration as the first phase comparison unit 14. . The second phase comparison unit 24 is a pulse corresponding to the phase difference between the phase modulation signal epa output from the second displacement amount detection unit 23 and the phase modulation signal epm output from the first displacement amount detection unit 13. The width modulation signal PWS2 is output.
[0032]
As is clear from the above equation (2), the phase difference φa (ti) of the phase modulation signal can be expressed by the following equation (4).
[0033]
φa (ti) = 2π × x (ti) / λa (4)
Therefore, the phase difference Δφ (= φa (ti) −φm (ti)) is
Δφ = φa (ti) −φm (ti)
= 2π × x (ti) (1 / λa−1 / λm) (5)
Here, since the wavelength λm of the first scale 11 and the wavelength λa of the second scale 21 are selected to have a relationship of 16λm = 17λa, Δφ (ti) is
Δφ = 2π × x (ti) / 16λm
= 2π × x (ti) / 17λa (5) ′
This is a repetitive signal that coincides when the position reaches 16 times (N = 16) the wavelength λm of the first scale.
[0034]
FIG. 9 shows the relationship between the phase amount φm detected by the first displacement detector 13 and the phase amount φa detected by the second displacement detector 9 in the absolute measurement section.
[0035]
FIG. 10 shows the phase difference Δφ between φa and φm and the phase amount φm detected by the first displacement detection unit 13 for an arbitrary position x in the 16-fold section of the wavelength λm of the first scale. This shows the relationship with (= φa−φm). As shown in FIG. 10, the phase amount φm is 2π for each period of λm, and Δφ is 2π / 16 for each period of λm.
[0036]
Therefore, the phase difference Δφ is divided by a phase amount corresponding to one cycle of λm (hereinafter referred to as interval phase amount) 2π / 16, and the λm address Aj in the absolute measurement interval where x is located is detected as the quotient. it can.
[0037]
Here, the phase difference Δφ is guided to the second gate circuit 25 having the same configuration as the first gate circuit 15 together with the clock pulse M × f M times the carrier frequency f, and corresponds to the phase difference Δφ. Converted to the pulse train SAPa.
[0038]
Therefore, the section phase amount 2π / 16 is equivalent to the section pulse number M / 16, and by dividing the pulse train SAPa by the section pulse number M / 16, the absolute position of λm unit where x is located as the quotient. That is, the λm address Aj within 16λm can be detected.
[0039]
Table 1 below shows the relationship between Δφ and λm address Aj when the displacement amount x is changed from 0 to 16λm and the number of pulses ΔD corresponding to Δφ when 16λm = 17λa. It is shown.
[0040]
[Table 1]

[0041]
The computing unit 17 specifies the λm address Aj using the above relationship, and generates an absolute position in a 16λm section with a resolution of 1 μm by using a pulse train SAPm with a resolution of 1 μm generated based on the φm. For example, it can be realized by software processing using a microcomputer (CPU) 73 having a counter function (timer) as shown in FIG.
[0042]
Here, the basic principle relating to the detection of the λm address Aj, which is the main function of the calculation unit 17, will be described.
[0043]
The arithmetic unit 17 includes a pulse counter 71 that counts the pulse train SAPm with 1 μm resolution output from the first gate circuit 15 and a pulse counter 72 that counts the pulse train SAPa output from the second gate circuit 25. As a trigger signal for fetching the measured values of the pulse counters 71 and 72 into the CPU 73, a reference signal MODS as a set input of the pulse width modulation signals PWSm and PWSa is supplied from the timing signal generator 18 to the interrupt terminal to the CPU 73. ing.
[0044]
When the reference signal MODS corresponding to the reset timing of the phase amount φm (ti) shown in the above equation (2) is input at the sampling time ti, the CPU 73 starts an interrupt operation and the first time measured by the pulse counter 71 is measured. The data Dm (ti) corresponding to the absolute position within the wavelength λm of the scale 11 and the measured value ΔD (ti) of the pulse train SAPm corresponding to the phase difference Δφ of the two phase modulation signals measured by the pulse counter 72 are The data is taken into the CPU 73 and stored in a predetermined memory area in the RAM 74.
[0045]
Next, the CPU 73 stores the difference ΔD of these count values corresponding to the phase difference Δφ in a predetermined area in the RAM 74, and divides by the number of section pulses M / 16, and uses the λm address Aj as the quotient. Then, the obtained Aj is multiplied by λm, and the absolute value Dm (ti) in λm at timing ti is added to generate an absolute position with a resolution of 1 μm over the entire 16λm section.
[0046]
The absolute position generated in this way is stored in a predetermined memory area in the RAM 74 and is output to the outside as measurement data, for example.
[0047]
By the way, the operation of the arithmetic unit 17 has been described in an ideal state. In an actual system, incompleteness of a detected signal, for example, a shift of DC superimposed on each of the detection heads 12 and 22 is detected. Furthermore, a deviation from the 90 ° phase of the balanced modulation signal output from the heads of both channels, that is, a detection error caused by a phase deviation of the output, that is, occurs when detecting within the wavelength λm with high resolution. Since the interpolation error is included and these interpolation errors are superimposed on the switching unit of the λm address Aj, an erroneous operation may occur in the Aj determination of the λm address.
[0048]
Here, the relationship between the above-described deviation and the interpolation error will be described using mathematical expressions.
[0049]
As a specific embodiment of the present invention, the case where an MR sensor is used as a detection head has been described. As is well known, since a mask is manufactured based on lithography in the MR sensor, the distance between the heads of both channels can be accurately set to 90 ° (λ / 4) with respect to the wavelength λ.
[0050]
Therefore, among the error factors described above, the influence of “output phase shift” can be ignored.
[0051]
Hereinafter, in order to simplify the description, an interpolation error caused by a shift in DC superimposed on the detection head and a shift in output level will be described.
[0052]
The phase modulation signals shown in the above equations (1) and (2) are theoretical equations based on the premise that an ideal signal can be obtained from a two-channel detection head.
[0053]
However, in an actual system, the deviation of the DC superimposed on each detection head or the deviation of the output level causes a deviation from the complete phase modulation signal as shown in the equations (1) and (2). An insertion error occurs.
[0054]
Assuming that the deviation of DC and the deviation of output level are general expressions and the balanced modulation signals obtained from the CH1 detection head and the CH2 detection head on the scale of wavelength λ are e1 and e2, respectively, the balanced modulation signals e1 and e2 are
e1 = {A + (1 + a) × Sin (2πx / λ)} × Cosωt (6)
e2 = {B + (1 + b) × Cos (2πx / λ)} × Sinωt (7)
However,
a: Deviation from reference value of CH1 output, b: Deviation from reference value of CH2 output
A: Superimposition of DC on CH1 head, B: Superimposition of DC on CH2 head
In order to simplify the equation, if 2πX / λ = X and ωt = T, then equations (6) and (7) are
e1 = {A + (1 + a) × SinX} × CosT (6) ′
e2 = {B + (1 + b) × CosX} × SinT (7) ′
It can be expressed as.
[0055]
Next, an interpolation error pattern in the case of a single error factor will be described.
[0056]
Here, in order to simplify the description, the case where there is a deviation only in DC and the case where there is a deviation only in output level will be described separately.
(1) When there is a deviation only in DC (no deviation in output level)
The phase modulation signal epm (D) when there is a deviation only in DC is a signal obtained by adding a = b = 0 in the equations (6) ′ and (7) ′, and is expressed as follows: be able to.
[0057]

When an approximate expression is obtained assuming that A and B are sufficiently smaller than 1,

However,
δ = tan ^-1 (A / B)
θ = tan ^-1 {(A + SinX) / (B + CosX)} (9)
Here, {} in the expression (8) represents the envelope of the phase modulation signal epm (D), and the amplitude whose phase is delayed by δ from the reference position of X with respect to the reference amplitude 1 is √ (A ² + B ² It can be seen that this is a sinusoidal signal having a first-order component with respect to X, with the ripples of
[0058]
In addition, θ in equation (9) represents the amount of displacement, so the theoretical value when there is no error {θ = tan ^-1 Subtracting from (SinX / CosX)} represents an error when the phase modulation signal deviates from the ideal, that is, when there is a deviation of DC.
[0059]
Now, if an interpolation error within one wavelength ranging from 0 to 2π with respect to X is ΔX (D), the interpolation error is ΔX (D)
ΔX (D) = tan ^-1 {(A + SinX) / (B + CosX)}
-Tan ^-1 (SinX / CosX) (10)
So here,
α = {(A + SinX) / (B + CosX)},
β = SinX / CosX
If you
tan ^-1 α-tan ^-1 β = tan ^-1 {(Α−β) / (1 + α × β)}
ΔX (D) = tan ^-1 {(A * CosX-B * SinX) / (1 + A * SinX + B * CosX)}
It becomes. Further, when an approximate expression of ΔX (D) is obtained considering that (A × SinX + B × CosX) is sufficiently smaller than 1,
ΔX (D) = tan ^-1 (A x CosX-B x SinX) (11)
It becomes.
[0060]
Furthermore, in a small angle, it can be approximated as tan {ΔX (D)} = ΔX (D).
[0061]
Therefore, an approximate expression of an error when a deviation occurs only in DC can be expressed as follows.
[0062]

However,
γ = tan ^-1 (A / B) (13)
As apparent from the equation (12), the interpolation error when only the DC is shifted is proportional to the magnitude of the shift, and the position where the amplitude is maximum (minimum) (hereinafter referred to as phase) is It turns out that it becomes a sine wave of one period with respect to one period of X which changes with deviation | shift, ie, wavelength (lambda).
[0063]
(2) When there is a deviation only in the output level (no deviation in DC)
The phase modulation signal epm (G) when there is a shift only in the output level is a signal obtained by adding A = B = 0 in the equations (6) ′ and (7) ′. For simplicity, a = 0 If b is a change with respect to the output on the CH1 side, it can be expressed as follows.
[0064]

Here, when an approximate value is obtained assuming that b is sufficiently smaller than 1,

However,
θ '= tan ^-1 {SinX / (1 + b) × CosX} (15)
Here, {} in the expression (14) represents the envelope of the phase modulation signal epm (G), and the sign of b, that is, the output of CH1 and CH2 with respect to the reference amplitude (1 + b / 2). Depending on the magnitude relationship, it can be seen that the amplitude of the polarity is (b / 2) and a sinusoidal signal having a second order component is superimposed on X.
[0065]
In addition, θ ′ in the equation (15) represents the amount of displacement, so the theoretical value when there is no error {θ = tan ^-1 Subtracting from (SinX / CosX)} represents an interpolation error when the phase modulation signal deviates from the ideal, that is, when the outputs of both channels deviate. Now, 0 to 2π with respect to X If the interpolation error within one wavelength is ΔX (G), the interpolation error ΔX (G) is
ΔX (G) = tan ^-1 {SinX / (1 + b) × CosX} -tan ^-1 (SinX / CosX)
And where
α ′ = {SinX / (1 + b) × CosX},
β ′ = (SinX / CosX)
If you
ΔX (G) = tan ^-1 (Α′−β ′) / (1 + α ′ × β ′)
Therefore,

It becomes.
[0066]
Furthermore, when b is sufficiently smaller than 1, tan {ΔX (D)} = ΔX (D) and the denominator can be approximated to 1.
[0067]
Therefore, an approximate expression of an error when a deviation occurs in the output level can be expressed as follows.
[0068]
ΔX (G) = (− b / 2 × Cos2X) (17)
As can be seen from the equation (17), the interpolation error ΔX (G) when the deviation occurs only in the output level has an amplitude and a polarity (output magnitude relationship), an absolute value of b and a polarity (both It can be seen that the signal is proportional to the magnitude of the output of the channel), has a constant phase, and changes in a sinusoidal shape having a component of two periods with respect to one period of wavelength λ.
[0069]
In the case of a single error factor, the vibration period and the position where the maximum (mountain) and minimum (valley) appear within one period of the wavelength λ are shown in Table 2 below. From Table 2, it can be seen that each factor is independent.
[0070]
[Table 2]

[0071]
Next, a case where two or more error factors are superimposed will be described.
[0072]
Similarly, when two or more error factors are superimposed, there are a case where a DC shift occurs between two heads, and a case where a DC shift and an output level shift occur with respect to one of the heads.
[0073]
(1) The positive and negative DC shifts of the CH1 head (A> 0, A <0) and the positive and negative DC shifts of the CH2 head (B> 0, B <0) are superimposed in four ways. The interpolation error of 1 has a component of one period with respect to the wavelength λ, and the position of the maximum (mountain) and the minimum (valley) changes depending on each combination. Examples of interpolation errors when the coefficient is set to 1 are shown in FIGS.
[0074]
(2) Four types of positive and negative DC deviations of the CH1 head and output level deviations are superimposed, and FIG. 16 shows an example of interpolation error when the coefficient is set to 1 in the same manner as described above. Although there are two periodic components with respect to the wavelength λ, the positions of a pair of local maximum (mountain) and local minimum (valley) change depending on each combination.
[0075]
(3) Four types of positive and negative DC shifts of the CH2 head and output level shifts are superimposed, and FIG. 17 shows an example of the interpolation error when the coefficient is set to 1 as described above. Similar to (2), it has a component of two periods with respect to the wavelength λ, the position of the maximum (crest) and the minimum (valley) changes depending on the combination, and is independent of (2).
[0076]
Next, a case where three or more factors are superimposed will be described.
[0077]
The superposition of three or more error factors is when an output level shift and a DC shift between two heads occur simultaneously.
[0078]
(1) In the state where the output of CH1 is smaller than the output of CH2, there are four types in which the positive and negative DC shifts of the CH1 head and CH2 head are superimposed, and FIG. 18 shows an example of the interpolation error. There is only one maximum (mountain) or minimum (valley) within the wavelength λ, and its position changes depending on each combination.
[0079]
(2) FIG. 19 shows an example of four types of interpolation errors when the positive and negative DC shifts of the CH1 head and the CH2 head are superimposed while the output of CH1 is larger than the output of CH2. Similar to the above, there is only one maximum (mountain) or minimum (valley) in the wave λ, and is independent of (2) above.
[0080]
Table 3 below shows a list of interpolation error periods due to two or more error factors, and positions of local maxima (peaks) and (valleys).
[0081]
[Table 3]

[0082]
As described above, when the phase modulation signal deviates from the ideal state, a ripple component having a specific period and phase is superimposed on the envelope of the phase modulation signal according to the cause of the deviation, and the period unique to the cause of the deviation. It can be seen that an interpolation error having a phase occurs.
[0083]
As is clear from these, the switching unit of the λm address Aj is a position corresponding to an integral multiple of the wavelength λm of the first scale 11, but for the wavelength λa of the second scale 21, the λm address Aj Since each corresponds to a position shifted by an integral multiple of 1/16 of the wavelength λa, the interpolation error of the wavelength λa of the second scale 21 is superimposed on Δφ detected by the λm address switching unit. Therefore, a reversal phenomenon of the phase difference Δφ occurs, and based on the relationship shown in Table 1 above, simply obtaining the detected Δφ with respect to the interval phase amount φZ (= 2π / 16), the λm address Aj A malfunction will occur in detection.
[0084]
FIG. 20 is a graph in which the DC error in the positive direction occurs in the CH1 detection head and the CH2 detection head of the second scale 21, that is, the interpolation error shown in the equation (12) is plotted over one period of the wavelength λa. Table 4 below shows the correspondence with the λm address Aj where the interpolation error is superimposed.
[0085]
[Table 4]

[0086]
Next, the actual system applied in the present invention, that is, the configuration and operation of the arithmetic unit 17 that can detect the λm address Aj correctly even when an interpolation error occurs will be described in detail.
[0087]
First, with reference to FIG. 21, the basic principle for determining the λm address Aj in the present invention will be described.
[0088]
(1) The phase difference Δφ detected at the switching unit of λm address Aj, that is, both ends of λm, varies by the phase amount φiaa corresponding to the interpolation error Iea of the wavelength λa of the second scale 21. Therefore, there is a possibility that the λm address Aj that should be detected as j is actually detected before or after that, that is, as j−1 or j + 1.
[0089]
(2) However, in the region that is further inward by the phase amount φiem corresponding to the interpolation error Iem in the first scale with respect to the region that has the possibility of erroneous detection, the region is not affected by the interpolation error. There is an area where the λm address Aj can be uniquely determined.
[0090]
(3) Further, in (1) above, the position within the wavelength λm to which the phase φm detected on the first scale belongs is, for example, the left side (x By adding a determination condition as to whether it is ≦ λm / 2) or located on the right side (x> λm / 2) and correcting based on the determination result, the λm address (Aj) can be correctly determined.
[0091]
That is, in the switching unit of the λm address Aj, a hysteresis is given to the determination region of the λm address (Aj), and in the hysteresis region, a determination condition based on the position of the first scale λm is added, thereby interpolating errors. It is possible to detect the λm address Aj that is not affected by the error and that does not malfunction.
[0092]
Next, the setting of hysteresis and other determination conditions to be given to the switching unit of λm address Aj in the present invention will be described by applying specific numerical values.
[0093]
As described above, the phase difference Δφ is detected as the pulse number ΔD, and the section phase amount φZ (= 2π / 16) corresponds to the section pulse number ND (= 64).
[0094]
Since the wavelengths of the first scale and the second scale are substantially equal, assuming that the interpolation error assumed in the system is substantially equal and the amplitude is Ie, the phase amount φie and the number of pulses Die corresponding to the interpolation error Can be expressed as:
[0095]
φie = 2π × (Ie / λm) (17)
Die = M × φi / 2π = 1,024 × (Ie / λm) (18)
That is, in the switching unit of the λm address (Aj), the pulse number ΔD may vary by Die.
[0096]
Here, the interpolation error rate Ie / λm with respect to the wavelength λ is normally about 1%, and the Die at that time is about 10 pulses from the equation (18). In the switching unit of the λm address Aj, the second It can be considered that a fluctuation of ± 10 pulses occurs with respect to the interval pulse number ND (= 64) due to the interpolation error of the scale λa. Furthermore, if the pulse number Die due to the influence of the interpolation error in the first scale 11 is also 10 pulses, an area within ± 20 pulses with respect to the section pulse number ND (= 64) obtained by adding this value, that is, In the section where the pulse number ΔD is 20 or more (= 0 + 20) to less than 44 (= 64-20), the area where the λm address Aj can be uniquely determined without being affected by the interpolation error (hereinafter referred to as unconditional determination area). It is.
[0097]
On the outside of the unconditional determination area, that is, in the lower area where ΔD is 0 or more and less than 20, and the upper area where ΔD is 44 or more and less than 64, the phase amount φm detected in the first scale is λm An intermediate value, that is, a condition for discriminating whether the Dm is 512 pulses (1,024 / 2) or not is added, and the λm address Aj temporarily determined according to the determination result may be corrected.
[0098]
Next, a procedure for determining the λm address (Aj) in the present invention will be described according to the processing flow shown in FIG.
[0099]
First, in the first step S1, when the reference signal MODS is input at the sampling time ti, the CPU 73 starts an interrupt operation, and the count value Dm (ti) of the first pulse counter 71 and the second pulse counter are counted. The count value ΔD (ti) of 72 is fetched and stored in a predetermined memory area of the RAM 74.
[0100]
In the next step S2, the CPU 73 divides ΔD by the number of interval pulses NZ (= 64), sets the quotient j as a λm address candidate, and sets the remainder Res as data for λm address determination. Store in the area.
[0101]
In the next step S3, the CPU 73 classifies into three areas according to the value of the remainder Res, and determines the λm address Aj by the next determination process.
[0102]
That is, when 0 ≦ Res <20 (step S41), it is determined whether or not Dm (ti) exceeds the median value 512 (step S51).
[0103]
If the determination result in step S51 is “Yes”, that is, Dm (ti) ≧ 512, 1 is subtracted from the quotient j calculated in step S2, and this value is set as the λm address Aj (step S61). If the determination result is “No”, that is, Dm (ti) <512, Aj detected in step S2 is directly used as the λm address Aj (step S62).
[0104]
When 20 ≦ Res <44 (step S42), the quotient j detected in step S2 is directly used as the λm address Aj (step S62).
[0105]
Further, when 44 ≦ Res <64 (step S43), it is determined whether or not Dm (ti) is smaller than the median value 512.
[0106]
If the determination result in step S53 is “Yes”, that is, Dm (ti) <512, 1 is added to the quotient j calculated in step S2, and this value is set as the λm address Aj (step S63). If the determination result is “No”, that is, Dm (ti) ≧ 512, Aj detected in step S2 is directly used as the λm address Aj (step S62).
[0107]
In the next step S7, after the λm address Aj is determined by the above procedure, the CPU 73 multiplies the λm address Aj by the wavelength λm, and then adds the above Dm (ti) to obtain the absolute position X in the 16λm section at time ti. (Ti) is generated.
[0108]
Thereafter, the same procedure (S1 to S7) is repeated for each sampling time, and the absolute position in the 16λm section is obtained. Further, the generated absolute position X (ti) can be used as necessary, for example, as a display or output to an external system as measurement data.
[0109]
By the way, in the phase detection type absolute encoder 100 according to the present invention, the λm address switching unit is provided with a determination zone that anticipates the interpolation error assumed in the system in advance, thereby preventing a malfunction due to the detection of the λm address Aj. .
[0110]
However, even when the accuracy is high at the time of shipment, the interpolation error deteriorates due to, for example, the deterioration of the component parts or the change of the adjustment state, and exceeds the estimated amount of the interpolation error assumed in advance. As a result, the λm address Aj An error occurs in the detection.
[0111]
Furthermore, if these interpolation errors are estimated too much, the above N cannot be increased, and as a result, there is a problem that the measurement section is narrowed in absolute.
[0112]
Therefore, the absolute encoder 100 monitors deterioration of the interpolation error and the like and activates an automatic adjustment function as necessary to ensure that the interpolation error is always kept at an appropriate level or less. The value can be set, so that a high-precision and wide absolute measurement section can be realized.
[0113]
Here, the monitoring and automatic adjustment of the interpolation error in the absolute encoder 100 will be described.
[0114]
That is, the absolute encoder 100 includes, for example, a λa pulse generation unit 16 configured as shown in FIG. The λa pulse generation unit 16 receives a D-type flip-flop 62 to which the phase modulation signal epa is supplied from the second displacement amount detection unit 23 via the waveform shaping circuit 61, and an output of the D-type flip-flop 62 is input. The timing differentiation circuit 63 supplies the reference signal MODS to the D flip-flop 62 and the clock pulse M × f to the synchronization differentiation circuit 63. The λa pulse output from the λa pulse generation unit 16 is input to the CPU 73 of the calculation unit 17 shown in FIG. 11, and functions as an interrupt signal for the CPU 73 as necessary. A part of the RAM 74 of the arithmetic unit 17 is composed of a nonvolatile memory.
[0115]
Next, the basic principle for realizing the above-described interpolation error monitoring and automatic adjustment will be described.
[0116]
In general, the scales (wavelengths) of the first scale 11 and the second scale 21 are formed (recorded) on the basis of a light wave interferometer, so that they have extremely high accuracy. Further, as apparent from the above formulas 12 and 17, the period of X, that is, the pitch of the λa pulse generated for each movement of the wavelength λa, is always constant regardless of the shift of the DC or the shift of the output level. Value.
[0117]
2) The first λ pulse generator 16 shown in FIG. 1 generates a λa pulse for each movement of the wavelength λa of the second scale 21 in the absolute measurement section. A λa pulse is generated. Here, in order to identify the λa pulse generated for each address of the wavelength λa of the second scale 21, the λa pulse generated when the λa address (Ak) changes from k to k + 1 is expressed as λa (k). Then, the position within the wavelength λm of the first scale 11 corresponding to the generation position of the λa pulse (λa (k)) is (16−k) × λm / 17.
[0118]
Next, if the recording conditions of the scales 11 and 21 are constant and the scale material maintains a constant quality, the interpolation error within each wavelength in a relatively narrow range, for example, an absolute measurement section, is similar. It is thought to have.
[0119]
Therefore, the 17 λa pulses (λa (k)) generated at an accurate pitch during the movement over the absolute measurement section correspond to sampling within one wavelength of the first scale 11 at λm / 17 pitch. .
[0120]
Here, the following Table 5 shows the position (16−k) × λm / 17 within the wavelength λm sampled by the λa pulse (λa (k)) and the theoretical value Dmt (k) of the number of pulses corresponding to the position. Actually, a deviation occurs due to an interpolation error in each scale.
[0121]
[Table 5]

[0122]
Therefore, by detecting the difference between the actually measured value and the theoretical value shown in Table 5, the interpolation error in each of the scales 11 and 21 is measured.
[0123]
By the way, in this embodiment, the inside of the wavelength λm of the first scale 11 is sampled 17 times, and the period of the interpolation error caused by the deviation of DC or the deviation of output level as described above is the wavelength λ. However, it contains only two period components at the maximum. Therefore, as is clear from the sampling theorem, the interpolation error associated with these factors can be completely evaluated.
[0124]
Next, a specific method for monitoring and automatic adjustment of the interpolation error in the absolute encoder 100 will be described.
[0125]
When automatic adjustment of the interpolation error is performed, the interpolation error is monitored at an appropriate timing, the monitoring result is compared with a predetermined judgment value, and automatic adjustment is started to maintain the allowable interpolation error for the product. Need to be configured.
[0126]
1) Setting of “standard value of interpolation error”
As already described, the ratio of the interpolation error (Ie / λ) to the wavelength λ is usually about 1%. Therefore, if the interpolation error standard value Ir larger than the above value and acceptable as a product is set to 2%, for example, the standard value Ir of the interpolation error is 0.002 × λ. In this embodiment, For example, it can be set to 20 μm.
[0127]
The standard value Ir of this interpolation error may be stored in a predetermined non-volatile area in the RAM 74 of the computing unit 17 so that it can be referred to when monitoring the interpolation error. Conveniently stored as the number of pulses Dr.
[0128]
2) Acquisition of interpolation error at the time of shipment
As described above, the “standard value Ir of interpolation error” allowed for a product can be used as a determination value when automatic adjustment is started.
[0129]
However, even if the product is the same, the state of the interpolation error differs for each product. In addition, it is always best to monitor the change over time and activate the automatic adjustment function based on the monitoring result. From the viewpoint of maintaining the accuracy of the product, an interpolation error at the time of shipment, which is assumed to be the best state of the product, is acquired, and a value obtained by adding a predetermined margin to the acquired interpolation error is set as a determination value Sometimes it is more convenient.
[0130]
Hereinafter, the acquisition of the interpolation error performed in the production process or at the time of shipment will be described.
[0131]
Procedure 11 When the optimal electrical adjustment in the first scale 11 is completed, a predetermined signal is given to the control input of the arithmetic unit 17 to activate the “interpolation error storage mode”. In this interpolation error storage mode, for example, the CPU 73 is interrupted by the λa pulse λa (k), and the count value of the pulse counter 72, that is, the absolute value within the wavelength λm corresponding to the λa pulse λa (k) is obtained. Conveniently configured to read.
[0132]
Procedure 12 Theoretical absolute position within the wavelength λm of the first scale 11 corresponding to the λa pulse λa (k), that is, the count value Dm (of the pulse counter 71 corresponding to (16−k) × λm / 17 k).
[0133]
Step 13 Next, the count value ΔD (k) of the pulse counter 72 is fetched, the λa address k where the λa (k) is generated is specified according to the λm address specifying procedure, and the specified k is used. Then, the interpolation error Ies (k) at the position is measured as a difference from the theoretical value Dmt (k) at the position within the wavelength λm corresponding to λa (k), that is, at the position (16−k) × λm / 17. Then, the address k is stored as an index in a predetermined area of the nonvolatile memory (hereinafter referred to as a shipping interpolation error table Tis (k)).
[0134]
Step 14 The above steps 11 to 13 are repeated for all λa (k), that is, k = 0 to 16, and an interpolation error of λm / 17 units is acquired over the entire region within the wavelength λm. Stored in the time interpolation error table Tis (k).
[0135]
If the interpolation error measured as described above is larger than the interpolation error Ir that should be allowed for the product, the electrical adjustment is performed again and then the procedures 11 to 14 are repeated.
[0136]
Here, the identification of the λa address is performed by exchanging the relationship between λm and λa, the number of pulses corresponding to one section of the λa address is M / 17 (about 60), and Dm (k) and ΔD (k ) To obtain the absolute value Da (k) within the wavelength λa, the λa address can be specified in the same procedure as the λm address described above, but the λa (k) pulse is input. The absolute position X (k) in the N-fold section of the wavelength λm at this time is obtained, and the X (k) is divided by λa.
[0137]
3) Monitoring of interpolation error
The monitoring of the interpolation error is normally performed on the user side. For example, the user automatically activates the “interpolation error monitoring mode” through a keyboard or the like when the power is turned on, and the interpolation error of the product in advance. Standard value Ir or a value Iec (k) obtained by adding a predetermined margin to a value Ies (k) acquired at the time of shipment and stored in the shipping interpolation error table Tis (k) is stored in a predetermined area in the RAM 74. Conveniently, it is stored in the provided interpolation error determination value table Tic (k) and used as a determination value for each predetermined position (λm / 17 pitch) within the wavelength λm. Further, in this mode, it is convenient to configure so that normal absolute position measurement by interruption with the reference signal MODS and interpolation error monitoring by interruption with the λa pulse are executed in parallel. Any selection can be made without departing from the spirit of the invention.
[0138]
Here, only the interpolation error monitoring function will be described for the sake of simplicity.
[0139]
Procedure 21 When λa (k) is input to the calculation unit 17, the CPU 73 starts an interrupt operation, captures the count value Dm (k) of the pulse counter 71, and stores it in the temporary storage area of the RAM 74.
[0140]
Procedure 22 Simultaneously, the count value ΔD (k) of the pulse counter 72 is fetched, the λa address k is specified, and the position within the wavelength λm corresponding to the λa (k) using the specified k, that is, (16 -K) The interpolation error Ie (k) at the position is detected as a difference from the theoretical value Dt (k) at the position of [lambda] m / 17, and a predetermined area (hereinafter referred to as a monitoring interpolation error table) using the address k as an index. Ti (k)).
[0141]
Step 23 Next, the measured interpolation error Ie (k) is used as an interpolation error determination value Iec (k) stored in a predetermined area in the interpolation error determination value table Tic (k) of the RAM 74. When the standard value is exceeded, the interpolation error error Er (k) is written in the corresponding position in the monitoring interpolation error table Ti (k).
[0142]
Procedure 24 The above procedures 21 to 23 are executed for all (all) λa (k), and the interpolation error Ie (k) and the interpolation at all the positions of λm / 17 units within the wavelength λin. The error error Er (k) is measured, and the result is stored in the monitoring interpolation error table Ti (k).
[0143]
4) Judgment of interpolation error error
Ie (k) measured in the monitoring of the interpolation error in 3) is compared with the interpolation error determination value Iec (k), and the interpolation is performed immediately when the first interpolation error error Er (k) is detected. It is also possible to determine that the error has exceeded the determination value and immediately output an alarm signal. However, if the determination is made with only one interpolation error Er (k), it may be confused with, for example, a temporary reaction due to noise or the like, resulting in an excessive alarm signal. As will be described later, the interpolation error is usually a signal that vibrates in one cycle or two cycles within the wavelength λ.
[0144]
Therefore, by making the determination as follows, it is possible to reliably determine an interpolation error error without malfunction.
[0145]
The number of interpolation error errors Er (k) that occur within one period of the wavelength λm is counted, and the following processing is performed according to the number.
[0146]
Process 1 When the error occurs in only one place, it is assumed that the interpolation error error has occurred only in the next cycle when the error error Er (k) occurs at the same position, that is, the same λa (k). judge.
[0147]
Process 2 When an interpolation error error Er (k) occurs at two or more locations within the wavelength λm, it is determined that an interpolation error error has occurred unconditionally.
5) Identification of interpolation error factors
If it is determined that the error is an interpolation error, the CPU 73 starts an error factor specifying process as a preparation for providing an appropriate adjustment signal ADJ to the electronic volumes DC1, DC2, and GAJ for correcting the error factor.
[0148]
As an error factor identification method, the interpolation error Ie (k) detected over one period of the wavelength λa and stored in the monitoring interpolation error table Ti (k) is analyzed by Fourier analysis, and one period for the wavelength λm. Alternatively, the error factor can be specified by decomposing it into the level and phase of the two period components.
[0149]
However, the fact that the error factors are extremely limited and that the period and phase of the interpolation error within the wavelength λ have an independent pattern for each error factor as shown in Table 2 is used. , Using the relationship between the period of the interpolation error detected within the wavelength λ and the position where the maximum (crest) and minimum (valley) of the interpolation error are, It is convenient to specify the error factor from the viewpoint of cost and processing speed.
[0150]
FIG. 24 shows a procedure for specifying the error factor, which will be described in detail below.
[0151]
First, with respect to the position where Er (k) occurs, the interpolation error Ie (k) is checked from the descending order of k, that is, from the 0 side within the wavelength λm, and the maximum interpolation error Ie (k) is considered in consideration of the sign. ) Λa (k) max in which max is detected and λa (k) min in which the minimum interpolation error Ie (k) min is detected are searched (step S11).
[0152]
Next, a position within the wavelength λk is obtained from λa (k) max where the maximum interpolation error Ie (k) max is detected and λa (k) min where the minimum interpolation error Ie (k) min is detected. The first maximum value (mountain) and the first minimum value (valley) are stored in a predetermined storage area of the RAM 74 (step S12). Here, as already described, the position within the wavelength λm of the first scale 11 corresponding to λa (k) is obtained as (16−k) × λm / 17.
[0153]
Next, the number of the first maximum value (peak) and the first minimum value (valley) detected in step S12 is determined (step S13).
[0154]
When one each of the first maximum (mountain) and the first minimum value (valley) is detected, the process proceeds to step S14, and the first maximum (mountain), ie, Ie (k) max or the first When only one minimum (valley), ie, Ie (k) min is detected, it is determined that there are three factors, and the process proceeds to step S22.
[0155]
In the next step S14, the second maximum value (mountain) at which the second largest interpolation error Ie (k) max2 is detected at the position excluding the Ie (k) max and Ie (k) min and the second value are detected. The second minimum value (valley) in which the small interpolation error Ie (k) min2 is detected is searched for. Then, the positions of the second maximum value (mountain) and the second minimum value (valley) are obtained (step S15), the presence and the number thereof are determined (step S16), and the following processing is performed.
[0156]
That is, when neither Ie (k) max2 nor Ie (k) min2 is detected, it is determined that only one period component, that is, a DC shift, is the first maximum value stored in the RAM 74. (Crest) and the value of the first minimum value (valley) are analyzed using the rules of Tables 2 and 3, and the number of heads on which DC is superimposed (single or both) and the direction of magnetization (A and B) Sign) is determined (step S17).
[0157]
When Ie (k) max2 and Ie (k) min2 are detected, that is, when two period components are detected, the detected second maximum value (peak) and second minimum value ( Valley) is stored in a predetermined storage area of the RAM 74, and the magnitude relationship between these two sets of local maximum values and local minimum values stored in the RAM 74 is determined (step S18). It is determined whether it is due to two factors or not (step S19).
[0158]
In step S20, when there is a clear difference between the levels of Ie (k) max and Ie (k) max2 and Ie (k) min and Ie (k) min2, the DC deviation of the head on one side and the deviation of the output level are detected. Are analyzed and the positions of the first maximum value (peak) and the minimum value (valley) stored in the RAM 74 are analyzed using the rules shown in Table 3, and the head is magnetized. (A or B) and the direction of deviation (positive or negative) and the direction of deviation of output level (sign of b) are specified.
[0159]
In step S21, when the levels of Ie (k) max and Ie (k) max2 and Ie (k) min and Ie (k) min2, that is, when the interpolation error is substantially equal, the difference between the CH1 output level and the CH2 output level is caused. It is determined as a factor, and the two sets of local maximum values (peaks) and local minimum values (valleys) are analyzed using the rules in Table 2 to specify the output level magnitude relationship, that is, the error factor b code.
[0160]
In step S22, when only one of Ie (k) max (maximum (mountain)) or Ie (k) min (minimum (valley)) can be detected, the DC deviation of the heads of both channels and the deviation of the output level are detected. The direction (polarity of A, B, b) is specified.
[0161]
Then, after storing the error factors (A, B, b) specified by the above procedure and the direction (positive / negative) of the deviation in a predetermined storage area of the RAM 74 (step S23), an alarm signal is output (step S23). Step S24).
[0162]
By the way, in the explanation so far, for the sake of easy understanding, the absolute value of the deviation is assumed to be equal with respect to the DC deviation of the two-channel head.
[0163]
In practice, however, the absolute values of the deviations are often not equal. In this case, the position of the maximum value (peak) and the minimum value (valley) is λ, as is apparent from the above equation (13). / 8 (π / 4) unit position, and before and after that, for example, in the case of sampling at a wavelength λ / 16 unit as in this example, before and after the maximum (peak) and minimum (valley) Must be taken into account.
[0164]
Further, in this embodiment, Table 3 is shown as a table in which the wavelength λ is divided into 16 parts. However, it is divided into 17 parts and calculated as a rounded value, and the maximum (crest) or minimum (valley) at the rounded position is calculated. ) Is calculated theoretically, and the value may be used as a table for comparison.
[0165]
5) Automatic adjustment
When the alarm signal is output, the system activates the “automatic adjustment mode” and can start the automatic adjustment based on the information of the specified error factor.
[0166]
As a method of shifting to the “automatic adjustment mode”, for example, a method of automatically shifting in conjunction with the output of an alarm signal, a user or another device, etc., recognizing that an alarm signal has been output, There is a method of starting via an external command signal or the like, and it may be selected as necessary.
[0167]
Basically, as described above, the error factor and the direction of the deviation are clear. Therefore, the cause is specified, the adjustment signal ADJ is given from the CPU 73, and the electronic volumes DC1, DC2, GAJ are arranged in the direction to cancel the cause. Should be controlled.
[0168]
The automatic adjustment process when the “automatic adjustment mode” is activated will be described below.
[0169]
In the actual adjustment operation, it is necessary to secure a table area for primary storage for storing the interpolation error at the time of adjustment, but this is omitted for the sake of simplicity.
[0170]
The automatic adjustment process is performed according to the following procedures 31 to 35 in principle.
[0171]
Step 31 When the “automatic adjustment mode” is activated, the CPU 73 refers to the RAM 74 storing the error factors, identifies the electronic volumes DC1, DC2, GAJ to be adjusted from the error factors, and adjusts the adjustment signal. Selection signals CS1, CS and CS3 are supplied to the electronic volume selected and designated by the ADJ selection signal SLT.
[0172]
Procedure 32 Next, referring to the displacement direction from the RAM 74, a wiper control signal (U / D, INC) for changing the wiper position of the electronic volumes DC1, DC2, GAJ in a direction to compensate for the displacement direction is given.
[0173]
Step 33 After that, the interpolation error Ie (k) max at the maximum (crest) position specified by the error factor and the interpolation error Ie (k) min at the minimum (valley) position are measured. Then, if the interpolation error at the position changes in a decreasing direction, the detected interpolation error is compared with a value stored in the interpolation error table Tis (k) at the time of shipment, and the difference Step 32 is repeated until is minimized. Further, when the detected interpolation error changes in the increasing direction, a wiper control signal is given in the direction of moving the wiper of the electronic volume in the opposite direction, and the detected interpolation error is set at the time of shipment. Compared with the interpolation error table Tis (k), the procedure 32 is repeated until the difference is minimized.
[0174]
Procedure 34 When the interpolation error in the best adjustment state can reproduce the interpolation error Ies (k) at the time of shipment by the above procedures 31 to 33, an end signal is output and the “automatic adjustment mode” is ended.
[0175]
The basic processing for automatic adjustment has been described above. However, in practice, a plurality of error factors may be superimposed, and it is necessary to consider the order of adjustment. In this case, it is convenient to adjust in the following order in consideration of the characteristics of the error pattern.
[0176]
(A) When the deviation of DC and the deviation of output level are superimposed, first, the deviation of DC is adjusted, and the interpolation error Ie (k) max at the first maximum value (peak) and the second maximum value are adjusted. The difference between the interpolation error Ie (k) max2 at the value (peak) or the interpolation error Ie (k) min at the first minimum value (valley) and the interpolation error Ie (k) min2 at the second minimum value It is convenient to move to the adjustment of the output level after adjusting so as to minimize the output.
[0177]
(B) In the case where the DC deviation of the two channels and the deviation of the output level are superimposed, first, the output level is adjusted, the interpolation error pattern due to the deviation of DG is reproduced, and then one cycle with respect to the wavelength λ. After that, it is convenient to specify and adjust the head having the larger DC deviation from the phase of the maximum point, and finally adjust the DC deviation of the head having the smaller DC deviation.
[0178]
Further, when these plural factors are superimposed, the pattern of the interpolation error Ies (k) at the time of shipment is not necessarily the case when the best condition can be adjusted and the adjusted result satisfies the standard value. For example, there may be a case where the error period and the positions of (peaks) and minimums (valleys) do not match.
[0179]
In this case, the interpolation error at the time of adjustment to the best state is used as a determination reference when automatic adjustment is activated next time. For example, the content of the shipping interpolation error table Tis (k) is used as the best adjustment. The values can be rewritten with the values of the interpolation error table Ti (k) in the state, and adjusted with this value as a reference next time, and can be freely handled without departing from the gist of the present invention.
[0180]
As a specific embodiment of the present invention, in order to obtain a desired resolution R of 1 μm, the wavelength λm of the first scale 11 is set to 1,024 times the desired resolution of 1 μm, and the carrier frequency f of the phase modulation signal is 1 , 024 (M = 1,024) times as many clock pulses as the first scale 11 in order to detect the absolute position of the 16th section (N = 16) of the wavelength λm of the first scale 11. A case has been described in which the relationship between the wavelength λm of the second scale 21 and the wavelength λa of the second scale 21 is set to a binary series of 16 × λm = 17 × input a.
[0181]
However, as described above, in order to obtain the desired resolution λm / M, the wavelength λm of the first scale and the design conditions M and N are decimal values, for example, λm = 1,000, M = 1000, N = 20. Needless to say, it can be set as follows.
[0182]
In this embodiment, the phase difference Δφ between the first scale and the second scale is directly calculated using the second phase comparison unit 24.
[0183]
However, as in the first phase comparison unit 14, the second phase comparison unit 24 also detects the absolute position within the wavelength λa of the second scale 21 by phase comparison between the phase modulation signal epa and the reference signal. The phase difference is detected as φa (ti), and then detected as the pulse number Da (ti) through the second gate circuit 25 and the pulse counter 72, and the arithmetic unit 17 calculates ΔD = Da (ti) −. Needless to say, it can be detected as Dm (ti).
[0184]
Further, in the present invention, the example in which only the first scale 11 serving as a reference for the measurement, that is, the first displacement amount detection unit 13 is adjusted has been described. However, as shown by the broken line in FIG. The second displacement amount detector 23 is configured to be automatically adjusted using the λm pulse output from the second λ pulse generator 26 by supplying the output from the displacement detector 13 and the reference signal. You can also.
[0185]
Furthermore, in monitoring the interpolation error, various methods can be applied without departing from the gist of the present invention, such as when the system is powered on or when the monitoring is started at a regular timing by a timer or the like.
[0186]
Also, at the start of automatic adjustment, various modes such as not only automatically starting the adjustment mode together with the output of the alarm signal but also starting automatic adjustment when the user inputs an external control signal, etc. It is applicable to.
[0187]
Furthermore, although the magnetic type has been described in the above-described embodiment, the present invention is not limited to the magnetic type. For example, as in the optical method, the two channels orthogonal sine corresponding to the relative movement of the scale and the head. Similarly, in a system in which a wave signal is output, if the phase-modulated signal is extracted by performing quadrature modulation on the detected two-phase sine wave signal with a signal having a carrier frequency f and then adding the same, Applicable.
[0188]
Furthermore, in the present invention, the application to the linear encoder has been described, but the present invention can also be applied to absolute detection of the rotation amount by making the repeated section coincide with the circumferential length.
[0189]
【The invention's effect】
As described above, according to the present invention, since the number of tracks is small, the structure of the system is simplified, and an absolute position detection system can be realized with a small size and low cost.
[0190]
In addition, since the position detection system uses a phase modulation signal, the frequency M × f of the clock pulse is set higher than the carrier frequency f of the phase modulation signal, that is, high resolution can be easily achieved simply by increasing M. Can be realized. Further, since the AC coupling method is not affected by DC drift as in the differential transformer method, for example, measurement can be performed immediately without warm-up. Furthermore, in the differential transformer system, fluctuations in the output level immediately cause errors, but in this system, even if the output level from the two-channel detection head changes, the error will not occur unless the output ratio varies. Therefore, a stable and highly accurate linear encoder can be realized.
[0191]
In addition to monitoring the interpolation error with this system alone without the help of other measurement systems, analyze the measured interpolation error and identify the cause of the interpolation error. Can do.
[0192]
Therefore, not only can an alarm signal be output when the interpolation error exceeds a predetermined level, but also an automatic adjustment can be performed in response to a command from a self-supporting or external control signal when the alarm signal is detected. Therefore, it is possible to detect and correct in advance the deterioration of the interpolation error caused by the change in the mounting state or the deterioration of the parts, and it is possible to realize highly accurate detection over a long period of time.
[0193]
Further, since the interpolation error can be maintained at a predetermined level, the determination zone in the switching region of the λ address set in order to prevent malfunction of the λ address due to the interpolation error can be set to the minimum. Can be selected greatly, and the absolute measurement range can be expanded.
[Brief description of the drawings]
FIG. 1 is a block diagram showing a configuration of an absolute encoder according to the present invention.
FIG. 2 is a block diagram showing a configuration of a first displacement amount detection unit in the absolute encoder.
FIG. 3 is a diagram showing a configuration example of an MR sensor constituting a detection head in the absolute encoder.
FIG. 4 is a diagram illustrating a configuration example of a summing amplifier provided in the first displacement amount detection unit.
FIG. 5 is a diagram illustrating a configuration example of an electronic volume provided in the first displacement amount detection unit.
FIG. 6 is a diagram illustrating a configuration example of an adjustment signal control circuit provided in the first displacement amount detection unit.
FIG. 7 is a block diagram showing a configuration of a first phase comparison unit in the absolute encoder.
FIG. 8 is a diagram illustrating an operation timing of the first phase comparison unit.
FIG. 9 is a diagram showing a relationship between a phase amount φm detected by a first displacement amount detection circuit and a phase amount φa detected by a second displacement amount detection circuit in an absolute measurement section.
10 is a diagram showing a relationship between a phase amount φm detected by a first displacement amount detection unit and a phase difference Δφ (= φa−φm) between φa and φm. FIG.
FIG. 11 is a block diagram illustrating a configuration example of a calculation unit.
FIG. 12 is a diagram illustrating an example of an interpolation error in which positive and negative DC shifts of the CH1 head and the CH2 head are superimposed.
FIG. 13 is a diagram illustrating an example of an interpolation error in which positive and negative DC shifts of a CH1 head and a CH2 head are superimposed.
FIG. 14 is a diagram illustrating an example of an interpolation error in which positive and negative DC shifts of a CH1 head and a CH2 head are superimposed.
FIG. 15 is a diagram illustrating an example of an interpolation error in which positive and negative DC shifts of the CH1 head and the CH2 head are superimposed.
FIG. 16 is a diagram illustrating an example of an interpolation error in which positive and negative DC shifts of the CH1 head and the CH2 head are superimposed.
FIG. 17 is a diagram illustrating an example of an interpolation error in which positive and negative DC shifts of a CH1 head and a CH2 head are superimposed.
FIG. 18 is a diagram illustrating an example of an interpolation error in which positive and negative DC shifts of the CH1 head and the CH2 head are superimposed.
FIG. 19 is a diagram illustrating an example of an interpolation error in which positive and negative DC shifts of the CH1 head and the CH2 head are superimposed.
FIG. 20 is a diagram illustrating an example of an interpolation error ΔX of the first scale when the DC bias is multiplied.
FIG. 21 is a diagram for explaining a basic concept relating to determination of a λm address Aj.
FIG. 22 is a flowchart showing a procedure for determining a λm address Aj.
FIG. 23 is a block diagram illustrating a configuration example of a λ pulse generation unit provided in the absolute encoder.
FIG. 24 is a flowchart showing a procedure for specifying an error factor.
[Explanation of symbols]
11 First scale, 12A, 12B MR sensor, 12 First detection head, 12A ₁ , 12A ₂ , 12A ₃ , 12A ₄ , 12B ₁ , 12B ₂ , 12B ₃ , 12B ₄ MR element, 13 first displacement amount detection unit, 14 first phase comparison unit, 15 first gate circuit, 17 calculation unit, 18 timing signal generation unit, 21 second scale, 22 second detection head, 23 Second displacement amount detection unit, 24 Second phase comparison unit, 25 Second gate circuit, 31, 32 multiplier, 33 Addition amplifier, 34 Adjustment signal control circuit, 34A decoder, 41 Waveform shaping circuit, 42, 43 1st and 2nd synchronous differentiation circuit, 44 JK flip-flop, 61 waveform shaping circuit, 62 D-type flip-flop, 63 synchronous differentiation circuit, 71, 72 pulse counter, 73 CPU, 74 RAM, 91 up / down counter, 92 Position decoder, 93 variable resistor, DC1, DC2, GAJ electronic volume, 100 absolute encoder

Claims (5)

A first scale having a scale of wavelength λm, a first detection head for detecting a signal of the first scale, and a relative displacement amount of the first scale and the first detection head as a carrier A first displacement amount detecting means configured to be extracted as a phase modulation signal having a frequency f; a second scale having a scale having a wavelength λa different from that of the first scale; and a signal of the second scale. And a second displacement amount detecting means configured to extract a relative displacement amount between the second detection head for detecting the second scale and the second detection head as a phase modulation signal of the carrier frequency f. And detecting the phase difference between the phase modulation signal obtained from the first displacement amount detection means and the phase modulation signal obtained from the second displacement amount detection means, and using the phase difference, the first scale is detected. Wavelength λm N times wavelength To obtain a periodic signal that repeats, in absolute encoder configured to detect the first scale N times the interval of the wavelength of the absolute,
A pulse corresponding to the absolute position of the first detection head within the wavelength λm of the first scale by comparing the phase of the phase modulation signal of the carrier frequency f obtained from the first displacement amount detection means with the reference signal of the frequency f. First phase comparison means for generating a width modulation signal;
A resolution corresponding to the absolute position of the first detection head within the wavelength λm of the first scale by interpolating a clock pulse M times the carrier frequency f into the pulse width modulation signal generated by the first phase comparison means. a first gate circuit for converting to a pulse train of λm / M;
Second phase comparison means for generating a pulse width modulation signal corresponding to the phase difference between the phase modulation signal output from the first displacement amount detection means and the phase modulation signal output from the second displacement amount detection means; ,
A second gate circuit for interpolating a clock pulse M times the carrier frequency f into the pulse width modulation signal generated by the second phase comparison means, and converting it to a pulse train corresponding to the phase difference;
Pulse generating means for comparing the second displacement amount detecting means and the reference signal to generate a pulse for each displacement amount corresponding to the wavelength λa of the second scale (hereinafter referred to as λa pulse);
The output of the second gate circuit is used as the address of the wavelength λm of the first scale (hereinafter referred to as λm address), and the pulse train output from the first gate circuit is counted to obtain the first By obtaining an absolute position within the wavelength λm of one scale and combining it with the specified λm address, detection is performed with a resolution λm / M over an N-fold section of the wavelength λm of the first scale, The absolute position within the wavelength λm of the first scale corresponding to the generation position of the λa pulse output from the second pulse generation means is measured, and the deviation of the measured value from the theoretical value is detected as an interpolation error. And storing the detected interpolation error and a preset standard value in a storage means, analyzing the detected interpolation error, and providing an electronic adjustment provided in the first displacement amount detecting means. Provide an adjustment signal to the means And an arithmetic means for performing electrical adjustment so as to minimize the interpolation error in the first scale. The calculation means detects a position within the wavelength λm of the first scale corresponding to the generation position of the λa pulse output from the pulse generation means corresponding to the movement in the absolute measurement section, and the detected value And the theoretical value at the position is regarded as a representative value of the interpolation error within the wavelength λm in the first scale, and the interpolation error superimposed on the first scale by Fourier analysis of the representative value. The period and the phase component are specified, the magnitude of the error factor and the direction of deviation are specified by the specified period and phase, and the first displacement is in a direction to cancel out the magnitude of the specified error factor and the direction of deviation. 2. The absolute air according to claim 1, wherein an adjustment signal is supplied to an electronic adjustment means provided in the quantity detection means to perform automatic adjustment so as to minimize an interpolation error in the first scale. Coder. The calculation means detects a position within the wavelength λm of the first scale corresponding to the generation position of the λa pulse output from the pulse generation means corresponding to the movement in the absolute measurement section, and the detected value And the theoretical value at the position is regarded as a representative value of the interpolation error within the wavelength λm in the first scale, and the representative value takes a maximum value or a minimum value at the wavelength λm of the first scale, and The position of the error is detected, the magnitude of the error factor and the direction of the deviation are specified on the basis of the period and the position where the maximum value or minimum value specific to the cause of the interpolation error is generated, and the size of the specified error factor An automatic adjustment is performed so as to minimize the interpolation error in the first scale by supplying an adjustment signal to the electronic adjustment means provided in the first displacement amount detection means in a direction to cancel out the direction of deviation. Characterize Motomeko 1 absolute encoder according. The arithmetic means monitors an interpolation error at the time of measurement based on a value stored in a predetermined memory as a preset standard value, and detects that the detected interpolation error exceeds the standard value. The detected interpolation error is analyzed at the time, and an adjustment signal is supplied to the electronic adjustment means of the first displacement amount detection means based on the analysis result, so that the detected interpolation error is minimized. The absolute encoder according to claim 1, wherein the automatic adjustment mode is executed. The arithmetic means monitors an interpolation error at the time of measurement with reference to a value stored in a predetermined memory as a preset standard value, and detects that the detected interpolation error exceeds the standard value. And outputting an alarm signal, analyzing the interpolation error based on the alarm signal or an externally supplied command signal, and using the analysis result to the electronic adjustment means of the first displacement amount detection means. 2. The absolute encoder according to claim 1, wherein an automatic adjustment mode is provided to supply an adjustment signal and minimize the detected interpolation error.

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