JP5272482B2 - Rotor phase estimator for synchronous motor - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a rotor phase estimating device, which is used for a drive controller for a synchronous motor having a permanent magnet as a rotor and is provided with a rotor phase estimating means which is based on the same dimension state observer with a simpler structure and guarantees a stable phase estimation and specifies an eigenvalue-convergence speed of observer. <P>SOLUTION: The phase estimating device is composed of the same dimension state observer using corresponding values of drive voltage and drive current as input signals and using a stator interlinkage magnetic flux and a rotor flux as state variables. The observer gain used for the same dimension state observer is specified in such a way that eigenvalue-convergence speed of observer can be specified and stable convergence is guaranteed. <P>COPYRIGHT: (C)2009,JPO&amp;INPIT

Description

Detailed Description of the Invention

  The present invention relates to a rotor phase estimation device used in a sensorless drive control device for a synchronous motor having a permanent magnet in a rotor. In particular, the present invention relates to a rotor phase device that estimates a rotor phase using a voltage / current for driving an electric motor. In the description of the present invention, phase is used synonymously with position.

The following are typical prior inventions related to the present invention.
Koko, Machiko Tomioka, Kou Nakano, and Kim Tokai: “Position sensorless control of brushless DC motor by adaptive observer”, IEEJ Transactions D, 113, 5, pp. 579-586 (1993-5) Shinji Shinnaka and Kosuke Sano: “A new sensorless vector control method of PMSM using a fixed coordinate fourth-order same-dimensional state observer combined with integral feedback type velocity estimation method”, IEEJ Transactions D, 125, 8, pp. 11-27. 830-831 (2005-8) Yoshihiko Kanehara and Toshiyuki Kaiya: “Control Device for Synchronous Motor”, International Publication Number WO2002 / 091558, Japanese Patent Application No. 2002-565151 (2001-4-24) Yoshihiko Kanehara: “Position sensorless control of PM motor using adaptive observer on rotating coordinates”, IEEJ Transactions D, 123, 5, pp. 600-609 (2003-5) Shinnaka Shinji and Daisuke I: “Applicability of Generalized Integral PLL Method to Sensorless Vector Control of Synchronous Motors Using Fourth Order Dimensional State Observer”, IEEJ Transactions D, 124, 11, pp. 11-27. 1164-1165 (2004-11) Yasuhiro Yamamoto: “Synchronous motor control device”, Japanese Patent No. 4032845 (2002-6-26) Hiroshi Kogo and Tsutomu Mita: “Introduction to System Control Theory”, pp. 173-178, Jikkyo Publishing (1979-12)

  The γδ general coordinate system is a general coordinate system including the αβ fixed coordinate system and the dq synchronous coordinate system as special cases (see FIG. 1). In the present invention, among the γδ general coordinate system, a coordinate system aiming at synchronization without phase difference to the dq synchronous coordinate system is referred to as a γδ quasi-synchronous coordinate system.

  As a rotor phase estimation method using voltage and current for driving for sensorless vector control of a synchronous motor having a permanent magnet in a rotor (hereinafter abbreviated as a synchronous motor), a one-dimensional state observer (hereinafter referred to as appropriate) A method of estimating a rotor magnetic flux using a state observer (or abbreviated as an observer) and extracting rotor phase information included in the rotor magnetic flux estimated value is proposed in the documents (1) to (5). Yes.

  As the same-dimensional state observer on the αβ fixed coordinate system, as shown in the literatures (1) and (2), the object such as a saddle intended only for the non-salient synchronous motor, non-salient / saliency Regardless of whether there is a new medium or the like for synchronous motors. The same-dimensional state observer of 楊 uses a stator current and a rotor magnetic flux as state variables. On the other hand, the new one-dimensional state observer uses the stator reaction magnetic flux and the rotor magnetic flux as state variables. As is well known to those skilled in the art, in the case of a non-salient-pole synchronous motor, the stator current and the stator reaction magnetic flux are in a proportional relationship. From the viewpoint of the state observer structure, the new state observer is It includes a state observer.

  As the same-dimensional state observer on the γδ quasi-synchronous coordinate system aiming at synchronization with the rotor N pole, there are those by Kanehara, Shinnaka, etc., as shown in documents (3) to (5). Both the Kanbara and Shinnaka state observers can be applied regardless of whether they are non-salient or salient, and stator reaction magnetic flux and rotor magnetic flux are used as state variables. As described above, the same-dimensional state observer of the prior invention selects the stator reaction magnetic flux (or the stator current equivalent to the observer structure) and the rotor magnetic flux as state variables. As is well known to those skilled in the art, in the case of a non-salient pole synchronous motor, the stator current and the stator reaction magnetic flux are in a proportional relationship, and the state observer uses the stator current and the rotor magnetic flux as state variables, A state observer that uses stator reaction magnetic flux and rotor magnetic flux as state variables is substantially equivalent from the viewpoint of observer structure theory.

ここに、Ｉは２ｘ２単位行列であり、Ｊは次式で定義された２ｘ２交代行列である。

（３）式で定義された同相インダクタンス（非突極同期電動機の場合には、固定子インダクタンスと同一）である。

In general, the estimation performance of a state observer is dominated by the matrix-type observer gain used for this. Tsuji et al. Also proposed an observer gain design method for non-salient-pole synchronous motors through Reference (1).ゲ イ ン 's gain design method uses the following matrix gains for the observer gain G _i (2 × 2 matrix) corresponding to the stator reaction magnetic flux (proportional to the stator current) and the observer gain G _m (2 × 2 matrix) corresponding to the rotor magnetic flux: It is something to design.

Here, I is a 2 × 2 unit matrix, and J is a 2 × 2 alternating matrix defined by the following equation.

In-phase inductance defined by equation (3) (same as stator inductance in the case of a non-salient pole synchronous motor).

As is clear from the equation (1), the gain design method of the kite has the following features.
(A) Keep the diagonal elements of the two types of 2 × 2 matrix observer gains at a non-zero constant value. At this time, the diagonal element is proportional to the stator resistance.
(B) On the other hand, the opposite diagonal elements of the two types of observer gains are both increased in accordance with the speed improvement. At this time, the inverse diagonal element is proportional to the stator inductance.
However, unfortunately, the observer using the observer observer gain is destabilized in the vicinity of the rated speed as pointed out in documents (1), (4), and (6).

ここに、ω_ｍａｘ，ω_ｈは、各々、最大駆動電気速度、オブザーバの等価的帯域幅（設計パラメータ）である。In order to solve the above destabilization problem, Yamamoto et al. Have proposed a new gain design method for a saddle state observer for non-salient synchronous motors through document (6). One of Yamamoto's gain design methods is given by the following equation.

Here, ω _max and ω _h are the maximum drive electric speed and the equivalent bandwidth (design parameter) of the observer, respectively.

Another one of Yamamoto's gain design methods is given by the following equation.

The features of Yamamoto's two design methods are summarized as follows.
(A) The diagonal elements of the gain G _i (2 × 2 matrix) corresponding to the stator reaction magnetic flux (proportional to the stator current) are always kept constant. On the other hand, all elements of the rotor magnetic flux-corresponding gain G _m (2 × 2 matrix) are reduced according to the speed improvement. The speed-corresponding change in the rotor magnetic flux-corresponding gain is the opposite of the heel gain.
Polar diagonal element g _m1 of (b) the rotor flux corresponding gain G _m is dominated by polar rotation, in the case of normal rotation positive, if negative rotation becomes negative.
(C) The inverse diagonal element g _m2 of the rotor magnetic flux-corresponding gain G _m is always positive regardless of whether the rotation is positive or negative.
In the case of Yamamoto's gain, as will be elucidated when explaining the effect of the present invention of claim 2 (see paragraphs 33 and 34), stable phase estimation cannot be achieved.

In contrast to the above-mentioned gain design method of Yamamoto, Kanehara has shown a gain design method based on stability theory through references (3) and (4). Kanehara's gain design method uses the Kalman filter gain design method introduced in Document (7) and the like, and has the following features.
(A) In order to determine each element of the observer gains G _i and G _m (2 × 2 matrix), it is necessary to solve a Riccati equation for the purpose of averaging a certain frequency characteristic over the entire drive region. The computational load of this solution over the entire drive area is enormous. Therefore, the solution work needs to be completed separately prior to the execution of the observer.
(B) Each element obtained by solving the Riccati equation is basically non-zero and changes with speed. In other words, each element of the observer gain has a functional relationship with the speed, and the function at this time is basically a non-linear function with respect to the speed. For this reason, it is necessary to store the gain element obtained by the solution in a table (referred to as a gain table in Document (3)).
(C) In the case of using this observer gain, in principle, stable convergence of the state variable estimated value is guaranteed in the entire drive region.

  The gain of Kanehara has an excellent merit that it can guarantee stable convergence of the state observer, unlike the gain of Sakai and the gain of Yamamoto. However, as described above, enormous pre-calculations that require table formation are required. According to Kanehara's gain, the fastest observer eigenvalue (corresponding to the pole of the time-invariant observer) can be roughly specified through one design parameter when solving the Riccati equation, but for the slowest observer eigenvalue, Like the gain and Yamamoto's gain, it has the disadvantage of being unknown. The slowest observer eigenvalue dominates the convergence speed, which is the practical performance of stable convergence, and this has the disadvantage that it is unknown as well as the 楊 gain and the Yamamoto gain.

  The present invention has been made under the above-mentioned background, and its object is to be used in a drive control device for a synchronous motor having a permanent magnet in a rotor, and a rotor based on a one-dimensional state observer with a simpler structure. An object of the present invention is to provide a rotor phase estimation device having phase estimation means. Rotor phase estimation means that does not require complicated Riccati equation solution and observer gain table, can guarantee stable phase estimation, and can specify the observer eigenvalue and convergence speed. It is to provide an estimation device.

  In order to achieve the above object, the invention of claim 1 is a rotor phase estimation device used in a drive control device for a synchronous motor having a permanent magnet in a rotor, wherein an equivalent value of a drive voltage is It comprises at least means for generating a rotor phase estimation value by constituting at least a state observer having a corresponding value of the driving current as an input signal and having stator interlinkage magnetic flux and rotor magnetic flux as state variables.

ただし、δ_ｉ１２，δ_ｉ２１：ゼロを代表値とする任意の微小値、Ｌ_ｄ：ｄ軸インダクタンス、Ｌ_ｑ：

The invention of claim 2 is used in a drive control device for a synchronous motor having a permanent magnet in a rotor, and an equivalent value of a drive voltage and an equivalent value of a drive current are input signals, and the stator linkage Rotor comprising at least means for generating a rotor phase estimation value by constituting at least a state observer having any one of two combinations of magnetic flux and rotor magnetic flux, stator reaction magnetic flux and rotor magnetic flux as state variables A phase estimator for two types of 2 × 2 matrix observer gains G _i and G _m used for the state observer, wherein the matrix observer gain corresponding to the stator linkage magnetic flux or the stator reaction magnetic flux is G _i , and the rotor When the matrix observer gain corresponding to the magnetic flux is G _m , two types of matrix observer gains are set according to the following equations.

However, δ _i12 , δ _i21 : Arbitrary minute values with zero as a representative value, L _d : d-axis inductance, L _q :

  A third aspect of the present invention is the rotor phase estimation device according to the second aspect, wherein all elements of the two-type matrix observer gain are made constant regardless of the speed.

  The invention according to claim 4 is the rotor phase estimation device according to claim 2, wherein the eigenvalue of the state observer changes substantially in proportion to the absolute velocity value in an intended operation region of the state observer. All elements of the two-type matrix observer gain are characterized by being proportionally determined by multiplying an absolute value of a speed equivalent value by a constant proportional coefficient corresponding to each element.

  The term “equivalent value” used above means a true value of the signal, a good approximate value of the true value, an estimated value, or a signal having a good correlation with the true value.

（７）〜（１４）式において、２ｘ１ベクトルν_１、ｉ_１、φ_１は、それぞれ固定子の電圧、電流、鎖交磁束を意味している。２ｘ１ベクトルφ_ｉ、φ_ｍは固定子鎖交磁束φ_１を構成する成分を示しており、φ_ｉは固定子電流ｉ_１に起因した固定子反作用磁束であり、またφ_ｍは回転子永久磁石に起因した回転子磁束である。なお、本発明では、固定子鎖交磁束と固定子磁束とは同義で使用する。また記号ｓは、微分演算子またはラプラス演算子として使用する。Hereinafter, the effects of the present invention will be described clearly with reference to the drawings and mathematical expressions. Consider a γδ general coordinate system rotating at an arbitrary speed ω _γ as shown in FIG. Further, it is assumed that the rotor N pole of the synchronous motor has a phase θ _γ instantaneously with respect to the main axis γ-axis. A mathematical model (circuit equation) of the synchronous motor on the γδ general coordinate system is described by the following equations (7) to (14).

In the equations (7) to (14), 2 × 1 vectors ν ₁ , i ₁ , and φ ₁ mean stator voltage, current, and flux linkage. 2 × 1 vectors φ _i and φ _m indicate components constituting the stator linkage magnetic flux φ ₁ , φ _i is a stator reaction magnetic flux caused by the stator current i ₁ , and φ _m is a rotor permanent magnet. The rotor magnetic flux resulting from In the present invention, the stator interlinkage magnetic flux and the stator magnetic flux are used synonymously. The symbol s is used as a differential operator or a Laplace operator.

  The circuit equations shown in the equations (7) to (14) can be amended into a state space expression. In prior inventions, stator current and rotor flux or stator reaction flux and rotor flux have been selected as state variables for the state space. In the case of a non-salient pole, the stator current and the stator reaction magnetic flux are in a proportional relationship, and it can be said that the prior invention generally selects the stator reaction magnetic flux and the rotor magnetic flux as state variables. From prior invention results, the selection of state variables for the same-dimensional state observer for synchronous motors tends to give recognition that there is only a combination of stator reaction flux and rotor flux. However, this recognition is not correct. State space expression using the stator magnetic flux and the rotor magnetic flux as state variables is possible. As a result, there exists a one-dimensional state observer for estimating these states. This is shown below based on the present invention.

（１５）式における記号∧は、関連信号の推定値を意味する。固定子電流と同推定値の偏

ーバゲインは、状態オブザーバによる状態変数（すなわち、固定子磁束、回転子磁束）の推定性能を支配する重要な設計事項である。A one-dimensional state observer using the stator magnetic flux and the rotor magnetic flux as state variables can be newly constructed using a D-factor D (•, •) of a 2 × 2 matrix as follows.

The symbol ∧ in equation (15) means an estimated value of the related signal. The bias of the same estimated value as the stator current

The observer gain is an important design item that governs the estimation performance of the state variables (that is, the stator magnetic flux and the rotor magnetic flux) by the state observer.

The same-dimensional state observer (hereinafter referred to simply as A-type state observer) of equation (15) is configured on a γδ general coordinate system including an αβ fixed coordinate system and a γδ quasi-synchronous coordinate system as special cases. Please be careful about points. This state observer can be illustrated as shown in FIG. The structure of FIG. 2 corresponds to the simplest A-type vector block diagram (A-type dynamic simulator) of the synchronous motor shown in the following documents (8) and (9), and is the same dimension of the synchronous motor. It is the simplest state observer. As is apparent from the above description, according to the first aspect of the present invention, the same-dimensional state observer having the simplest structure is constructed, and as a result, the rotor phase estimation having the simple-structure rotor phase estimating means based thereon. The effect that the apparatus can be configured is obtained.
Shinji Shinnaka: “Block diagram of AC rotating machine using vector signal”, IEEJ Transactions D, 118, 6, pp. 715-723 (1998-6) Shinji Shinnaka: “Block diagram of general-purpose vector signal of AC motor with stator iron loss”, IEEJ Transactions D, 120, 12, pp. 1492-1500 (2000-12)

（１６）式の状態オブザーバは、図３のように図示することができる。図３の構造は、文献（８）、（９）に示された同期電動機のＢ形ベクトルブロック線図（Ｂ形動的シミュレータ）に対応するものである。以降では、本状態オブザーバを、簡単に、Ｂ形状態オブザーバと呼称する。Next, the effect of the invention of claim 2 will be described. The circuit equations shown in the equations (7) to (14) can be revised to a state space expression in which the stator reaction magnetic flux and the rotor magnetic flux are selected as the state variables. As a result, the same-dimensional state observer corresponding to this state space expression can be comprised. This observer on the γδ general coordinate system can be constructed as follows using the D factor.

The state observer of equation (16) can be illustrated as shown in FIG. The structure of FIG. 3 corresponds to the B-type vector block diagram (B-type dynamic simulator) of the synchronous motor shown in documents (8) and (9). Hereinafter, this state observer is simply referred to as a B-type state observer.

When the state variable is estimated by the A-type state observer of the equation (15) or the B-type state observer of the equation (16), whether or not the state variable estimated value converges to the same true value is determined by two types of 2 × 2 observer gains G. _i and G _m are dominantly affected. It can be understood from the fact that the same-dimensional state observer of the equation (15) or (16) corresponds to the A type and B type vector block diagrams (or A type and B type dynamic simulators) of the same synchronous motor. As described above, the same observer gain is used for both the same-dimensional state observers of the expressions (15) and (16). Further, as understood from the fact that these observers are configured on the γδ general coordinate system including the αβ fixed coordinate system and the dq synchronous coordinate system, these observer gains are represented on the αβ fixed coordinate system on the dq synchronous coordinate system. Any of the above can be designed.

（１６）式のＢ形状態オブザーバに用いる２種の２ｘ２オブザーバゲインＧ_ｉ，Ｇ_ｍを（６ａ）式の形とし、このとき、ゼロを代表値とする任意の微小値δ_ｉ１２，δ_ｉ１２に関しては、代表値

として）評価すると、この誤差方程式は次式のように整理される。

（１８）式の誤差方程式の特性を支配する４ｘ４行列Ａ_ｒの固有値λは、次の４次方程式の根として与えられる。

Based on the above recognition, regarding the B-type state observer of Equation (16), it is assumed that the true value is known for all signals other than the state variable to be estimated, and this is the method for designing the observer gain that guarantees stable convergence. This will be described based on the invention. First, the state variable estimation error is defined as follows.

Two types of 2 × 2 observer gains G _i and G _m used for the B-type state observer of the equation (16) are in the form of the equation (6a), and at this time, regarding arbitrary minute values δ _i12 and δ _{i12 having} zero as a representative value Is the representative value

As a result, this error equation is organized as follows:

(18) the eigenvalues λ of 4x4 matrix A _r which governs the characteristics of the type of error equations are given as the root of the next fourth order equations.

（１８ｂ）式を、（２０）式の２式に用いることにより、次式を得る。

（２１）式に固定子抵抗の性質Ｒ_１＞０を考慮すると、磁束推定値の安定収束の十分条件として請求項２で規定した条件である（６ｂ）式を得る。All necessary and sufficient condition that the real part is a negative eigenvalues λ of the matrix A _r in the case of omega _2n ≠ 0, by applying a stability criterion method Hurwitz (19) to the polynomial expression, be determined it can. From this, the following equation (20) is obtained as a condition for stabilizing the error equation.

The following formula is obtained by using the formula (18b) in the two formulas (20).

When considering the property R ₁ > 0 of the stator resistance in the equation (21), the equation (6b) which is the condition defined in claim 2 is obtained as a sufficient condition for stable convergence of the estimated magnetic flux.

  As apparent from the above description, the expression (6b) for the matrix observer gain having the structure of the expression (6a) according to the invention of claim 2 is a sufficient condition for stable convergence. That is, it is not a necessary and sufficient condition. Special attention should be paid to this point. It should be pointed out that stable convergence is possible even if the elements of the matrix observer gain having the structure of equation (6a) are configured under conditions not defined by equation (6b).

  As is clear from the above analytical explanation, according to the invention of claim 2, in the same-dimensional state observer using the stator interlinkage magnetic flux and the rotor magnetic flux or the stator reaction magnetic flux and the rotor magnetic flux as the state variables. On the other hand, an effect that the estimated magnetic flux value can be stably converged to the same true value can be obtained. As a result, it is possible to obtain an effect that a rotor phase estimation device including means for generating an appropriate rotor phase estimation value can be configured.

（２２）式は、固定子側磁束対応のゲインの対角要素ｇ_ｉ１１、ｇ_ｉ２２を調整することにより、概略的ながら、オブザーバの収束速度を制御できることを示すものである。換言するならば、請求項２の発明によれば、固定子鎖交磁束と回転子磁束、または、固定子反作用磁束と回転子磁束を状態変数とする同一次元状態オブザーバに対し、磁束推定値を、その収束速度を制御した形で、同真値に収斂させることができる効果が得られるようになる。ひいては、適切な回転子位相推定値を生成する手段を備える回転子位相推定装置が構成できると言う効果が得られる。Strictly speaking, the convergence speed of the observer is the eigenvalue of the slowest mode among the four eigenvalues of the observer, in other words, the maximum value of the eigenvalue negative real part (the minimum value in the absolute value evaluation of the real part), Ruled. In general, it is governed by the average value of the real part. (19b), (18b

Expression (22) shows that the convergence speed of the observer can be controlled roughly by adjusting the diagonal elements g _i11 and g _i22 of the gain corresponding to the stator side magnetic flux. In other words, according to the invention of claim 2, the estimated magnetic flux value is obtained for the same-dimensional state observer with the stator interlinkage magnetic flux and the rotor magnetic flux or the stator reaction magnetic flux and the rotor magnetic flux as the state variables. Thus, an effect of converging to the same true value can be obtained by controlling the convergence speed. As a result, it is possible to obtain an effect that a rotor phase estimation device including means for generating an appropriate rotor phase estimation value can be configured.

As is well known to those skilled in the art, in the case of a non-salient pole synchronous motor, the stator reaction magnetic flux and the stator current are in a proportional relationship with the stator inductance as a proportional coefficient. Thus, for non-salient synchronous motor, in the case of constituting the same dimensions state observer for the stator current and the rotor flux and the state variable is a stator inductance observer gain G _i which defines according (6) The value obtained by dividing can be used as the observer gain corresponding to the stator current. Thus, the invention of claim 2 can be applied to a non-salient-pole synchronous motor when the same-dimensional state observer having the stator current and the rotor magnetic flux as the state variables is configured, and is intended. It is pointed out that.

ここに、δ_ｉは、ゼロを代表値とする任意の微小値である。対称簡略化した（２３）式は、（６）式に比較し、オブザーバゲインの選択範囲が小さくなっているが、対称簡略化に応じてより使い易いものとなっている。The defining expression for the observer gain in claim 2 (that is, the expression (6)) includes the following simplified defining expression in which the diagonal elements and the antidiagonal elements of the observer gain are defined by four parameters with good symmetry. I will point out that.

Here, δ _i is an arbitrary minute value having zero as a representative value. The symmetrically simplified expression (23) is smaller in the selection range of the observer gain than the expression (6), but is easier to use in accordance with the simplified symmetry.

The observer gain (23a) expression according to the present invention is partly similar to the 楊 gain expression (1a) and the Yamamoto gain expression (4a). The expression (1a) is used together with the conditional expression (1b), the expression (4a) is used together with the conditional expression (4b), and the expression (23a) is meaningful together with the expression (23b). Special attention should be paid to the fact that it is used and meaningful. These three gain methods are mutually exclusive. For example, the heel gain in the equation (1) is in an exclusive relationship with the equation (23) based on the present invention with respect to the inverse diagonal element of the stator side corresponding gain G _i . At 楊 gain, this inverse diagonal element is non-zero and increases proportionally as the speed estimate increases, but in the present invention, this inverse diagonal element always represents zero regardless of the speed increase. Keep the minute value as the value. If we consider that the intended drive region of the state observer is the medium to high speed region, these mean exclusive relationships.

(4) Yamamoto gain equation relates the diagonal elements and reverse diagonal elements of the rotor flux corresponding gain G _m, an exclusive relationship with that according to the present invention (23). In Yamamoto gain, the diagonal elements of the rotor flux corresponding gain G _m is the time of forward rotation takes positive values, induces unstable. On the other hand, in the equation (23) based on the present invention, this diagonal element always takes a negative value, including during positive rotation. Thereby, the stable convergence at the time of forward rotation is maintained. In Yamamoto gain, reverse diagonal elements of the rotor flux corresponding gain G _m is not polarity inversion corresponding to the speed, at all times, including during negative rotation, maintaining a positive polarity, inducing destabilization during negative rotation To do. On the other hand, in the equation (23) based on the present invention, the reverse diagonal element inverts its polarity according to the speed and takes a negative polarity during negative rotation. Thereby, the stable convergence at the time of negative rotation is maintained. As described above, the Yamamoto gain is anxious in both the positive rotation and the negative rotation, whereas in the case of the equation (23) that is exclusively related to the Yamamoto gain, in either the positive rotation or the negative rotation. Can guarantee stable convergence. In addition to the exclusive relationship described above, attention should be paid to the characteristics of the heel gain shown in paragraph 08 and the characteristics of the Yamamoto gain shown in paragraph 11.

Next, the effect of the invention of claim 3 will be described. As is clear from the analytical explanation using the equations (17) to (21), for example, the matrix observer gain is further simplified as the equation (23) and selected as the following equation with all elements as constants. It is also possible.

ここに、δ_ｉ，δ_ｍ偏は、ゼロを代表値とする微小定数である。Furthermore, it is also possible to select a shape in which the symmetry of the gain corresponding to the rotor magnetic flux is remarkably broken, as in the following formula (25) or (26).

Here, the δ _i and δ _m deviations are minute constants with zero as a representative value.

（２７）式より明らかなように、固定ゲインの場合には、パラメータの設定を介して、概略的ながら磁束推定値の収束速度（概ね一定値）を制御できる。When using the simplified fixed gain of the equations (24) to (26), the equation (22) is approximated as follows.

As is clear from the equation (27), in the case of a fixed gain, the convergence speed (substantially constant value) of the estimated magnetic flux value can be controlled roughly via the parameter setting.

  As can be understood from the examples of the expressions (24) to (26) and the expression (27), according to the invention of claim 2, the observer gain is made constant by making all elements of the matrix gain constant in the entire speed region. Can be remarkably simplified. As a result, the effect that the rotor phase estimation apparatus used for a drive control apparatus can be simplified is acquired. Even when all the elements of the matrix gain are constant, as is clear from the analytical explanation using the equations (17) to (21), the state variable estimated value (magnetic flux estimated value) is set to the same value. Stable convergence is guaranteed and the convergence speed is constant, but this can be controlled roughly.

（２８ｃ）式におけるω_ｍｉｎは、（６ｂ）式から得られる安定収束のための条件｜ｇ_ｍ１｜＋｜ｇ_ｍ２｜≠０を保証するために導入したものであり、オブザーバの所期動作領域の下限速度を示している。なお、請求項４の発明では、ゼロの行列要素は、ゼロ一定係数を速度相当値の絶対値に乗じたものとして、捕らえている。Next, the effect of the invention of claim 4 will be described. As is clear from the analytical explanation using the equations (17) to (21), all the elements of the matrix observer gain are set to a constant coefficient corresponding to the absolute value of the speed equivalent value in the intended operation region of the observer. It can also be determined by multiplying the value. One example of this can be expressed as:

Ω _min in the equation (28c) is introduced in order to guarantee the condition | g _m1 | + | g _m2 | ≠ 0 for stable convergence obtained from the equation (6b). The lower limit speed is shown. In the fourth aspect of the invention, the zero matrix element is captured as a value obtained by multiplying the zero constant coefficient by the absolute value of the speed equivalent value.

回転時の条件下では、誤差方程式の特性を支配する４ｘ４行列Ａ_ｒの固有値λは、次の４次方程式の根として与えられる。

（２９）式は、（２９ａ）式を満足する４個の全固有値は、高速回転時には、速度の絶対

The conditions at the time of rotation, the eigenvalue λ of 4x4 matrix A _r which govern the characteristic of the error equation is given as the root of the next fourth order equations.

In equation (29), all four eigenvalues satisfying equation (29a) are absolute values of speed during high-speed rotation.

  The fact that all eigenvalues change approximately in proportion to the speed means that the convergence speed of the state observer changes approximately according to the speed. Therefore, according to the fourth aspect of the invention, there is an effect that the convergence speed of the state observer can be varied in accordance with the speed. As a result, it is possible to stably estimate the state variable in a wide operation region from low speed to high speed, and as a result, an effect is obtained that a rotor phase estimation device having a wide operation range can be realized. Note also the difference between the gain exemplified in equation (28) and the heel gain shown in equation (1) and the Yamamoto gain shown in equations (4) and (5). Furthermore, it should be noted that the difference from the gain of Kanehara described in paragraphs 12 and 13, which requires solving the Riccati equation for the purpose of averaging a certain frequency characteristic over the entire drive region. Unlike the present invention, the gain of Kanehara cannot specify the convergence speed.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.

Embodiment 1

  FIG. 4 shows an embodiment in which a drive control device using the rotor phase estimation device of the present invention is applied to a synchronous motor. The main point of the present invention is in a rotor phase estimation device (hereinafter also referred to as a phase speed estimator), but in order to clarify the positioning of the rotor phase estimation device (phase speed estimator) in the entire motor drive control system, The entire motor drive control system including the drive control device will be described. 1 is a synchronous motor, 2 is a power converter, 3 is a current detector, 4a and 4b are 3 phase 2 phase converters, 2 phase 3 phase converters, 5a and 5b are both vector rotators, Reference numeral 6 denotes a current controller, 7 denotes a phase velocity estimator (rotor phase estimation device) using the present invention, and 8 denotes a cosine sine signal generator. As will be readily understood by those skilled in the art, in FIG. 4, various devices from 2 to 8 except for one electric motor constitute a drive control device. The phase speed estimator directly related to the present invention is also used in torque control and speed control. For the sake of simplicity, this figure shows a system for performing current control substantially equivalent to torque control. In this figure, in order to clarify the coordinate system of voltage and current, these signals are given a foot symbol t (uvw coordinate system), s (αβ fixed coordinate system), r (γδ quasi-synchronous coordinate system). Yes. Furthermore, in order to ensure simplicity, the 2 × 1 vector signal is represented by one thick signal line.

  Among the drive control devices, the phase speed estimator 7 is related to the present invention. FIG. 4 is particularly characterized in that the phase velocity estimator is configured on an αβ fixed coordinate system. FIG. 5 shows the internal structure of the phase velocity estimator. This phase speed estimator comprises a phase estimator 7a and a speed estimator 7b.

している。αβ固定座標系上の本位相推定器は、請求項１の発明に基づく（１５）式のγδ一般座標系上のＡ形状態オブザーバに従い構成することができる。具体的には、（１５）式のγδ一般座標系上のＡ形状態オブザーバに固定座標系の条件（ω_γ＝０，θ_γ＝θ_α）

doing. This phase estimator on the αβ fixed coordinate system can be configured according to the A-type state observer on the γδ general coordinate system of the equation (15) based on the invention of claim 1. Specifically, the condition of the fixed coordinate system (ω _γ = 0, θ _γ = θ _α ) is added to the A-type state observer on the γδ general coordinate system of equation (15).

とになる。FIG. 6 shows the detailed structure of the phase estimator (A-type state observer) configured on the αβ fixed coordinate system. In the figure, for the observer gains G _i and G _m , the fixed gain method according to the inventions of claims 2 and 3 is adopted. Of course, according to the inventions of claims 2 and 4, a gain method for changing the matrix observer gain according to the speed may be used. In this case

It becomes.

することにある。本速度推定器は、次の文献（１０）で使用されたものと同一の積分フィードバック形速度推定法に基づき構成されている。
新中新二：「永久磁石同期モータの最小次元Ｄ因子状態オブザーバとこれを用いたセンサレスベクトル制御法の提案」、電気学会論文誌Ｄ、１２３、１２、ｐｐ．１４４６−１４６０（２００３−１２）

There is to do. This speed estimator is configured based on the same integral feedback speed estimation method used in the following document (10).
Shinji Shinnaka: “Proposal of minimum dimension D-factor state observer of permanent magnet synchronous motor and sensorless vector control method using the same”, IEEJ Transactions D, 123, 12, pp. 11-27. 1446-1460 (2003-12)

  It should be pointed out that, instead of the speed estimator exemplified above, a speed estimator having a similar function may be configured based on an adaptive method or another method.

素を変更することになる。The phase estimator can also be configured using a B-type state observer. The main changes to obtain a practical B-type state observer on the αβ fixed coordinate system from the B-type state observer on the γδ general coordinate system of the equation (16) are the same as in the case of the A-type state observer. . Therefore, this description is omitted. FIG. 7 shows the detailed structure of the phase estimator (B-type state observer) configured on the αβ fixed coordinate system. In the figure, for the observer gains G _i and G _m , the fixed gain method according to the inventions of claims 2 and 3 is adopted. Of course, in accordance with the inventions of claims 2 and 4, a gain method for changing the matrix observer gain according to the speed may be used.

It will change the element.

  Even when the phase estimator is configured using a B-type state observer, the same speed estimator as in the case of the A-type state observer can be used. The versatility of the speed estimator is the same when it is based on an adaptive speed estimation method. For this reason, description regarding this is omitted.

Embodiment 2

  In particular, exemplary embodiments relating to claims 2 and 3 will be described. The design of the observer gain according to the present invention requires motor parameters. Table 1 summarizes the characteristics of the test synchronous motor.

本設計は，（２４）式において，α_ｉ＝４．４３，α_ｇ＝１の選定を意味する。According to the equations (23) and (24) based on the inventions of claims 2 and 3, the following values can be obtained as the observer gain.

This design means selection of α _i = 4.43 and α _g = 1 in the equation (24).

  A numerical experiment system having the same configuration as that shown in FIGS. 4 to 6 was prepared, and a numerical experiment was performed. In the experimental system, a load device was connected to the test synchronous motor so that the speed and torque of the test synchronous motor could be set independently. With the load device, the test synchronous motor was immediately rotated at a rated speed of 180 (rad / s) immediately after the start of driving. Simultaneously with the rotation of the test synchronous motor, the sensorless vector control system for the test synchronous motor was turned on and sensorless drive control was started. The current command value at this time was almost the rated value. The error of the phase estimation value with respect to the rotor phase true value at the start of driving is the maximum ± π (rad), and the speed estimation value is zero. Accordingly, the phase error immediately after the start of estimation is ± π (rad), and the speed error is −180 (rad / s).

位相真値から理解されるように，供試同期電動機は駆動直後から負荷装置により所期の速度に維持されている。また，駆動直後の大きな位相誤差，速度誤差にもかかわらず，駆動開始約０．１（ｓ）後には，位相推定値，速度推定値は，各々同真値に収斂し，この結果，固定子電流が適切に制御されている様子が確認される。同様な応答は，負回転時にも確認している。

As can be understood from the true phase value, the synchronous motor under test is maintained at the desired speed by the load device immediately after driving. Despite the large phase error and speed error immediately after driving, the phase estimated value and speed estimated value converge to the same true value after about 0.1 (s) of driving, and as a result, the stator It is confirmed that the current is properly controlled. Similar response has been confirmed during negative rotation.

Embodiment 3

本ゲインの特色は、固定ゲイン方式であると同時に、回転子磁束に対応したゲインの逆対角要素が著しい非対称になっている点にある。According to the equations (23) and (25) based on the inventions of claims 2 and 3, the following values can be obtained as the observer gain.

The feature of this gain is not only the fixed gain system but also the fact that the anti-diagonal element of the gain corresponding to the rotor magnetic flux is significantly asymmetric.

  Except for the gain design, the experiment was performed under the same conditions as in FIG. The experimental results are shown in FIG. The meanings of the waveforms in FIGS. 9A and 9B are the same as those in FIGS. 8A and 8B. FIG. 9 shows good estimation performance with no significant difference from FIG. Similar estimation performance was obtained in the case of negative rotation.

Embodiment 4

図８、９と同一の実験を行った結果、これらと有意の差のない安定した収束性能を得た。負回転の場合も同様な性能であった。According to the equations (23) and (26) based on the second and third aspects of the invention, the following value of the asymmetrical gain corresponding to the rotor magnetic flux can be obtained as the observer gain.

As a result of conducting the same experiment as FIGS. 8 and 9, stable convergence performance without significant difference was obtained. Similar performance was obtained in the case of negative rotation.

Embodiment 5

得ることができる。

また，（２８ｃ）式におけるω_ｍｉｎは，ω_ｍｉｎ＝１０とする。In particular, exemplary embodiments relating to claims 2 and 4 will be described. Based on the inventions of claims 2 and 4 (2

Can be obtained.

Also, ω _min in the equation (28c) is ω _min = 10.

The same experiment as the experiment of FIG. 8 was performed using the observer gains G _i and G _m using these parameters. However, the initial phase error was π / 4 (rad) and the initial velocity error was −180 (rad / s). The experimental results are shown in FIG. The meanings of the waveforms in FIGS. 9A and 9B are the same as those in FIGS. After 0.08 (s) of the estimation start, the phase / speed estimation is completed, and as a result, it is confirmed that the stator current is satisfactorily sensorlessly controlled. Similar sensorless control was confirmed in the case of negative rotation under similar conditions.

  The estimation examples illustrated in FIGS. 8 to 10 are cases in which the rotor phase estimation device is configured using FIGS. 4, 5 and 6. When the B-type state observer in FIG. 7 is used instead of the A-type state observer in FIG. 6, the same observer gain can be used, and as long as the same observer gain is used, the A-type and B-type states are used. It should be noted that there is an estimated response with no significant difference between observers.

Embodiment 6

  FIG. 11 shows another embodiment in which a drive control device using the rotor phase estimation device of the present invention is applied to a synchronous motor. The fundamental difference of the motor drive control system between FIG. 11 and FIG. 4 is the position of the rotor phase estimation device (phase speed estimator). In the embodiment of FIG. 4, the phase velocity estimator is configured on an αβ fixed coordinate system. In contrast, FIG. 11 shows an embodiment in which the phase velocity estimator is configured on a γδ quasi-synchronous coordinate system.

  In FIG. 11, the functions of the other devices except the phase velocity estimator are the same as those in FIG. The internal structure of the phase velocity estimator in FIG. 11 is shown in FIG. The phase deviation estimator is composed of a phase deviation estimator 7c and an in-phase device 7d.

差推定器は、請求項１の発明に基づく（１５）式のγδ一般座標系上のＡ形状態オブザーバに従い構成することができる。具体的には、（１５）式のγδ一般座標系上のＡ形

The phase deviation estimator 7c adds the coordinate system speed in addition to the actual stator current value and the stator voltage command value.

The difference estimator can be configured according to the A-type state observer on the γδ general coordinate system of the equation (15) based on the invention of claim 1. Specifically, form A on the γδ general coordinate system of equation (15)

ることになる。FIG. 13 shows the detailed structure of the phase deviation estimator (A-type state observer) configured on the γδ quasi-synchronous coordinate system. In the figure, for the observer gains G _i and G _m , the fixed gain method according to the inventions of claims 2 and 3 is adopted. Of course, according to the inventions of claims 2 and 4, a gain method for changing the matrix observer gain according to the speed may be used. This place

Will be.

用されたものと同一の一般化積分形ＰＬＬ法に基づき構成されている。

This is based on the same generalized integral PLL method used.

  It should be pointed out that instead of the phase synchronizer exemplified above, a phase synchronizer having a similar function may be configured based on an adaptive method or another method.

インＧ_ｉ，Ｇ_ｍの要素を変更することになる。The phase deviation estimator can also be configured using a B-type state observer. The main changes to obtain a practical B-type state observer on the γδ quasi-synchronous coordinate system from the B-type state observer on the γδ general coordinate system of equation (16) are the same as in the case of the A-type state observer. is there. Therefore, this description is omitted. FIG. 14 shows the detailed structure of the phase deviation estimator (B-type state observer) configured on the γδ quasi-synchronous coordinate system. In the figure, for the observer gains G _i and G _m , the fixed gain method according to the inventions of claims 2 and 3 is adopted. Of course, according to the inventions of claims 2 and 4, a gain method for changing the matrix observer gain according to the speed is used.

The elements of in G _i and G _m are changed.

  Even when the phase deviation estimator is configured using a B-type state observer, the same phase synchronizer as in the case of the A-type state observer can be used. The versatility of the phase synchronizer is the same when it is based on an adaptive method. For this reason, the description regarding this is abbreviate | omitted.

Embodiment 7

  The observer gain used for the phase deviation estimator (A-type, B-type state observer) configured on the γδ quasi-synchronous coordinate system is the phase estimator (A-type, B-type state observer) configured on the αβ fixed coordinate system. ) Is the same as the observer gain used. The present embodiment based on claims 2 to 4 has already been described in detail through FIGS.

  The present invention has been described specifically and in detail using a plurality of embodiments with reference to various drawings. The present invention described above can be variously modified and changed by those having ordinary knowledge in the technical field to which the present invention belongs without departing from the technical scope of the present invention. It should be pointed out that it is not limited to.

  The present invention can be widely used for applications in which sensorless driving is desired in a medium to high speed range.

  The figure which shows the example of 1 relationship of three types of coordinate systems and a rotor phase   The block diagram which shows the basic composition of the A form state observer on the gamma-delta general coordinate system based on this invention   Block diagram showing the basic configuration of a B-type state observer on the γδ general coordinate system   Block diagram showing an example of a drive control system using a phase velocity estimator based on the present invention   The block diagram which shows the internal structure example of the phase velocity estimator based on this invention   Block showing the basic configuration of an A-type state observer having two types of observer gains on the αβ fixed coordinate system according to the present invention   Block diagram showing the basic configuration of a B-type state observer with two types of observer gains on the αβ fixed coordinate system   Example of estimation response using phase velocity estimator and observer gain based on the present invention   Example of estimation response using phase velocity estimator and observer gain based on the present invention   Example of estimation response using phase velocity estimator and observer gain based on the present invention   Block diagram showing an example of a drive control system using a phase velocity estimator based on the present invention   The block diagram which shows the internal structure example of the phase velocity estimator based on this invention   Block showing the basic configuration of an A-type state observer having two types of observer gains on the γδ quasi-synchronous coordinate system according to the present invention   Block diagram showing the basic configuration of a B-type state observer with two types of observer gains on the γδ quasi-synchronous coordinate system

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 Synchronous motor 2 Power converter 3 Current detector 4a 3 phase 2 phase converter 4b 2 phase 3 phase converter 5a Vector rotator 5b Vector rotator 6 Current controller 7 Phase speed estimator 7a Phase estimator 7b Speed estimator 7c Phase deviation estimator 7d Phase synchronizer 8 Cosine sine signal generator

Claims (3)

Used in a drive control device for a synchronous motor having a permanent magnet in the rotor, and the equivalent value of the drive voltage and the equivalent value of the drive current are input signals, and the stator flux linkage and rotor flux are fixed. A rotor phase estimation device comprising at least a means for generating a rotor phase estimation value by configuring at least a state observer using any one of two kinds of combinations of a child reaction magnetic flux and a rotor magnetic flux as a state variable,
Regarding the two types of 2 × 2 matrix observer gains Gi and Gm used for the state observer, the matrix observer gain corresponding to the stator linkage magnetic flux or the stator reaction magnetic flux is Gi, and the matrix observer gain corresponding to the rotor magnetic flux is Gm. Then, the rotor phase estimation apparatus is characterized in that two types of matrix observer gains are set according to the following equations.

However, (delta) i12, (delta) i21: Arbitrary micro values which make zero a representative value, Ld: d-axis inductance, Lq: q-axis inductance,

: Equivalent value of electric velocity, sgn: sign function. 2. The rotor phase estimation apparatus according to claim 1 , wherein all elements of the two-type matrix observer gain are made constant regardless of the speed. The rotor phase estimation apparatus according to claim 1 , wherein the eigenvalue of the state observer changes substantially in proportion to the absolute velocity value in an intended operation region of the state observer. A rotor phase estimation apparatus characterized in that all elements are proportionally determined by multiplying a constant proportional coefficient corresponding to each element by an absolute value of a speed equivalent value.

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