JP5437174B2 - LED driving circuit and semiconductor device - Google Patents

Description

  The present application relates to an LED drive circuit and a semiconductor device for driving an LED.

  Conventionally, an LED (Light Emitting Diode) is driven with a constant current. And the electric power loss reduction is aimed at regarding the constant current drive of LED. For example, regarding constant current driving of an LED using a current control unit that includes a constant current source that draws a predetermined amount of current and adjusts the current amount by PWM (Pulse Width Modulation), the power in the current control unit Techniques for reducing loss are known.

JP2007-129862A

  For example, as shown in FIG. 14, constant current driving of the LED using a regulator and a current setting resistor is also performed. In this method, a voltage Vmine determined by the current setting resistor RLED connected in series to the LED 5 and the current ILED flowing through the LED 5 is fed back to the regulator 1. The regulator 1 adjusts the output voltage Vo applied to the LED 5 so that the fed back voltage Vmine is equal to the reference voltage Vref generated through the RC filter based on the pulse-shaped dimming signal. Thereby, LED5 can be driven with a constant current.

  In FIG. 14, in order to improve the current accuracy, the voltage Vmine of the current setting resistor RLED fed back to the regulator 1 is increased relative to the input offset voltage of the amplifier in the regulator 1 to make the influence of the offset relatively. It is possible to make it smaller. Since the regulator 1 works to satisfy Vmine = Vref, the reference voltage Vref is increased in order to increase the voltage Vmine. Here, in FIG. 14, assuming that the forward voltage drop per LED 5 connected in series is VF, Vo = VF × (number of LEDs 5) + Vref, and the output voltage Vo depends on the reference voltage Vref. Therefore, when the reference voltage Vref is increased, the output voltage Vo is increased. On the other hand, since the current ILED flowing through the LED 5 is constant, it is necessary to increase the current setting resistance RLED. However, when the current setting resistor RLED is increased, there is a problem that power loss due to the current setting resistor RLED increases. The above-mentioned Patent Document 1 does not touch on the reduction of power loss from the viewpoint of such current accuracy.

  The present invention has been proposed in view of the above problems, and an object of the present invention is to provide an LED driving circuit capable of reducing power loss while considering current accuracy, and a semiconductor device including the same.

  An LED drive circuit disclosed in the present application is an LED drive circuit that drives an LED with a constant current, a regulator that adjusts a voltage applied to the LED based on a reference voltage, and a current corresponding to the current flowing through the LED Is converted to a voltage by flowing through the current sense resistor and fed back to the regulator, and a protection circuit that stops the output of the regulator based on the output of the level converter. The semiconductor device disclosed in the present application includes the LED drive circuit.

  According to the disclosed LED driving circuit and semiconductor device, the output voltage of the regulator does not depend on the reference voltage by providing the level converter in the feedback loop that feeds back the current flowing through the LED to the regulator. Therefore, a more optimal output voltage that is not limited by the reference voltage can be set. Therefore, for example, without increasing the output voltage, the voltage on the input side of the regulator can be increased to reduce the influence of the offset, so that it is possible to reduce the power loss in consideration of the current accuracy.

It is a circuit block diagram of a 1st embodiment. It is a circuit block diagram of a 2nd embodiment. FIG. 1 is an equivalent circuit diagram 1 of a level converter. It is explanatory drawing of a soft start signal. It is a circuit block diagram which shows the specific example of 2nd Embodiment. It is a circuit block diagram of a 3rd embodiment. It is a circuit block diagram of a 4th embodiment. It is an operation | movement timing chart of 4th Embodiment. It is a circuit block diagram of a fifth embodiment. FIG. 3 is an equivalent circuit diagram 2 of the level converter. It is a circuit block diagram which shows the specific example of 5th Embodiment. It is a circuit block diagram of a 6th embodiment. It is a circuit block diagram which shows the specific example of 6th Embodiment. It is a circuit block diagram which shows a prior art example.

  FIG. 1 shows a circuit block diagram of the first embodiment. The LED drive circuit of the first embodiment includes a regulator 1, a level converter 2, and a protection circuit 3. The non-inverting input terminal of the regulator 1 receives a reference voltage Vref generated through an RC filter based on a pulsed dimming signal given from the outside. A voltage Vmine fed back from the output side is input to the inverting input terminal of the regulator 1. The regulator 1 adjusts the output voltage Vo so that the voltage Vmine becomes equal to the reference voltage Vref, and supplies it to the LED 5.

  The level converter 2 is equivalently expressed by a configuration including an amplifier and a constant current source. In the level converter 2, the voltage across the resistor R1 connected in series to the LED 5 is input to the amplifier and amplified. A current flows from the constant current source to the current sense resistor Rs according to the output of the amplifier. As a result, the level converter 2 passes a current corresponding to the current ILED flowing through the LED 5 to the current sense resistor Rs to convert it into the voltage Vmine and feeds it back to the regulator 1.

  The output of the level converter 2 is input to the protection circuit 3. The protection circuit 3 outputs an enable signal enb to the regulator 1 based on the output of the level converter 2, and controls permission and stop of the output of the regulator 1.

  With the above configuration, the LED 5 can be driven with a constant current ILED. Regarding the output voltage Vo, in the conventional example of FIG. 14, Vo = VF × (number of LEDs 5) + Vref. On the other hand, in the first embodiment, if the resistance R1 is sufficiently small, Vo≈VF × (the number of LEDs 5). Thus, by providing the level converter 2 in the path for feeding back the current ILED flowing through the LED 5 to the regulator 1, the output voltage Vo of the regulator 1 does not depend on the reference voltage Vref. Therefore, a more optimal output voltage Vo can be set without being restricted by the reference voltage Vref.

  Focusing on the voltage Vmine fed back to the regulator 1, in the conventional example of FIG. 14, the voltage Vmine is determined by the current setting resistor RLED connected in series to the LED 5 and the current ILED flowing through the LED 5. On the other hand, in the first embodiment, the voltage Vmine is determined by the current sense resistor Rs at the output stage of the level converter 2 and the current flowing from the constant current source of the level converter 2. Therefore, in the first embodiment, in order to increase the voltage Vmine from the viewpoint of current accuracy, it is only necessary to increase the current sense resistor Rs, and it is not necessary to increase the resistor R1 connected in series to the LED 5. Therefore, the resistance R1 can be set small, and power loss can be reduced. Further, since the protection circuit 3 stops the output of the regulator 1 based on the output of the level converter 2, damage or the like can be prevented.

  FIG. 2 shows a circuit block diagram of the second embodiment. The LED drive circuit according to the second embodiment includes a regulator 1, a level converter 2, a protection circuit 3, and a soft start control circuit 4. A reference voltage Vref generated based on the dimming signal and a soft start signal output from the soft start control circuit 4 are input to the two non-inverting input terminals of the regulator 1, respectively. A voltage Vmine fed back from the output side is input to the inverting input terminal of the regulator 1. The regulator 1 adjusts the output voltage Vo so that the voltage Vmine is equal to the lower one of the reference voltage Vref and the soft start signal, and supplies the output voltage Vo to the LED 5.

  The level converter 2 will be described with reference to the equivalent circuit diagram of FIG. In the second embodiment, an n-channel MOS transistor 6 is connected to the LED 5 in series. N channel MOS transistor 21 has a gate connected to n channel MOS transistor 6 in common. The n-channel MOS transistors 6 and 21 are both turned on and off based on an external control signal CTL. The drain of the n-channel MOS transistor 6 is connected to the inverting input terminal of the amplifier 22. The drain of the n-channel MOS transistor 21 is connected to the non-inverting input terminal of the amplifier 22 and is connected to the power supply via the p-channel MOS transistor 24. Here, the size ratio of n channel MOS transistor 21 is set so that current flows at a ratio of 1: 100 as compared with n channel MOS transistor 6. Therefore, for example, when the current ILED flowing through the LED 5 is 30 mA, the n-channel MOS transistor 21 passes a current of 0.3 mA. Thereby, the power loss in the level converter 2 can be suppressed. The output of the amplifier 22 is connected to the gates of the p-channel MOS transistors 23 and 24. The drain of p channel MOS transistor 23 is connected to current sense resistor Rs. As a result, the level converter 2 passes the current Is corresponding to the current ILED flowing through the LED 5 to the current sense resistor Rs, converts it to the voltage Vmine, and feeds it back to the regulator 1.

  Referring to FIG. 2 again, the description of the second embodiment is continued. The protection circuit 3 includes two comparators 31 and 32. The output of the level converter 2 is input to the non-inverting input terminal of the comparator 31, and a threshold voltage (for example, 10 mV) is input to the inverting input terminal. When the LED 5 is dropped, the current ILED does not flow, so the output voltage Vmine of the level converter 2 decreases to near 0V. Therefore, the output of the comparator 31 changes from H level to L level. Thereby, the drop-off of the LED 5 can be detected. On the other hand, a threshold voltage (eg, 1.1 V) is input to the non-inverting input terminal of the comparator 32, and the output of the level converter 2 is input to the inverting input terminal. When an overcurrent flows through the LED 5, the current ILED becomes larger than the set value, so that the output voltage Vmine of the level converter 2 rises abnormally. Therefore, the output of the comparator 32 changes from H level to L level. Thereby, an overcurrent can be detected.

  The output of each comparator is input to the AND gate 33 together with the control signal CTL. The output of the AND gate 33 is output to the regulator 1 as the enable signal enb. In the present embodiment, the regulator 1 stops output when the enable signal enb is at L level. When the enable signal enb is at L level, the gate of the n-channel MOS transistor 6 is kept at L level and the n-channel MOS transistor 6 is turned off. As a result, the protection circuit 3 stops the output of the regulator 1 and shuts off the current ILED flowing through the LED 5 not only when the control signal CTL is turned off due to the L level input but also when the dropout of the LED 5 is detected or when an overcurrent is detected. .

  The soft start control circuit 4 outputs a soft start signal to the regulator 1. Here, the soft start signal is a signal that gradually rises from 0 V to the power supply voltage level as shown in FIG. As described above, the regulator 1 adjusts the output voltage Vo so that the voltage Vmine is equal to the lower one of the reference voltage Vref and the soft start signal. Therefore, the regulator 1 adjusts the output voltage Vo so that the voltage Vmine becomes equal to the voltage level of the soft start signal until the soft start signal reaches the reference voltage Vref after the regulator 1 starts output. When the soft start signal becomes equal to or higher than the reference voltage Vref, the regulator 1 adjusts the output voltage Vo so that the voltage Vmine becomes equal to the reference voltage Vref. Thereby, since the output voltage Vo can be gradually raised when the output of the regulator 1 is started, the inrush current can be suppressed and the stress on the LED 5 can be reduced.

  The soft start control circuit 4 outputs a soft start completion signal to the comparator 31 of the protection circuit 3 and masks the output of the comparator 31 until the output voltage Vmine of the level converter 2 reaches the threshold voltage of the comparator 31. It is a configuration. As a result, when the output of the regulator 1 starts, when the output voltage Vmine of the level converter 2 rises as the output voltage Vo gradually rises from 0 V, the comparator 31 outputs an L level regardless of the dropout of the LED 5 Can be prevented.

  With the above configuration, the LED 5 can be driven with a constant current ILED. The output voltage Vo is Vo = VF × (the number of LEDs 5), and the output voltage Vo of the regulator 1 does not depend on the reference voltage Vref in the second embodiment. Therefore, a more optimal output voltage Vo can be set without being restricted by the reference voltage Vref. Also in the second embodiment, the voltage Vmine fed back to the regulator 1 can be increased to improve the stability and reduce the power loss. In the first embodiment described above (see FIG. 1), the resistor R1 is connected to the LED 5 in series, whereas in the second embodiment, the n-channel MOS transistor 6 is connected to the LED 5 in series. The Thereby, it is possible to suppress the dark current from flowing when the output of the regulator 1 is stopped. Further, since the protection circuit 3 detects the dropout or overcurrent of the LED 5 and stops the output of the regulator 1, damage or the like can be prevented.

  FIG. 5 shows a specific example in which the semiconductor device 8 is configured by applying a current mode control step-up switching regulator as the regulator 1 in the second embodiment. The error amplifier 11 amplifies the difference between the lower one of the reference voltage Vref and the soft start signal and the voltage Vmine fed back from the output side. The output of the error amplifier 11 is input to the non-inverting input terminal of the comparator 12. The current flowing through the inductor L is input to the inverting input terminal of the comparator 12 via the current-voltage conversion circuit 15 and the slope compensation circuit 16 and compared with the output of the error amplifier 11. The output of the comparator 12 is input to the PWM control circuit 13 to control the on-duty, and a drive signal is output from the PWM control circuit 13 via the driver 14. The drive signal is input to the gates of the n-channel MOS transistors Tr1 and Tr2 in the current-voltage conversion circuit 15. The drain of n channel MOS transistor Tr1 is connected to inductor L, and the drain of n channel MOS transistor Tr2 is connected to resistor R. As a result, the current-voltage conversion circuit 15 converts a current corresponding to the current flowing through the inductor L to the resistor R and converts it into a voltage. The slope compensation circuit 16 is a circuit that prevents oscillation that occurs when the on-duty is 50% or more.

  When the n-channel MOS transistor Tr1 is turned on, energy from the input voltage VIN is accumulated in the inductor L. When the n-channel MOS transistor Tr1 is turned off, the energy stored in the inductor L is released, and the current flows to the capacitor C and the LED 5 via the diode D. When the n-channel MOS transistor Tr1 is turned on again, energy is stored in the inductor L again. At this time, a current flows through the LED 5 due to the energy accumulated in the capacitor C, and the diode D prevents backflow. With the above configuration, the switching regulator performs on / off control of the n-channel MOS transistor Tr1 according to the state of the voltage Vmine fed back from the output side, and supplies the output voltage Vo higher than the input voltage VIN to the LED 5.

  In this specific example, the current sense resistor Rs is not built in the level converter 2 but is externally attached to the semiconductor device 8. In addition, when the enable signal enb output from the protection circuit 3 is at L level, the driver 14 is stopped and the output of the switching regulator is stopped. Others are as described with reference to FIGS.

  With the above configuration, the semiconductor device 8 can drive the LED 5 with a constant current ILED. In a step-up type switching regulator, the efficiency deteriorates as the step-up ratio increases, so it is better to suppress the step-up ratio as much as possible. In this specific example, the output voltage Vo = VF × (the number of LEDs 5), and the output voltage Vo can be set without being restricted by the reference voltage Vref. Therefore, by boosting to the minimum necessary output voltage Vo and suppressing the boost ratio, it is possible to prevent deterioration in efficiency and reduce the loss. Also, by providing the current sense resistor Rs externally, it is possible to suppress variations in resistance value (for example, ± 20% for the built-in resistor but ± 0.5% for the externally mounted resistor). Thus, the current accuracy can be improved.

  FIG. 6 shows a circuit block diagram of the third embodiment. Comparing the third embodiment and the second embodiment (see FIG. 2), in the third embodiment, the reference voltage Vref input to the non-inverting input terminal of the regulator 1 is based on a dimming signal given from the outside. It is not generated, but is set to be variable. Further, a PWM signal is input instead of the control signal CTL. Others are the same as in the second embodiment.

  In the second embodiment described above (see FIG. 2), the drive control of the LED 5 is performed by the DC control signal CTL. However, the drive control of the LED 5 can also be performed by the PWM signal as in the third embodiment. . The same effects as those of the second embodiment can be obtained by the configuration of the third embodiment.

  FIG. 7 shows a circuit block diagram of the fourth embodiment. The fourth embodiment is a form obtained by developing the third embodiment. In the fourth embodiment, an inverter 34 and an OR gate 35 are added to the protection circuit 3. A PWM signal is input to the inverter 34, and an output of the AND gate 33 and an output of the inverter 34 are input to the OR gate 35. The output of the OR gate 35 is output as the enable signal enb.

  In the fourth embodiment, the sample hold circuit 7 is added to the path through which the output of the level converter 2 is input to the regulator 1. The sample hold circuit 7 is switched on when the PWM signal is at the H level, and is switched off when the PWM signal is at the L level. The period during which the PWM signal is at the L level is the level in the immediately preceding H level period. The value of the output voltage Vsense of the converter 2 is held and output.

  The operation of the fourth embodiment will be described with reference to the operation timing chart of FIG. As shown in FIG. 8, current ILED flows when the PWM signal is at the H level. Therefore, dimming can be performed by adjusting the average value of the current ILED flowing through the LED 5 according to the duty ratio of the PWM signal. During the period in which the PWM signal is at the L level, the value of the output voltage Vsense of the level converter 2 in the immediately preceding H level period is held by the sample hold circuit 7 described above. Therefore, the voltage Vmine fed back to the regulator 1 is kept stable, and the output voltage Vo can be prevented from becoming unstable. With the above configuration, the protection circuit 3 does not stop the output of the regulator 1 by masking the protection function when the PWM signal is turned off by the L level input. Therefore, the output voltage Vo is maintained. Further, when the drop of the LED 5 or the overcurrent is detected when the PWM signal is turned on by the H level input, the protection circuit 3 stops the output of the regulator 1 as indicated by the timing T in FIG. The current ILED flowing through is interrupted. Therefore, damage etc. can be prevented. The effect similar to 2nd Embodiment is acquired also by the structure of 4th Embodiment.

  FIG. 9 shows a circuit block diagram of the fifth embodiment. In the fifth embodiment, as compared with the second embodiment (see FIG. 2), a p-channel MOS transistor 41 is connected in series with the LED 5 instead of the n-channel MOS transistor 6. In the p-channel MOS transistor 41, the output voltage Vo of the regulator 1 is applied to the source, and the drain is connected to the LED 5. Control signal CTL is input to the gate of p-channel MOS transistor 41 through inverter 42.

  A level converter 2 corresponding to the fifth embodiment will be described with reference to an equivalent circuit diagram of FIG. The gate of p-channel MOS transistor 41 and the gate of p-channel MOS transistor 25 are connected in common to output A of inverter 42 (see FIG. 9). Both p-channel MOS transistors 41 and 25 are on / off controlled based on a control signal CTL input via an inverter 42. Similarly to the p-channel MOS transistor 41, the output voltage Vo of the regulator 1 is applied to the source of the p-channel MOS transistor 25. Here, the size ratio of p channel MOS transistor 25 is set so that current flows at a ratio of 1: 100 as compared with p channel MOS transistor 41. Therefore, for example, when the current ILED flowing through the LED 5 is 30 mA, the p-channel MOS transistor 25 passes a current of 0.3 mA. Thereby, the power loss in the level converter 2 can be suppressed. The drain of p channel MOS transistor 25 is connected to current sense resistor Rs. As a result, the level converter 2 passes the current Is corresponding to the current ILED flowing through the LED 5 to the current sense resistor Rs, converts it to the voltage Vmine, and feeds it back to the regulator 1.

  Since the other points are the same as those of the second embodiment, the same reference numerals are given to the respective parts corresponding to those in FIG. The same effects as those of the second embodiment can be obtained by the configuration of the fifth embodiment in which the p-channel MOS transistor 41 is connected in series to the LED 5. Further, comparing the level converter 2 described in FIG. 3 in the second embodiment with the level converter 2 in FIG. 10, the circuit configuration of the level converter 2 is simplified in the fifth embodiment, and the circuit scale can be suppressed. .

  FIG. 11 shows a specific example in which a step-up DCDC converter is applied as the regulator 1 in the fifth embodiment. An input voltage VIN supplied from a rechargeable battery such as a lithium ion battery is applied to one end of the inductor L. The other end of the inductor L is grounded via an n-channel MOS transistor Tr1. The anode of the diode D is connected to the connection point between the inductor L and the n-channel MOS transistor Tr1. Capacitor C that smoothes output voltage Vo of regulator 1 is connected to the cathode of diode D via p-channel MOS transistor 41.

  The DCDC converter controller 10 is realized, for example, by a configuration including the error amplifier 11, the comparator 12, the PWM control circuit 13, the driver 14, the current-voltage conversion circuit 15, and the slope compensation circuit 16 described in FIG. Since the operation of the step-up DCDC converter has been described with reference to FIG. 5, detailed description thereof is omitted here. The DCDC converter controller 10 controls on / off of the n-channel MOS transistor Tr1 according to the state of the voltage Vmine fed back from the output side, and generates an output voltage Vo higher than the input voltage VIN.

  Next, advantages of this example will be described. In general, portable devices drive an RTC (Real Time Clock) or the like with a built-in backup button battery. However, when a battery that supplies an input voltage VIN is inserted, a battery with a large supply power is driven by the button battery. Switch to drive by. At that time (that is, when a battery is inserted into the device), a rush current for charging the capacitor C flows from the battery via the inductor L and the diode D. When a voltage drop occurs in the input voltage VIN due to the rush current and the internal resistance of the battery, the microcomputer that holds the settings of the RTC and the portable device may be reset. Therefore, there is a need to prevent rush current when the battery is inserted. However, in the case of the specific example of the second embodiment described with reference to FIG. 5, the path from the input voltage VIN to the capacitor C through the inductor L and the diode D is formed even when the regulator is stopped. Can not be prevented. On the other hand, in the case of the specific example of the fifth embodiment shown in FIG. 11, the path from the input voltage VIN to the capacitor C is blocked by the p-channel MOS transistor 41 that is off when the regulator is stopped. The above rush current can be prevented.

  Further, as described above, the comparator 32 of the protection circuit 3 detects an overcurrent of the current ILED flowing through the LED 5. However, in the case of the specific example of the second embodiment described with reference to FIG. 5, it is not possible to detect an overcurrent that flows when the front stage of the LED 5 is shorted. On the other hand, in the case of the specific example of the fifth embodiment shown in FIG. 11, since the current is detected at the front stage of the LED 5, it is possible to detect the overcurrent that flows when the front stage of the LED 5 is short-circuited.

  Up to this point, the fifth embodiment in which the n-channel MOS transistor 6 is changed to the p-channel MOS transistor 41 with respect to the second embodiment has been described. FIG. 12 shows a circuit block diagram of a sixth embodiment in which the n-channel MOS transistor 6 is changed to a p-channel MOS transistor 41 with respect to the third embodiment (see FIG. 6). FIG. 13 shows a specific example in which a step-up DCDC converter is applied as the regulator 1 in the sixth embodiment of FIG. In FIG. 12 and FIG. 13, the same reference numerals are given to the respective parts corresponding to those described so far, and the description thereof is omitted. The sixth embodiment in which the n-channel MOS transistor 6 is changed to the p-channel MOS transistor 41 with respect to the third embodiment and its specific example are also described in the fifth embodiment and its specific example (FIGS. 9 to 11). There are benefits similar to benefits.

  Here, the comparators 31 and 32 included in the protection circuit 3 are examples of a first comparator and a second comparator, respectively. N-channel MOS transistors 6 and 21 are examples of a first n-channel MOS transistor and a second n-channel MOS transistor, respectively. The p channel MOS transistors 41 and 25 are examples of a first p channel MOS transistor and a second p channel MOS transistor, respectively. The n-channel MOS transistor Tr1 is an example of a switching element. The diode D is an example of a rectifying element. The control signal CTL and the PWM signal are examples of control signals, respectively.

  As described above in detail, according to the first to sixth embodiments, the output voltage Vo of the regulator 1 is provided by providing the level converter 2 in the path for feeding back the current ILED flowing through the LED 5 to the regulator 1. Is independent of the reference voltage Vref. Therefore, it is possible to set a more optimal output voltage Vo that is not limited by the reference voltage Vref. Therefore, the voltage Vmine fed back to the regulator 1 can be increased to increase the stability, and the power loss can be reduced. It is possible to reduce power loss in consideration of current accuracy.

  Needless to say, the present invention is not limited to the above-described embodiment, and various improvements and modifications can be made without departing from the spirit of the present invention.

  For example, the regulator 1 is not limited to a step-up switching regulator, and may be a step-down or step-up / step-down switching regulator. Even in this case, the loss can be reduced.

1 regulator 2 level converter 3 protection circuit 4 soft start control circuit 5 LED
6 n-channel MOS transistor 7 sample hold circuit 8 semiconductor device 31, 32 comparator 41 p-channel MOS transistor Rs current sense resistor

Claims (8)

In an LED drive circuit for driving an LED with a constant current,
A regulator for adjusting a voltage applied to the LED based on a reference voltage;
A level converter that converts a current corresponding to a current flowing through the LED into a voltage by flowing the current through a current sense resistor, and feeds back to the regulator;
A protection circuit for stopping the output of the regulator based on the output of the level converter;
An LED driving circuit comprising: The protection circuit is
A first comparator for comparing the output of the level converter with a first threshold voltage to detect the dropout of the LED;
A second comparator for comparing the output of the level converter and a second threshold voltage to detect an overcurrent;
With
The LED drive circuit according to claim 1, wherein when the dropout or overcurrent of the LED is detected, the output of the regulator is stopped. The soft start control circuit for masking the output of the first comparator until the output of the level converter reaches the first threshold voltage at the start of output of the regulator. The LED driving circuit described. A first n-channel MOS transistor connected in series to the LED and controlled to be turned on and off based on a control signal;
The level converter is
A second n-channel MOS transistor having a gate connected in common with the first n-channel MOS transistor;
An amplifier in which a drain of the first n-channel MOS transistor and a drain of the second n-channel MOS transistor are respectively connected to input terminals;
A p-channel MOS transistor having a gate connected to the output of the amplifier and a drain connected to the current sense resistor;
The LED drive circuit according to claim 1, further comprising: A first p-channel MOS transistor that is on / off controlled based on a control signal;
The regulator is
An inductor with an input voltage applied to one end;
A switching element connected to the other end of the inductor;
A rectifying element having one end connected to a connection point between the inductor and the switching element;
A capacitor connected to the other end of the rectifying element via the first p-channel MOS transistor and smoothing the output voltage;
The LED driving circuit according to claim 1, wherein the LED driving circuit is a step-up DCDC converter. The level converter is
6. The LED drive according to claim 5, further comprising: a second p-channel MOS transistor having a gate and a source connected in common with the first p-channel MOS transistor and a drain connected to the current sense resistor. circuit.   A semiconductor device comprising the LED drive circuit according to claim 1.   The semiconductor device according to claim 7, wherein the current sense resistor is externally attached.

Images