JP5962703B2 - Load drive device - Google Patents

Description

  The present invention relates to a load driving device that drives a load such as an LED (light emitting diode) with a constant current, and more particularly, to a load driving device that reduces the loss of a constant current control element in a configuration in which a plurality of loads are connected in parallel.

  In Patent Document 1, as the prior art, the voltage across the constant current source inserted and connected in series with the LED string is detected, and the voltage across the voltage and the reference voltage from the reference voltage source are compared by an error amplifier. A configuration for controlling the output voltage of the converter is disclosed. In this case, since the product of the set current flowing through the constant current source and the voltage across the constant current source becomes a loss in the constant current source, the efficiency is lowered particularly at light loads. Therefore, Patent Document 1 proposes a method of changing the reference voltage according to the load current.

  Patent Document 2 discloses an LED drive device in which a plurality of LED strings connected in series on the output side of a converter are connected in parallel. In this case, each LED string is inserted and connected in series with the LED string. The voltage across the constant current source is detected, the voltage across the reference voltage source and the reference voltage from the reference voltage source are compared with an error amplifier (gm amplifier), the current output from the gm amplifier is converted into a voltage, and the output of the converter The voltage is controlled.

JP 2010-63332 A Japanese Patent No. 5203320

  As a problem of the above-mentioned background art, the constant current sources disclosed in Patent Documents 1 and 2 are used in such a configuration that a switching element such as an FET (field effect transistor) is connected in series with the LED string. The switching element is operated in the saturation region (active region in the bipolar transistor) without operating in the switching region in order to keep the current flowing in the LED string constant. In order to pass a current for operating a switching element, for example, an FET in a saturation region, it is necessary to apply a voltage of a certain value or more between both ends of the switching element (drain-source for FET and collector-emitter for bipolar transistor).

  However, since the voltage across the switch element required to operate in the saturation region fluctuates due to temperature changes, component variations, etc., the fluctuation margin is set to the minimum voltage required for saturation across the switch element. It is necessary to add it in advance. Therefore, it is necessary to add a margin in advance to a reference voltage for operating the converter at a constant voltage. For this reason, the voltage across the constant current source cannot be brought close to 0, and the loss corresponding to the product of the voltage across the switch element and the set current has become a problem as a load driving device. This means that when a plurality of LED drive devices are connected to a converter operating at a constant voltage, the LED row having the maximum sum of the forward drop voltages Vf of the LEDs in each LED drive device is at both ends of the switch element. This means that even if the LED string is used, the loss occurs at the constant current source.

  Therefore, the present invention provides a case where one or a plurality of LED driving devices including a series of LEDs are used in parallel, or when the number of serial LEDs or the number of parallel LED driving devices increases or decreases. If the number of LEDs in series is the same, it is possible to minimize the loss of the constant current source even when the maximum value of the sum of the forward drop voltages Vf of the LEDs in each LED driving device changes. An object of the present invention is to provide a control method for a load driving device.

A load driving device according to the present invention includes a power conversion circuit that supplies power to one or a plurality of parallel loads, and a switch element that is connected in series with the load of each row and controls a current flowing through the connected load. A constant current source having a built-in voltage, a voltage obtained by converting a current flowing through the load, a voltage at a control terminal of the switch element, or a voltage at a positive terminal of the switch element through which a current flows from the load. Select and measure each time in the process of increasing the output voltage applied to each column from the power conversion circuit in a stepwise manner with a value arbitrarily specified in advance for each predetermined period, Statistical processing for calculating the sum value, average value, minimum value, maximum value, etc. between the columns, and using any one of them as a monitor value makes the monitor value steep with respect to changes in the output voltage. Change changing inflection point By detecting as configured by and a control circuit for performing control for increasing or decreasing the output voltage as a target value of the change point.

  In this case, when the output voltage is increased, the change point is changed from the non-constant current state in which the load current changes following the change in the output voltage, and the change in the load current is eliminated with respect to the change in the voltage. It is preferable to detect a point at which the state changes to a constant current state, that is, a value when the monitor value changes from a variable value to a constant value.

  This is because this state is a point at which the operation is shifted to the constant current operation, the voltage across the switch element is the smallest, and the loss of the constant current source is the smallest.

Further, it is preferable that the control circuit is configured so that the output voltage is monotonously changed, that is, increased until the monitor value reaches a threshold value at which a constant current flows through all the switch elements.

  According to the load driving device of the present invention, when the output voltage from the power conversion circuit to the load is changed using the monitor value statistically processed by the control circuit, the monitor value before and after the change can be compared. For example, it is possible to detect a change point at which the load of each column can operate at a constant current by the constant current source regardless of the load of the column. Therefore, if the output voltage is controlled with the change point detected by the control circuit as the target value of the monitor value, the switch element may have temperature changes or component variations, or the output current setting value or load may fluctuate. Even so, the voltage across the switch element in the constant current source can be minimized, so that the loss of the switch element can be minimized. In addition, even if the load power and the number of columns increase, the monitor value can be generated in the same way, and the output voltage can be controlled so that the loss of the switching element of the constant current source is minimized. No reference value is required to compare values, so control can be performed reliably regardless of variations in switch element parts, and the same circuit configuration can be used even if the switch element is changed or the output current changes. it can.

  Preferably, the optimum change point can be detected by gradually increasing the output voltage from a relatively low voltage at the start of the operation of the power conversion circuit.

Preferably, when the output voltage is increased , the monitor value change point is not changed until the monitor value reaches the threshold value, so that the monitor value change point is not erroneously detected at another stage due to the influence of noise or the like. The output voltage is increased monotonically without detecting. Thereby, the change point of the monitor value can be correctly detected by the control circuit.

1 is a circuit diagram of a load driving device according to a first embodiment of the present invention. It is a graph which shows the relationship between the output voltage of a converter, and the voltage between the both ends of the resistance corresponding to the electric current value of each LED row | line | column same as the above. It is a graph which shows the relationship between the output voltage of a converter, and the monitor value obtained from the voltage between both ends of resistance same as the above. 3 is a flowchart showing a periodic operation procedure of the output voltage control means. It is a circuit diagram of the load drive device concerning a second embodiment of the present invention.

  A load driving device according to the present invention will be described with reference to the accompanying drawings.

  FIG. 1 shows a load driving device 1 according to a first embodiment of the present invention. The load driving device 1 includes a converter 3 corresponding to a power conversion circuit, a constant current source 4 including a switch element Q1 for current control, an LED array, in order to drive and control an LED array 2 in which a plurality of LEDs are connected in series. 2 and a microcomputer (microcomputer) 6 as a control circuit.

  The converter 3 converts the DC input voltage Vin applied between the input terminals + Vi and −Vi to the DC output voltage Vout and supplies it to the LED array 2. Here, the output voltage Vout is higher than the input voltage Vin. In order to perform conversion, a boost chopper circuit is configured by a choke coil Lcnv, a switching element Qcnv such as an FET, a diode Dcnv, and a capacitor Ccnv. More specifically, a series circuit of a choke coil Lcnv and a switch element Qcnv is connected between both ends of the input terminals + Vi and −Vi, and a series circuit of a diode Dcnv and a capacitor Ccnv is connected between both ends of the switch element Qcnv. In connection with the switching operation of the switch element Qcnv, the output voltage Vout is generated across the capacitor Ccnv.

  Further, a gate driver 8 that is a drive circuit and a PWM unit 16 in the microcomputer 6 that generates a PWM (pulse width modulation) pulse are sequentially connected to a control terminal (gate) of a switch element Qcnv that is a main switch of the converter 3. Is done. Thus, upon receiving a PWM pulse from the PWM unit 16 corresponding to the pulse generation circuit, the gate driver 8 supplies a pulse drive signal enabling the switching operation of the switch element Qcnv to the gate of the switch element Qcnv, and the PWM pulse The output voltage Vout from the converter 3 can be variably controlled by changing the on-duty.

  As an example, a load to be driven by the load driving device 1 includes one or a plurality of LED rows 2. In this embodiment, for convenience of explanation, suffixes -1, -2,.

  Hereinafter, focusing on one LED string 2-1, between the two ends of the capacitor Ccnv that generates the output voltage Vout, the LED string 2-1, the switch element Q1-1, and the current detection resistor R1-1 Is connected to operate the switch element Q1-1 so that the LED string 2-1 is driven at a constant current. The switch element Q1-1 of the constant current source 4 and the switch element Qcnv of the converter 3 described above are both N-channel MOS FETs, but even if another semiconductor element with a control terminal such as a bipolar transistor is used. Good. The resistor R1-1 converts the current value Iout-1 flowing through the LED array 2-1 into the voltage S1-1. As an alternative current detector, a current transformer or a Hall element may be used. Good.

  The constant current source 4 includes an operational amplifier A1-1, a reference voltage source 9 common to each LED array 2, and a resistor R2-1, in addition to the switch element Q1-1 and the resistor R1-1 described above. . The operational amplifier A1-1 is a voltage obtained by converting a current value Iout-1 flowing through the LED string 2-1 generated between both ends of the resistor R1-1, and is a source voltage S1-1 of the switch element Q1-1 and ADIM. The reference voltage V1 from the reference voltage source 9 that can be changed by the terminal is error-amplified, and the error-amplified output is used as the gate voltage S2-1 of the switch element Q1-1 and the switch element via the resistor R2-1. Q1-1 is given to the gate. Thus, the constant current source 4 detects the current value Iout-1 flowing through the LED array 2-1 as the source voltage S1-1 of the switch element Q1-1, and the voltage S1-1 and the reference voltage V1 are detected as the operational amplifier A1-. 1, the drain-source voltage corresponding to the voltage between the + side terminal and the − side terminal of the switch element Q <b> 1-1 is changed based on the result of error amplification at 1, and a constant current is supplied to the LED array 2-1. Perform the flow operation. This operation is performed independently of the variable control of the output voltage Vout of the converter 3.

  The dimming circuit 5 includes a reference voltage source 9 that also functions as a constant current source 4 and an analog dimming circuit using an ADIM terminal, as well as a switch element Q2-1 made of, for example, an NPN bipolar transistor, and each LED A PWM dimming circuit including a resistor R3 common to the column 2 and a PDIM terminal is combined. The analog dimming circuit varies the reference voltage V1 from the reference voltage source 9 using the ADIM terminal, thereby determining the maximum value of the current value Iout-1 flowing in the LED string 2-1, and the LED string 2- 1 is dimming. On the other hand, in the PWM dimming circuit, a resistor R3 is inserted and connected between the PDIM terminal and the base of the switch element Q2-1, the collector of the switch element Q2-1 is connected to the gate of the switch element Q1-1, and the switch element While the emitter of Q2-1 is grounded and the H (high) level voltage is output from the PDIM terminal, the switch element Q2-1 of the constant current source 4 is turned on as the switch element Q2-1 is turned on. It has a configuration for turning it off. Thus, when a low frequency dimming pulse signal of about 100 Hz to 1 kHz is given to the PDIM terminal, the LED row 2-1 operates intermittently at that frequency, and the on-duty of the dimming pulse signal is varied, whereby the LED row 2 -1 can be dimmed. The dimming pulse signal here is set to a frequency that is so high that the flickering of light in the LED array 2-1 is not noticeable and lower than the PWM pulse from the PWM unit 16.

  The configuration of the constant current source 4 and the dimming circuit 5 related to one LED string 2-1 has been described above, but the same can be said for the other LED strings 2-2,.

  The microcomputer 6 calculates a control command value for controlling the operation of the converter 3 by digital calculation, and includes a MUX 11, an ADC 12, an I / O unit 14, a CPU 15, and a PWM unit 16.

  The MUX 11 receives the selection control signal from the CPU 15 and outputs the value of the output voltage Vout from the converter 3 and the source voltages S1-1, S1-2, and the switching elements Q1-1, Q1-2,. ..., the value of S1-n, the values of the gate voltages S2-1, S2-2, ..., S2-n of the switching elements Q1-1, Q1-2, ..., Q1-n, and the switching elements Q1-1, This corresponds to a multiplexer that sequentially outputs any one of the values of the drain voltages S3-1, S3-2,..., S3-n of Q1-2,. .

  The ADC 12 corresponds to an analog-to-digital converter that converts a voltage value of a signal selectively output from the MUX 11 into a digital value.

  The I / O unit 14 corresponds to an input / output circuit that outputs a dimming pulse signal from the PDIM terminal to the subsequent CPU 15.

  The CPU (central processing unit) 15 takes in the digital value from the ADC 12 and the digital value of the dimming pulse signal output from the I / O unit 14, and the duty of the pulse drive signal applied to the gate of the switching element Qcnv. This corresponds to an arithmetic processing unit that generates a control command value for determining the ratio (on-duty) and thus the value of the output voltage Vout from the converter 3.

  The PWM (pulse width control) unit 16 generates a PWM pulse having a duty ratio based on a control command value calculated by the CPU 15 and sends the PWM pulse to the gate driver 8.

  With the configuration of the microcomputer 6, particularly in this embodiment, the gate voltages S2-1, S2-2,..., S2 of the switching elements Q1-1, Q1-2,. -N, drain voltages S3-1, S3-2, ..., S3-n, or current values Iout-1, Iout-2, ..., Iout-n flowing through the constant current source 4 as current detection resistors R1 −1, select one of the source voltages S1-1, S1-2,..., S1-n of the switching elements Q1-1, Q1-2,. The measured values of each column are statistically processed by calculating the total value, average value, minimum value, maximum value, and the like to generate monitor values. Next, immediately after the start of the operation of the converter 3, the on-duty of the PWM pulse from the PWM unit 16 is increased every predetermined period, and the output voltage Vout of the converter 3 is gradually increased. In the process, the current values Iout-1, Iout-2,..., Iout-n flowing through the LED arrays 2-1, 2-2,. Then, a change point where the monitor value changes from a variable value to a constant value is detected, and thereafter, an on-duty PWM pulse is always sent from the PWM unit 16 so that the monitor value always matches the change point. Output. In this manner, the output voltage control means 21 for controlling the output voltage Vout of the converter 3 is provided.

  More specifically, the output voltage control unit 21 sets a predetermined value in advance as one step, increases the output voltage Vout by one step, and the difference between the monitor values before and after changing the output voltage Vout is a constant threshold value. When the value is outside the range, the output voltage Vout is increased by one step in the next operation. Here, the arbitrarily specified step value can be defined by the resolution of the CPU 15 or the like.

  On the other hand, when the difference between the monitor values before and after changing the output voltage Vout is within the threshold range, that point becomes the change point, the output voltage Vout is decreased by one step in the next operation, and then again in the next operation. The output voltage Vout of the converter 3 is controlled by comparing the difference between the monitor values and the threshold value. Further, when the difference between the monitor values is within the threshold range, the operation of decreasing the output voltage Vout by one step is slower than the switching period (from the time constant between the choke coil Lcnv that is the output LC filter of the converter 3 and the capacitor Ccnv). (Which is also a long cycle).

  Further, as in the present embodiment, when there are a plurality of LED strings 2-1, 2-2,..., 2 -n as loads, current values Iout−1, Iout−2, .., Iout-n, when using the detection voltages (switch elements Q1-1, Q1-2,..., Q1-n source voltages S1-1, S1-2,..., S1-n) Any of the sum value, average value, or minimum value (excluding abnormal columns) of the detection voltages at the monitor value is used as a monitor value, and the gate voltages S2-1, S2 of the switch elements Q1-1, Q1-2,. When -2,..., S2-n are used, the monitor value is any one of the total value, average value, or maximum value of the gate voltages S2-1, S2-2,. The drain voltages S3-1, S3-2,..., S3 of the switch elements Q1-1, Q1-2,. When n is used, it is preferable to configure the output voltage control means 21 so as to use the minimum value of the drain voltages S3-1, S3-2,. .

  In the case where an abnormal row is detected and the abnormal row is removed from among the plurality of LED rows 2-1, 2-2,..., 2-n, The detection voltage of the abnormal row may be excluded. In that case, since the sum value in the monitor value is reduced by the exclusion, the reference may be lowered accordingly.

  A combination of the plurality of detection voltages described above may be used as a monitor value. A plurality of monitor values may be combined to finally obtain a necessary monitor value.

  Next, the operation of the configuration of the load driving device 1 will be described in detail with reference to FIGS. In the present embodiment, the output voltage Vout is controlled by the output voltage control means 21 by any one of the following three methods.

  The first control method is a method of generating monitor values from the current values Iout-1, Iout-2,..., Iout-n. First, when the input voltage Vin is input to the converter 3 at the start of the operation of the load driving device 1, no current flows through the LED rows 2-1, 2-2,. , A1-n constituting the constant current source 4 are switch elements Q1-1, Q1- corresponding to the current values Iout-1, Iout-2,. .., Q1-n based on the comparison result between the source voltages S1-1, S1-2,..., S1-n of the reference voltages V1,. It is in a state where it is completely conducted in the switching region. Thereafter, when the operation of the microcomputer 6 starts, the output voltage control means 21 gradually increases the on-duty of the PWM pulse output to the gate driver 8 so that the output voltage Vout from the converter 3 gradually increases. In each LED row 2-1, 2-2,..., 2-n, until the value of the output voltage Vout approaches the sum of the forward drop voltages Vf of the individual LEDs, The switch elements Q1-1, Q1-2,..., Q1-n continue to be conductive.

  FIG. 2 shows the current values Iout-1, Iout-2, Iout− flowing through the three LED rows 2-1, 2-2, 2-3 with the output voltage Vout of the converter 3 as the horizontal axis. The relationship between the voltages V (R1-1), V (R1-2), and V (R1-3) across the resistors R1-1, R1-2, and R1-3 that convert 3 into voltage Is shown in a graph. Here, 12Ω resistors R1-1, R1-2, and R1-3 are used as the current detectors. For example, the voltage V (R1-1) across the resistor R1-1 is Iout-1 [A ] X 12 [ohm]. The voltages V (R1-1), V (R1-2), and V (R1-3) across the resistors R1-1, R1-2, and R1-3 are the switch elements Q1-1 and Q1- shown in FIG. 2 and Q1-3 respectively correspond to source voltages S1-1, S1-2, and S1-3.

  In FIG. 3, the horizontal axis is the output voltage Vout of the converter 3, and the vertical axis is the voltage of each monitor value V (R1-n) sum, V (R1-n) avg, V (R1-n) min. Is a graph showing the relationship. In the figure, V (R1-n) sum is the sum of the voltages V (R1-1), V (R1-2), and V (R1-3) across the terminals, and V (R1-n) avg Is an average value of the voltages V (R1-1), V (R1-2), and V (R1-3) between both ends, and V (R1-n) min is the voltage V (R1-1 between the ends. ), V (R1-2), V (R1-3).

  FIG. 4 shows a periodic operation procedure of the output voltage control means 21 based on a timer (not shown) built in the microcomputer 6. This timer is obtained by dividing the clock frequency that is the frequency of the PWM pulse in the microcomputer 6. For example, when the clock frequency is 200 kHz, the timer frequency is set to 20 kHz. Then, in accordance with the period of this timer, control for increasing or decreasing the value of an output voltage Vout described later is performed.

  Hereinafter, the operation procedure of the output voltage control means 21 will be described. In the program step P1 of FIG. 4, the output voltage control means 21 obtains a measurement value necessary for calculating a monitor value in order to obtain each resistance R1- 1, R1-2, R1-3 voltage V (R1-1), V (R1-2), V (R1-3) are respectively taken in by MUX11, and these values are digitally converted from analog values by ADC12. Convert to value. Then, in the next program step P2, the output voltage control means 21 sets the monitor value M [0] as the monitor value M [0], the sum V of the voltages V (R1-1), V (R1-2), V (R1-3) across the terminals. (R1-n) sum, average value (R1-n) avg, and minimum value V (R1-n) min are calculated.

  As shown in FIG. 2 and FIG. 3 described above, in the region where the output voltage Vout of the converter 3 is 7.2 V or less, no current flows through the LED rows 2-1, 2-2 and 2-3. Voltages V (R1-1), V (R1-2), V (R1-3) between both ends, and monitor values V (R1-n) sum, V (R1-n) avg calculated in the microcomputer 6 , V (R1-n) min are both substantially zero. Since the output voltage control means 21 needs to first increase the output voltage Vout from the converter 3 until current begins to flow through the LED rows 2-1, 2-2, 2-3, the program step P3 in FIG. Thus, the monitor value M [0] of the current values Iout-1, Iout-2, and Iout-3 is an appropriate first voltage V1 corresponding to the target values of the current values Iout-1, Iout-2, and Iout-3. It is determined whether or not a threshold value (for example, a value corresponding to 50% of the voltage V1) is exceeded. If the monitor value M [0] is equal to or less than the first threshold value, the process proceeds to program step P7, and the gate driver By increasing the on-duty of the PWM pulse output to 8 by one step, the output voltage Vout from the converter 3 is increased by one step, and the program returns to the program step P1 to perform the next periodic operation procedure.

  On the other hand, if the monitor value M [0] exceeds the first threshold value in the program step P3, the output voltage control means 21 proceeds to the program step P4 and performs the program in the previous periodic operation procedure. Difference ΔM = M [0] −M [−1] between the monitor value M [−1] obtained in step P2 and the monitor value M [0] obtained in program step P2 in the current periodic operation procedure. Calculate In order to enable this calculation, the microcomputer 6 includes a storage unit (not shown) that stores and holds the previous monitor value M [−1].

  When the output voltage control means 21 calculates the difference ΔM in the program step P4, in the next program step S5, it is determined whether or not the difference ΔM is within the range between the second threshold value and the third threshold value. . Here, if the difference ΔM is larger than the second threshold and smaller than the third threshold, the output value Vout = 8.1 V is exceeded in FIG. 2 or FIG. 3, and the monitor value M [0] becomes flat. It is judged that it is in a state. At that time, since the output voltage control means 21 does not know where the output voltage Vout is controlled in the flat portion of the monitor value M [0], it does not stop the control for the converter 3 after the program step P5. In step P6, the on-duty of the PWM pulse is decreased by one step.

  On the other hand, if the difference ΔM is less than or equal to the second threshold value or greater than or equal to the third threshold value in the program step P5, it is determined in FIG. 1 that the monitor value M [0] with the output voltage Vout = 8.1V or less is increasing. In step P7, the on-duty of the PWM pulse is increased by one step. Thus, when the output voltage control means 21 determines the on-duty of the PWM pulse, the process returns to the program step P1 to perform the next periodic operation procedure.

  In this way, the boundary between the region where the monitor value M [0] is constant and flat by the constant current source 4 and the region where the monitor value M [0] is increasing is changed (for example, the output voltage Vout in FIGS. 2 and 3). Is 8.15V), and the loss of the constant current source 4 can be minimized by controlling the output voltage Vout from the converter 3 with the target change point of the monitor value M [0]. In the first control method for generating monitor values from the current values Iout-1, Iout-2, and Iout-3, the voltage V (R1-1 between both ends of the resistors R1-1, R1-2, and R1-3 is used. ), V (R1-2), and V (R1-3), the minimum value V (R1-n) min is preferably set to the monitor value M [0]. In this case, by controlling the output voltage Vout with the change point as the target value of the monitor value M [0], the constant current source 4 causes all the LED arrays 2-1, 2-2, 2-3 to The lowest output voltage Vout that can be operated can be supplied to each of the LED strings 2-1, 2-2, 2-3, and all the LED strings 2-1 are controlled while minimizing the loss of the constant current source 4. , 2-2, 2-3 can be reliably operated at a constant current.

  The second control method is a method of generating monitor values from the gate voltages S2-1, S2-2, and S2-3 of the switch elements Q1-1, Q1-2, and Q1-3. The output voltage control means 21 here is an operation voltage Vcc of operational amplifiers A1-1, A1-2, A1-3 and switching elements Q1-1, Q1-2, Q1-3 at program step P3 in FIG. A first threshold value is set between the threshold voltage Vthmax and the switch elements Q1-1, Q1-2, and Q1-3 connected to the LED arrays 2-1, 2-2, and 2-3. Thus, the value of one or more gate voltages S2-1, S2-2, S2-3 or the sum of the gate voltages S2-1, S2-2, S2-3 is three times the first threshold value (n It is determined whether or not the value has decreased to a value equal to or less than the value of the first threshold value. If the value exceeds three times the first threshold value (when n = 3), the process proceeds to program step P7, while the first If the value is equal to or smaller than three times the threshold value (when n = 3), the process proceeds to program step P4. Further, the output voltage control means 21 uses the gate voltages S2-1, S2 for the switch elements Q1-1, Q1-2, Q1-3 connected to all the LED rows 2-1, 2-2, 2-3, respectively. -2 and S2-3 are used as monitor values, or switch elements Q1-1, Q1- connected to normal LED strings 2-1, 2-2, 2-3, respectively, except for abnormal LED strings. 2 and Q1-3, the maximum value of the gate voltages S2-1, S2-2, and S2-3 is set to the monitor value M [0], and the operation procedure after the program step P4 shown in FIG. Run in the same way as

  Thereby, the same effect as the first control method can be obtained, and the loss of the constant current source 4 can be minimized. In the second control method for generating monitor values from the gate voltages S2-1, S2-2, and S2-3 of the switch elements Q1-1, Q1-2, and Q1-3, the normal LED array 2-1 , 2-2, and 2-3, the maximum values of the gate voltages S2-1, S2-2, and S2-3 for the switch elements Q1-1, Q1-2, and Q1-3 connected to the monitor elements M [0 ] Is preferable. In this case, by controlling the output voltage Vout with the change point as the target value of the monitor value M [0], the constant current source 4 causes all the LED arrays 2-1, 2-2, 2-3 to The lowest output voltage Vout that can be operated can be supplied to each of the LED strings 2-1, 2-2, and 2-3. , 2-2, 2-3 can be reliably operated at a constant current.

  In the third control method, the drain voltages of the switch elements Q1-1, Q1-2, and Q1-3 connected to the normal LED strings 2-1, 2-2, and 2-3, respectively, except for the abnormal LED string. In this method, monitor values are generated from S3-1, S3-2, and S3-3. In this case, the output voltage control means 21 is the program step P2 of FIG. 4 and the drain voltages S3-1, S3-2 and S3-3 of the switch elements Q1-1, Q1-2 and Q1-3 excluding the abnormal LED strings. Are calculated as a monitor value M [0]. Further, there is no processing of program step P3 shown in FIG. 4, and the operation procedure after program step P4 shown in FIG. 4 is executed in the same manner as the first control method and the second control method. The reason why the process of program step P3 can be omitted is that, as the output voltage Vout is increased by one step in program step P7, the current between the drain and source of each switch element Q1-1, Q1-2, Q1-3. Since the drain voltages S3-1, S3-2, and S3-3 are not easily affected by noise, the risk of erroneously detecting a change point from the monitor value M [0] can be avoided. It is.

  Thereby, the same effect as the first control method and the second control method is obtained, and the loss of the constant current source 4 can be minimized. Further, the drain voltages S3-1, S3-2, and S3-3 of the switch elements Q1-1, Q1-2, and Q1-3 connected to the normal LED rows 2-1, 2-2, and 2-3 are monitored. In the third control method for generating values, it is preferable to set the minimum value of the drain voltages S3-1, S3-2, and S3-3 as the monitor value M [0]. In this case, by controlling the output voltage Vout with the change point as the target value of the monitor value M [0], the constant current source 4 causes all the LED arrays 2-1, 2-2, 2-3 to The lowest output voltage Vout that can be operated can be supplied to each of the LED strings 2-1, 2-2, 2-3, and all the LED strings 2-1 are controlled while minimizing the loss of the constant current source 4. , 2-2, 2-3 can be reliably operated at a constant current.

  Next, a load driving device 1 according to a second embodiment of the present invention will be described with reference to FIG. In addition, the same code | symbol is attached | subjected to the same part as the said Example, and description of the same location is abbreviate | omitted. The present embodiment shows a modification of the constant current source 4, and the constant current source 4 </ b> A is based on the output of one comparator Comp <b> 1 in a plurality of LED rows 2-1, 2-2, 2-3. Is configured to operate at a constant current. Specifically, one pulse generation circuit 31 is connected to the output side of the operation comparator Comp1, and the same number as the LED rows 2-1, 2-2, 2-3 is connected to the output side of the pulse generation circuit 31. Each of the latch circuits 32 is connected to an LPF (low pass filter) 33, and the output side of the LPF 33 is connected to the gates of the switch elements Q1-1, Q1-2, and Q1-3. Has been.

  In the above configuration, any one of the voltages across the resistors R1-1, R1-2, and R1-3 is selectively output via the MUX 11, and is compared with the reference voltage V1 by the comparator Comp1. When the reference voltage V1 is higher, an H (high) level voltage is output from the comparator Comp1. Next, the pulse generation circuit 31 generates a pulse based on the reference timing signal from the pulse signal source 34, holds this in the latch circuit 32, and outputs it to the LPF 33. Next, the pulse is smoothed in the LPF 33, and the LED strings 2-1, 2-2, 2-3 selected as a waveform close to DC are operated in the non-switching region. By repeating the above operation, the LED strings 2-1, 2-2, 2-3 are controlled with constant current. In this embodiment, the switching elements Q1-1, Q1-2, and Q1-3 are controlled to be stopped by the switching element Q2 connected to the source side of the switching elements Q1-1. The switching element Q2 is a pulse signal source. The external filter reset signal from 35 operates.

  As described above, the load driving device 1 according to the present invention includes the converter 3 as a power conversion circuit that supplies power to a plurality of parallel LED rows 2-1, 2-2,. Switch elements Q1 connected in series with the LED strings 2-1, 2-2,..., 2-n and for controlling the current flowing through the connected LED strings 2-1, 2-2,. -1, Q1-2,..., Q1-n, respectively, and currents Iout-1, Iout-2,. , S1-n of the switch elements Q1-1, Q1-2,..., Q1-n, which are voltages corresponding to Iout-n, or the switch elements Q1-1, Q1- 2,..., Q1-n, which are gate voltages S2-1, S2-2,. ,..., 2-n, drains which are voltages at the positive side terminals of the switch elements Q1-1, Q1-2,..., Q1-n into which currents Iout-1, Iout-2,. .., S <b> 3-n is selected and used from the comparator 3 and is commonly applied to the LED rows 2-1, 2-2,. The output voltage Vout is measured each time in the process of increasing and stepping in a predetermined value at every predetermined cycle, and the total value, average value, minimum value, maximum value are measured between each column. Statistical processing for calculating values and the like, one of them is generated as a monitor value M [0], and the output voltage from the converter 3 to each of the LED rows 2-1, 2-2,. By detecting the inflection point of the monitor value M [0] that changes sharply as the change point, while changing Vout. A microcomputer 6 as a control circuit for controlling the output voltage Vout as a target value of the changing point monitor values M [0], and a.

  In this case, when the output voltage Vout from the converter 3 to the LED strings 2-1, 2-2,..., 2-n is changed using the monitor value statistically processed by the microcomputer 6, before and after the change. .., 2-n, the constant current source 4 causes each LED string 2-2 to be monitored regardless of which LED string 2-1, 2-2,. It is possible to detect a change point at which 1, 2-2,..., 2-n can operate at a constant current. Therefore, if the output voltage Vout is controlled using the change point detected by the microcomputer 6 as the target value of the monitor value M [0], the switch elements Q1-1, Q1-2,. , 2-n, the value of the reference voltage V1 that determines the set value of the output currents Iout-1, Iout-2,..., Iout-n flowing through the LED strings 2-1, 2-2,. -1, 2-2,..., 2-n itself may vary, and the voltage across the switching elements Q1-1, Q1-2,. Therefore, the loss of the switching elements Q1-1, Q1-2,..., Q1-n can be minimized. Further, even when the power of the LED strings 2-1, 2-2,..., 2-n increases or the number of strings increases, the monitor value M [0] is generated in the same manner, and the switch of the constant current source 4 is generated therefrom. The output voltage Vout can be controlled so as to minimize the loss of the elements Q1-1, Q1-2,..., Q1-n, and the reference value is not required for comparing the monitor values. -1, Q1-2,..., Q1-n can be reliably controlled regardless of component variations, and the switch elements Q1-1, Q1-2,. Even if Iout-2,..., Iout-n changes, the same circuit configuration can be used.

  In the present invention, when the output voltage Vout is increased, the output currents Iout-1, Iout-2,..., Iout-n, which are load currents, change from the non-constant current state following the change in the output voltage Vout. , A point at which the output currents Iout-1, Iout-2,..., Iout-n do not change with respect to a change in the output voltage Vout is detected, that is, a monitor value before and after changing the output voltage Vout. It is preferable that the microcomputer 6 detects a value at which M [−1] and M [0] change from a variably increased value to a constant value as the change point. This is the point where this state has shifted to the constant current operation, the voltage across the switching elements Q1-1, Q1-2,..., Q1-n is the smallest, and the loss of the constant current source 4 is the smallest. Because it is a point.

  In this case, the microcomputer 6 can detect the optimum change point by gradually increasing the output voltage Vout from a relatively low voltage when the operation of the load driving device 1 is started.

  In the present invention, the microcomputer 6 is configured to monotonously change or increase the output voltage Vout until a constant current flows through all of the switching elements Q1-1, Q1-2,..., Q1-n. .

  In this case, when the output voltage Vout is increased, the monitor value M [0] is set to the first value so that the change point of the monitor value M [0] is not erroneously detected at another stage due to the influence of noise or the like. Until the threshold value is reached, the output voltage Vout is monotonously increased without detecting the change point of the monitor value M [0]. Thereby, the change point of the monitor value M [0] can be correctly detected by the microcomputer 6.

  Although the embodiment of the present invention has been described above, this is an example for explaining the present invention, and the scope of the present invention is not limited to this embodiment. Of course, various modifications can be made without departing from the scope of the present invention. For example, the converter 3 is not limited to the illustrated step-up chopper circuit, but can be applied to a converter having any circuit configuration including one or more switching elements. Further, the same effect may be realized by changing the signal level and logic configuration of each part to those shown in the above embodiments. Furthermore, not only LEDs, but any load that performs constant current control may be used, and the number is not limited to one or more.

2-1, 2-2, ... 2-n LED row (load)
3 Converter (Power conversion circuit)
4 Constant current source 6 Microcomputer (control circuit)
Q1-1, Q1-2,... Q1-n switch element

Claims (2)

A power conversion circuit that supplies power to one or more rows of loads;
A constant current source which is connected in series with the load of each row and incorporates a switch element for controlling a current flowing through the connected load;
Using either the voltage corresponding to the current flowing through the load, the voltage of the control terminal of the switch element, or the voltage of the + side terminal of the switch element through which the current flows from the load, the total value between the columns Generating a monitor value subjected to statistical processing for calculating an average value, a minimum value, or a maximum value, and when the output voltage from the power conversion circuit to the load of each column is increased, the monitor value is changed from the variable value. A control circuit that performs control to increase or decrease the output voltage using the change point as a target value by detecting a value when the value becomes a constant value as a change point of the monitor value. Load driving device. Wherein all until the current flowing through the switching element, to increase the output voltage monotonically, the load driving device according to claim 1, wherein the configuring the control circuit.