JP7116288B2 - power transmission equipment - Google Patents

Description

Application of Article 30, Paragraph 2 of the Patent Law Feb. 24, 2018, presentation materials for the 2018 Young Researcher Research Conference in the Tohoku Region, "Sound, Light, Radio Waves, Energy, Systems and Their Applications"

TECHNICAL FIELD The present invention relates to a power transmission device, and more particularly to a new circuit design for a wireless power transmission device.

Since the 2005 proposal by Karalis et al. of the Massachusetts Institute of Technology (MIT), research on wireless power transmission has been active (see Patent Documents 1, 2 and 3). The inventions described in Patent Documents 1, 2, and 3 are power transmission technologies that use an AC power source as a power source. It is considered necessary to match the resonant frequency 2π√LC of the side resonant circuit (LC circuit) with the resonant frequency 2π√LC of the power receiving side resonant circuit (LC circuit). However, in practice, the power-supplying-side resonant circuit and the power-receiving-side resonant circuit interact with each other, thereby causing new resonance. It is technically common knowledge that resonance (multiple resonance) between the power supply side resonance circuit and the power reception side resonance circuit, including this new resonance, should not occur.

According to Patent Document 2, a power-supply-side resonant circuit and a power-receiving-side resonant circuit are coupled by a magnetic field component (magnetic field coupling), and a mutual inductance M is formed between the power supply coil and the power reception coil. A new resonant circuit formed by the mutual inductance M, the power-supply-side resonant circuit, and the power _- receiving-side resonant circuit has a resonant frequency _fr2 different from the resonant frequency fr1, and multiple resonance occurs. When trying to make the drive frequency fo of AC power supplied to the power supply _side resonance circuit follow the resonance frequency fr1, the drive frequency fo follows the resonance frequency _fr2 instead of the original target resonance frequency _fr1 . The resonance frequency fr ₂ is an undesirable resonance point, and it is desirable to remove it. In the conventional AC theory, multiple resonance has been avoided.

Moreover, the invention described in Patent Document 1 uses an AC power supply of 10 kHz to 50 GHz, the invention described in Patent Document 2 uses an AC power supply with a driving frequency fo of about 100 kHz, and the invention described in Patent Document 3 uses several hundred An AC power supply of kHz to several MHz is required. In particular, Patent Document 1 reports experimental data in a frequency band around 10 MHz. The frequency band power supply circuit (0th order circuit) as described in Patent Documents 1 to 3 uses an expensive switching power supply from a commercial power supply to create a highly accurate direct current, and then a large number of power semiconductor devices are complicated. It is also created by precisely switching and converting a DC pulse cut out on a rectangular wave into an AC pseudo or equivalently by means of PWM (Pulse Width Modulation) or the like. At this time, power loss such as resistance loss that occurs in the power semiconductor element and switching loss that rapidly increases with an increase in frequency occurs. In addition, switching elements are likely to be destroyed by back electromotive force induced in the coil or by excessive voltage rise due to resonance, and the higher the frequency and the greater the power, the more difficult the circuit design becomes. On the other hand, if you want to send power over long distances, you have to raise the frequency. As described above, the conventional wireless power transmission device based on the alternating current theory has problems that the device is complicated, the overall power transmission efficiency is low, the device is fragile, the reliability is low, and the cost is high. For these reasons, the conventional technology cannot realize wireless power transmission that efficiently transmits power over long distances, which will be required in the future. In other words, the circuit design itself of the power transmission device based on the AC theory is required to be studied.

U.S. Patent Application Publication No. 2008/0278264 Japanese Patent No. 5549745 Japanese Patent No. 5462953

SUMMARY OF THE INVENTION In view of the above problems, the present invention focuses on a transient response that is not based on the conventional AC theory, and aims to provide an inexpensive power transmission device that simplifies the circuit configuration and increases the power transmission efficiency.

In a first aspect of the present invention, (a) a power transmission side capacitor is connected in parallel to the power transmission side capacitor and electrostatic energy sent from the power transmission side capacitor is accumulated as magnetic energy, and this magnetic energy is circulated to the power transmission side capacitor. (b) a DC power source forming a circuit connecting between one terminal and the other terminal of the power transmission side capacitor; and (c) one terminal of the power transmission side capacitor and the DC power source. and (d) a receiving side coil that faces the transmitting side coil and receives magnetic energy from the transmitting side coil, the receiving side. A secondary circuit having a receiving side capacitor that is connected in parallel to the coil and accumulates the magnetic energy accumulated in the receiving side coil as electrostatic energy, and (e) connection between one terminal and the other terminal of the receiving side capacitor. and (f) a load-side diode whose anode is connected to one terminal of the power-receiving capacitor and whose cathode is connected to the load element. The gist is that it is a device. In the power transmission device according to the first aspect, electrical energy can be transmitted contactlessly from the primary circuit to the secondary circuit.

In a second aspect of the present invention, (a) a power transmission side capacitor is connected in parallel to the power transmission side capacitor and electrostatic energy sent from the power transmission side capacitor is stored as magnetic energy, and this magnetic energy is circulated to the power transmission side capacitor. (b) a DC power source forming a circuit connecting between one terminal and the other terminal of the power transmission coil; and (c) one terminal of the power transmission coil and the DC power source. and (d) one electrode connected to one node that connects the power transmission side capacitor and the power transmission side coil in parallel. (e) a second mutual coupling capacitor having one electrode connected to the other node connecting the power transmission side capacitor and the power transmission side coil in parallel; and (f) the first mutual coupling One electrode is connected to the other electrode of the capacitor, the other electrode is connected to the other electrode of the second mutual coupling capacitor, the receiving side capacitor receives electrostatic energy from the primary side circuit, and the receiving side capacitor is connected in parallel. and (g) a circuit connecting between one terminal and the other terminal of the power receiving side coil. and (h) a load-side diode having an anode connected to one terminal of the power-receiving-side coil and a cathode connected to the load element. This is the gist. Similarly to the power transmission device according to the first aspect, the power transmission device according to the second aspect can also transmit electrical energy from the primary side circuit to the secondary side circuit in a contactless manner.

According to the present invention, which is based on a circuit design that departs from the conventional AC theory, by using characteristic harmonic transmission, which is a phenomenon during transient response, it is possible to simplify the circuit configuration, improve the power transmission efficiency, and provide an inexpensive power transmission device. .

FIG. 1(a) is a circuit diagram showing an outline of an example of the power transmission device according to the first embodiment of the present invention, and FIG. 1(b) shows terminals of the power transmission side capacitor of the circuit shown in FIG. FIG. 10 is a waveform diagram of an inter-voltage. FIG. 2(a) is a waveform diagram of voltages between terminals of the power transmission side capacitor and the power reception side capacitor of the power transmission device according to the first embodiment, and FIG. 2(b) follows the waveform of FIG. 2(a). 4 is a detailed waveform diagram; FIG. It is a figure explaining the transient response at the time of step-inputting to an LC parallel circuit. FIG. 2 is a circuit diagram showing a mounted circuit of the power transmission device according to the first embodiment; FIG. 4 is a diagram illustrating a large-signal equivalent circuit of a MOSFET used in the power transmission device according to the first embodiment; FIG. 6(a) is a schematic diagram for explaining the importance of the plane spacing between the coils of the power transmission device according to the first embodiment, and FIG. 6(b) is for charging the battery of an electric vehicle (EV). FIG. 10 is a bird's-eye view for explaining a magnetic coupling degree control mechanism that adjusts the surface spacing between coils when applied. FIG. 7A is a bird's-eye view illustrating a mechanism for adjusting magnetic coupling of coils of the power transmission device according to the first embodiment, FIG. b) and FIG. 7(c) show an example of a magnetic coupling degree control mechanism that uses a magnetic plate to adjust the magnetic coupling of the coil. FIG. 8(a) is a block diagram illustrating an example of the hardware configuration of the magnetic coupling degree control mechanism of the power transmission device according to the first embodiment, and FIG. 8(b) illustrates another example. It is a block diagram. FIG. 4 is a schematic diagram for explaining the operation of the power transmission device according to the first embodiment divided for each timing along the time series; FIG. 10(a) is a circuit diagram showing an outline of an example of the power transmission device according to the second embodiment of the present invention, and FIG. 10(b) is a specific implementation circuit of the circuit of FIG. 10(a). It is a circuit diagram showing. FIG. 10 is a timing chart for explaining a power supply method of the power transmission device according to the second embodiment; 4A and 4B are schematic diagrams illustrating a power supply method of the power transmission device according to the second embodiment, (a) during charging, (b) during characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3, ( c) during transmission; and (d) during characteristic harmonic transmission between primary circuit 2 and secondary circuit 3 . FIG. 13(a) is a circuit diagram showing an outline of an example of the power transmission device according to the third embodiment of the present invention, and FIG. 13(b) is a specific implementation of the circuit shown in FIG. 13(a). It is a circuit diagram showing a circuit. 10 is a flowchart for explaining a power supply method for a power transmission device according to a third embodiment; FIG. 11 is a timing chart for explaining a method of supplying power to a power transmission device according to a third embodiment; FIG. 16(a) is a circuit diagram showing an outline of an example of the power transmission device according to the fourth embodiment of the present invention, and FIG. 16(b) is a specific implementation of the circuit shown in FIG. 16(a). It is a circuit diagram showing a circuit. FIG. 17A is a waveform diagram obtained by simulating the voltage between the terminals of the power transmission side capacitor and the power reception side capacitor in the power transmission device according to the fourth embodiment, and FIG. FIG. 10 is a waveform diagram obtained by a mounting circuit of a voltage between terminals of a power receiving side capacitor; FIG. 18(a) is a waveform diagram when the equivalent coupling coefficient K=0.00 of the voltage between the terminals of the power transmission side capacitor and the power reception side capacitor in the power transmission device according to the fourth embodiment, and FIG. FIG. 4 is a waveform diagram when the equivalent coupling coefficient K=0.1; FIG. 19(a) is a waveform diagram when the equivalent coupling coefficient K=0.6 of the voltage between the terminals of the power transmission side capacitor and the power reception side capacitor in the power transmission device according to the fourth embodiment, and FIG. It is a waveform diagram when the equivalent coupling coefficient K=0.8, and FIG. 19C is a waveform diagram when the equivalent coupling coefficient K=0.88. FIG. 20(a) is a waveform diagram when the equivalent coupling coefficient K=0.6 of the voltage between the terminals of the power transmission side capacitor and the power reception side capacitor in the power transmission device according to the fourth embodiment, and FIG. FIG. 10 is a waveform diagram when the equivalent coupling coefficient K of currents flowing through a power transmitting side coil and a power receiving side coil is K=0.6; FIG. 11 is a graph showing changes in transmission efficiency with respect to capacitance of a capacitor of the power transmission device according to the fourth embodiment; FIG. It is a graph which shows the efficiency of the power transmission apparatus which concerns on 4th Embodiment. FIG. 11 is a flow chart illustrating a first power supply method for a power transmission device according to a fourth embodiment; FIG. FIG. 24(a) is a timing chart for explaining the first power supply method of the power transmission device according to the fourth embodiment, and FIG. 24(b) is a timing chart for explaining the second power supply method. FIG. 14 is a flow chart illustrating a second power supply method of the power transmission device according to the fourth embodiment; FIG. FIG. 11 is a circuit diagram showing an outline of an example of a power transmission device according to a fifth embodiment of the present invention; FIG. 27(a) is a waveform diagram of currents flowing through the power transmission side coil and the power reception side coil of the power transmission device according to the fifth embodiment, and FIG. FIG. 4 is a waveform diagram of voltages between terminals of a power transmission side capacitor and a power reception side capacitor; FIG. 4 is a circuit diagram used in approximate simulation for explaining a bumpy sawtooth wave in the power transmission device according to the first embodiment of the present invention; FIG. 29 is a diagram showing transient response characteristics of a sawtooth wave with knobs obtained by approximate simulation for the circuit of FIG. 28; FIG. 29 is a diagram for explaining how the bumps of the sawtooth wave obtained by approximate simulation using the circuit of FIG. 28 change with repetition periods; 29 is a simplified circuit diagram in which parasitic capacitance, parasitic resistance, etc. are omitted from the circuit of FIG. 28; FIG. FIG. 32 is a diagram for explaining that a W-type transient response characteristic is obtained by approximate simulation for the simplified circuit of FIG. 31; FIG. 8 is a circuit diagram used when approximating the transient response characteristics of the power transmission device according to the second embodiment of the present invention by changing the power supply voltage and the load voltage; FIG. 34 is a diagram illustrating transfer of electric energy with transient response characteristics obtained by approximate simulation for the three circuits of FIG. 33; FIG. 4 is a schematic diagram illustrating W-type transient response characteristics of the power transmission device according to the first embodiment of the present invention;

Next, first to fifth embodiments of the present invention will be described with reference to the drawings. In the following description of the drawings, the same or similar parts are denoted by the same or similar reference numerals. However, it should be noted that the drawings are schematic, and the relationship between thickness and planar dimension, the ratio of thickness of each member, etc. are different from the actual ones. Therefore, specific thicknesses and dimensions should be determined with reference to the following description. In addition, it is a matter of course that there are portions with different dimensional relationships and ratios between the drawings.

Further, the first to fifth embodiments shown below are examples of apparatuses and methods for embodying the technical idea of the present invention. , shape, structure, arrangement, etc. are not specified as follows. Various modifications can be made to the technical idea of the present invention within the technical scope defined by the claims. Furthermore, the directions of "left and right" and "up and down" in the following description are merely definitions for convenience of description, and do not limit the technical idea of the present invention. Therefore, for example, if the page is rotated 90 degrees, "left and right" and "up and down" are read interchangeably, and if the page is rotated 180 degrees, "left" becomes "right" and "right" becomes "left". Of course it will be. Similarly, the helix orientation of the spiral, as shown in FIGS. 6(a) to 7(c), is merely a choice for convenience of explanation, and depending on the actual design circumstances, left-handed or right-handed. It is also possible to select right-hand winding.

(First embodiment)
The power transmission device according to the first embodiment of the present invention includes a primary side circuit 2 and a secondary side circuit 3, as shown in FIG. 1(a). The primary circuit 2 is connected in parallel to a power transmission side capacitor C ₁ for accumulating electrostatic energy, and is connected in parallel to the power transmission side capacitor C ₁ to store the electrostatic energy sent from the power transmission side capacitor C ₁ as magnetic energy. It is an LC resonance circuit having _a power transmission side coil L1 that circulates to the power transmission _side capacitor C1 and is magnetically coupled to the power reception side coil L2 of the _secondary side circuit 3 to transmit and receive magnetic energy. A DC power supply 5 and a _primary side drive switch SW1, which are connected in series, are connected in parallel to a transmission side capacitor C1. _A DC power supply 5 supplies a DC voltage to the transmission side capacitor C1.

As will be described later, the “primary side drive switch SW1” is a circuit element that limits free vibration of the primary side circuit 2 . By limiting free oscillation, the primary side drive switch SW1 implements transient current-voltage changes in the primary side circuit 2. FIG. The DC power supply 5 may be a pseudo-constant voltage source, or may be a DC power supply with a simple structure that is simply rectified and a power supply containing a large ripple component. Moreover, an inexpensive DC power supply 5 can be employed. The secondary circuit 3 is opposed to the power transmission side coil L ₁ and is separated from the power transmission side coil L ₁ . The power reception side coil L ₂ receives magnetic energy from the power transmission _side coil L ₁ . It is an LC resonance circuit having a power receiving side capacitor _C2 that accumulates accumulated magnetic energy as electrostatic energy.

A load-side diode D2 and a load element 6, which are connected in series with each other, are connected in parallel to the receiving-side capacitor _C2 . A rechargeable battery such as a lithium (Li) ion battery for use in an electric vehicle (EV) can be used as the load element 6, for example. FIG. 1A schematically shows an equivalent circuit of a lithium-ion battery by way of a series-parallel circuit of resistors and capacitors. Lithium-ion batteries include current collectors, electrolyte resistors, and electrical double layer capacitors and resistors formed at interfaces within the battery. The load-side diode D2 is connected so that its anode faces the receiving _- side resonator 2 and its cathode faces the load element 6, limiting the flow direction of the charging current IC to one direction.

In FIG. ₁ (a), E is the voltage across the DC power supply 5 and the equivalent internal resistance r1, VC1 is the voltage across the terminals of the transmission side capacitor C1, VC2 is the voltage across the terminals of the receiving _side capacitor _C2 , and VC2 is the voltage across the terminals of the receiving _side capacitor _C2 . Let VC _S be the charging voltage measured between the terminals of 6, and let I _C be the current flowing through the load-side equivalent floating resistance r _L . The equivalent internal resistance r1 _{approximately} indicates the internal impedance of the DC power supply 5 with a resistance value. FIG. ₁ (b) shows the transient response waveform of the inter-terminal voltage VC1 obtained by actual measurement when the primary side drive switch SW1 is turned on and off.

In the first embodiment, it is assumed that the value of the charging voltage _VCS of the load element 6 in the initial state is a high value close to the charge completion voltage (close to full charge). At time t= ₀ , the transmitting capacitor C1 is not charged and the terminal voltage VC1= ₀ . When the primary side drive switch SW1 is turned on at t=0, the inter-terminal voltage E of the DC power supply 5 is input stepwise. A step input at t=0 initially causes a charging current to capacitor C1, whose value is E/r1. At this time coil L1 generates a back EMF to block the incoming current, so the current into L1 is zero. As shown in FIG. 1B, the power transmission side capacitor C ₁ is charged with a time constant τ ₁ determined by the equivalent internal resistance r ₁ , the power transmission side capacitor capacitance C ₁ , the power transmission side coil inductance L ₁ , and the mutual inductance M. , the voltage _VC1 across the terminals increases. The time constant τ ₁ is mainly a value that depends on parameters related to the product C ₁ ·r ₁ of the capacitance C ₁ of the transmission side capacitor C ₁ and the resistance value C ₁ of the equivalent internal resistance r ₁ .

As the voltage of C1 gradually rises and the current becomes smaller, the back electromotive force of L1 becomes smaller and the current begins to flow to L1. This causes the voltage across C1 to drop slightly. At this point, SW1 is closed (t=t1). The closing time t1 of this switch should be such that SW1 is not destroyed by the voltage applied to SW1 caused by the back electromotive force generated by turning off the coil when the current starts to flow. When the _{primary side} drive switch SW1 is turned off at t=t1, current flows from the power transmission _side capacitor C1 to the power transmission coil L1, and full _- scale discharge starts. The electric energy stored in the power transmission _side capacitor C1 tries to move to the power transmission _side coil L1. At this time, due to the magnetic field generated around L1 due to the current flowing through the power transmission _side coil L1, _an electromotive force is generated in the power reception _side coil L2 coupled by the mutual inductance M, and current flows. As will be described later, if the characteristics of the primary side circuit 2 and the secondary side circuit 3 are in harmony at this time, the power receiving side capacitor _C2 is most efficiently charged by this transmitted power. That is, power is most efficiently transmitted from the primary side circuit 2 to the secondary side circuit 3 . Electromotive force is generated in L1 by a magnetic field generated around the power receiving _side coil L2 due to the current flowing through the power receiving _side coil L2. Due to the original voltage and this electromotive force, the voltage of the power transmission side coil L1 becomes _a sawtooth wave that deviates from a sine wave that is not a normal alternating current waveform. Slightly past the lowest voltage portion of the sawtooth wave, the sawtooth wave characteristic rises while rising like a bump.

The time constant τ ₂ until this bump-like swelling occurs is mainly determined by the capacitance C ₁ of the transmission side capacitor C ₁ , the inductance L ₁ of the transmission side coil L ₁ and the parasitic resistance of the transmission side coil L ₁ R _str (L ₁ ) is a value that depends on parameters expressed in a relationship similar to Equation (4) described later. However, it is necessary to consider the mutual inductance M=M(t) between the inductance of the power transmission side coil L ₁ and the power reception side coil L ₂ , which is a value dependent on time t.

At t=t ₂ , the voltage of the primary side capacitor becomes a maximum peak again, and at this time, the voltage of the secondary side capacitor becomes a value close to a minimum value. When the primary side drive switch SW1 is turned on again in accordance with this peak, the inter-terminal voltage E of the DC power supply 5 is input stepwise again. As before, a charging current flows to the capacitor C1 at _first , and its value is the value obtained by subtracting the voltage across the capacitor C1 at t ₌ t2 from the voltage E across the terminals of the DC power supply ₅ and dividing it by _r1 . is. _At this time coil L1 generates _a back EMF to block the incoming current, so the current into L1 is zero. As the voltage of C1 gradually rises and the current becomes smaller, the back electromotive force of L1 becomes smaller and the current begins to flow to L1. This _causes the voltage across C1 to drop slightly. At this point, SW1 is closed ₍ t=t3). _The time _t3 to close this switch should be such that _SW1 is not destroyed by the voltage applied to SW1 by the back electromotive force generated by the transmitting coil L1 when the current starts to flow in the coil when it is turned off. . After the _primary side drive switch SW1 is turned off at t=t3 _, current starts to flow from the power transmission side capacitor C1 to the power transmission coil L1, and full _- scale discharge starts. The electric energy stored in the power transmission _side capacitor C1 tries to move to the power transmission _side coil L1. At this time, due to the magnetic field generated around L1 due to the current flowing through the power transmission _side coil L1, _an electromotive force is generated in the power reception _side coil L2 coupled by mutual inductance M, and current flows. As will be described later, if the characteristics of the primary side circuit 2 and the secondary side circuit 3 are in harmony at this time, the power receiving side capacitor _C2 is most efficiently charged by this transmitted power. That is, power is most efficiently transmitted from the primary side circuit 2 to the secondary side circuit 3 . Electromotive force is generated in L1 by a magnetic field generated around the power receiving _side coil L2 due to the current flowing through the power receiving _side coil L2. Due to the original voltage and this electromotive force, the voltage of the power transmission side coil L1 becomes _a sawtooth wave that deviates from a sine wave that is not a normal alternating current waveform. Slightly past the lowest voltage portion of the sawtooth wave, the sawtooth wave characteristic rises while rising like a bump.

As a result _, after t=t3, as shown on the right side of FIG. ₁ (b), the voltage VC1 across the terminals of the power transmission side capacitor C1 decreases and becomes _a negative value again. When the voltage VC1 across the terminals of the power transmission _side capacitor C1 becomes _a negative value, the electrical energy stored in the power transmission _side coil L1 begins to circulate to the power transmission side capacitor C1, as shown on the right end of FIG. ₁ (b). At the same time, the terminal voltage VC1 of the power transmission _side capacitor C1 starts to increase due to the circulating current, becomes _a positive value, and further increases. The time up to this point is determined by the time constant τ ₂ determined by the capacitance C ₁ of the power transmission side capacitor, the inductance L ₁ of the power transmission side coil, and the mutual inductance M. As shown in FIG. 1(b), due to the step input and cutoff of the voltage E between the terminals of the DC power supply 5 by the _primary side drive switch SW1, the change in the voltage between the terminals VC1 is not a sinusoidal waveform in normal AC theory, but a , shows the transient response of a repetitive waveform with dull rising and falling sawtooth waves.

When the time constant inherent in the circuit characteristics of the primary circuit 2 and the time constant inherent in the circuit characteristics of the secondary circuit 3 are in harmony, the electrical energy of the primary circuit 2 is most efficiently transmitted to the secondary circuit 3. resulting in characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 . The transient response waveforms of the voltage VC1 between the terminals of the primary circuit ₂ and the voltage VC2 between the terminals of the _secondary circuit 3 when characteristic harmonic transmission occurs between the primary circuit 2 and the secondary circuit 3 are shown in FIG. 2(a). In FIG. 2(a), as in FIG. 1(b), due to the step input and cutoff of the voltage E between the terminals of the DC power supply 5 by the primary side drive switch SW1, the voltage VC1 between the terminals has _a sawtooth wave with knobs. It shows a repetitive transient response waveform with rising and falling characteristics. _In the _transient response waveform shown by the solid line in FIG. _{At i} , the primary side drive switch SW1 is turned on.

At this time, as indicated by the dashed line on the left side of FIG. After the power receiving side capacitor _C2 of the _secondary circuit 3 is charged and the voltage across the terminals of the power receiving side capacitor _C2 reaches the peak value, the power receiving side capacitor _C2 starts discharging and the voltage across the terminals becomes VC ₂ is starting to decrease. When the primary side drive switch SW1 is turned on at t=t _i , the inter-terminal voltage E of the DC power supply 5 is input stepwise. The terminal voltage VC2 of the receiving side capacitor _C2 of the _secondary side circuit 3 decreases to a negative value at t ₌ ti.

By the step _input of t= _{t i} , as shown near the left _side of the center of FIG. The power transmission side capacitor C1 of the _primary side circuit ₂ is charged with the time constant τ1, and the terminal voltage _VC1 increases. As shown in FIG. 2(a), the voltage VC1 across the terminals of the _primary circuit 2 reaches a peak value and then begins to decrease. When the primary side drive switch SW1 is turned off at t=t _{i+1 ,} the power transmission side capacitor C1 of the primary side circuit ₂ starts full-scale discharge, and the electric energy stored in the power transmission side capacitor C1 is transmitted. Move to _side coil L1. The terminal voltage VC2 of the receiving side capacitor _C2 of the _secondary side circuit 3 is a negative value between t=t _i and t=t _i+1 as indicated by the dashed line.

After t= _{t i+1 ,} as indicated by the solid line near the center of _FIG . _The voltage VC1 across the terminals of the power transmission side capacitor C1 decreases to _a negative value, and the electric energy accumulated in the power transmission _side capacitor C1 is transferred to the power transmission _side coil L1. The electrical energy stored in the power transmission _side coil L1 is transferred to the secondary side circuit 3 by characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 between the primary side circuit 2 and the secondary side circuit 3. It is wirelessly transmitted to the power receiving _side coil L2. The electrical energy wirelessly transmitted to the power receiving side coil L2 of the _secondary side circuit 3 is accumulated in the power receiving side capacitor _C2 of the secondary side circuit 3 .

As a result, after t=t _i+1 , the terminal voltage VC2 of the power receiving side capacitor _C2 of the _secondary circuit 3 starts to increase, as indicated by the dashed line near the center of FIG. 2(a). After reaching a peak value, the terminal voltage VC2 of the receiving side capacitor _C2 of the _secondary side circuit 3 starts to decrease and becomes a negative value as indicated by the dashed line on the right side of FIG. 2(a). As a condition of this embodiment, the value of the charging voltage _VCS of the load element ₆ in the initial state is a high value close to the charging completion voltage (close to full charge). Since _VC2 exceeds the charging voltage VCS of the load element ₆ near the peak value, current flows through the load element 6, and the electric energy accumulated in the receiving side capacitor C2 moves to the load element 6, and the load element 6 rechargeable batteries are charged.

After t=t _{i+1 ,} when the terminal voltage VC ₁ of the power transmission side capacitor C ₁ reaches the minimum negative value as indicated by the solid line in the center of FIG. The stored electric energy begins to circulate to the power transmission _side capacitor C1, and the voltage VC1 across the terminals of the power transmission _side capacitor C1 starts to increase due to the circulating current, becomes _a positive value, and further increases. As indicated by the dashed line on the right side of FIG. 2(a), when the terminal voltage VC1 takes _a positive value, the terminal voltage VC2 of the receiving _- side capacitor _C2 takes a negative value.

When the voltage VC1 across the terminals of the power transmission side capacitor C1 increases to _a positive value and the primary side drive switch SW1 is turned on again at t=t _{i +2} , the voltage E across the terminals of the DC power supply 5 is input stepwise. be. Due to the step input at t=t _{i +2} , the power transmission side capacitor C1 is charged as indicated by the solid line on the right side of FIG. 2( _a ), and the terminal voltage VC1 increases. As shown by the solid line in FIG. ₂ ( _a ), the terminal voltage VC1 reaches the peak value E0 and then starts to decrease again. When the primary side drive switch SW1 is turned off at t=t _{i +3 ,} the power transmission side capacitor C1 starts discharging again, and the electric energy stored in the power transmission side capacitor C1 moves to the power transmission _side coil L1. . The terminal voltage VC2 of the receiving side capacitor _C2 of the secondary side circuit 3 has a negative value between t=t _{i+2 and} t=t _i+3 as indicated by the dashed line.

The electrical energy stored in the power transmission _side coil L1 is transferred to the secondary side circuit 3 by characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 between the primary side circuit 2 and the secondary side circuit 3. It is wirelessly transmitted to the power receiving _side coil L2. The electrical energy wirelessly transmitted to the power receiving side coil L2 of the _secondary side circuit 3 is accumulated in the power receiving side capacitor _C2 of the secondary side circuit 3 . As shown in FIG. 2(a), the step input and cutoff of the inter-terminal voltage E by the primary side drive switch SW1 causes the change in the inter-terminal voltage VC1 of the _primary side circuit 2 to be as follows: It shows a transient response of a repetitive waveform with rising and falling characteristics with a dull sawtooth wave. On the other hand, the change in the inter-terminal voltage VC2 of the _secondary circuit 3 exhibits a transient response of a repetitive waveform like a thinned-out triangular wave, but it is not a sinusoidal waveform in normal AC theory. As can be seen from FIG. 2(a), the "decimated triangular wave" can be interpreted as a waveform obtained by reversing the polarity of the trapezoidal wave. In any case, the vibration waveform of the primary circuit 2 and the vibration waveform of the secondary circuit 3 are not mutually symmetrical.

_FIG . 2B shows the transient response waveforms of the terminal voltage _VC1 and the terminal voltage _VC2 shown in FIG. It is an actually measured _waveform of a transient response to which a charging current IC to a rechargeable battery, which is the load element 6, is added. At t=t _i , the primary side drive switch SW1 is turned on to store electric charge in the power transmission side capacitor C1, and then at t=t _{i+1 ,} the primary side drive switch SW1 is turned off. A characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 to the secondary side circuit 3 occurs. When the primary side drive switch SW1 is turned on, the equivalent internal resistance r1 of the DC power supply ₅ is small, so the voltage E across the terminals of the DC power supply 5 indicated by the thick solid line in FIG. It shows _a change superimposed on the voltage VC1 between the voltages.

Description will be made focusing on the transient response after t=t _i+1 in FIG. 2(b). As indicated by the solid line near the center of FIG. 2(b), the voltage VC1 across the terminals of the power transmission side capacitor C1 decreases after t=t _{i +1 and} becomes a negative value. The electric energy accumulated in the power transmission _side capacitor C1 is transferred to the power transmission _side coil L1 and accumulated in the power transmission _side coil L1. The electrical energy stored in the power transmission _side coil L1 is transferred to the secondary side circuit 3 by characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 between the primary side circuit 2 and the secondary side circuit 3. It is transmitted to the power receiving _side coil L2. Since the electrical energy transmitted to the power receiving coil L2 of the _secondary circuit 3 is stored in the power receiving capacitor _C2 of the _secondary circuit 3, the voltage VC2 across the terminals of the power receiving capacitor _C2 is As indicated by the dashed line near the center of 2(b), it starts increasing from a negative value after t=t _i+1 .

The voltage VC2 across the terminals of the receiving capacitor _C2 of the _secondary circuit 3 is a negative value between t=t _i and t=t _i+1 as indicated by the dashed line from the left side of the center. . _The voltage VC2 across the terminals of the power receiving side capacitor _C2 increases from a negative value at t=t _i+1 , becomes a positive value, further increases, reaches a peak value, and then reaches the right side of FIG. begins to decrease as indicated by the dashed line in . _When the terminal voltage VC2 decreases, the electrical energy accumulated in the power receiving side capacitor _C2 flows into the load element 6 as the charging current _IC indicated by the dashed line on the right side of FIG. charged. Almost in synchronism with the increase in the charging current IC indicated by the _dashed line, the voltage _VCS between the terminals of the load element 6 indicated by the dotted line on the right side of FIG. It shows a transient response that decreases in synchronism with the decrease in _charging current IC. When the charging current _I.sub.C decreases to zero, the voltage _VCS across the terminals of the load element 6 stops decreasing, starts increasing, and the voltage across the terminals of the load element 6 becomes a steady value. A change in Vcs according to Ic occurs due to the presence of a series-parallel circuit of resistors and capacitors, the equivalent circuit of which is illustrated in FIG. 1(a).

After t=t _{i+1 ,} when the terminal voltage VC ₁ of the power transmission side capacitor C ₁ reaches the minimum negative value as indicated by the solid line in the center of FIG. The stored electric energy begins to circulate to the power transmission _side capacitor C1, and the voltage VC1 across the terminals of the power transmission _side capacitor C1 starts to increase due to the circulating current, becomes _a positive value, and further increases. As indicated by the dashed line on the right side of FIG. ₂ (b), when the terminal voltage VC1 becomes _a positive value, the terminal voltage VC2 of the receiving side capacitor _C2 becomes a negative value.

When the voltage VC1 across the terminals of the power transmission side capacitor C1 increases to _a positive value and the primary side drive switch SW1 is turned on again at t=t _{i +2} , the voltage E across the terminals of the DC power supply 5 is input stepwise. be. Due to the step input at t=t _{i +2} , the power transmission side capacitor C1 is charged as indicated by the solid line on the right side of FIG. 2(b), and the inter _- terminal voltage VC1 increases. As described above, since the equivalent internal resistance r1 of the DC power supply ₅ is small, when the primary side drive switch SW1 is turned on, the voltage E across the terminals of the DC power supply 5 indicated by the thick solid line in FIG. _A change is superimposed on the voltage VC1. As indicated by the solid line superimposed on the terminal voltage E on the right end of FIG. ₂ (b), the terminal voltage VC1 reaches its peak value and then starts to decrease again. When the primary side drive switch SW1 is turned off at t=t _{i +3 ,} the power transmission side capacitor C1 starts discharging again, and the electric energy stored in the power transmission side capacitor C1 moves to the power transmission _side coil L1. . The terminal voltage VC2 of the receiving side capacitor _C2 of the secondary side circuit 3 has a negative value between t=t _{i+2 and} t=t _i+3 as indicated by the dashed line.

The behavior before t=t _i is similar, and the voltage VC2 across the terminals of the receiving side capacitor _C2 starts decreasing after reaching a peak value, as indicated by the dashed line on the left side of FIG. ₂ (b). . _When the terminal voltage VC2 increases and exceeds Vcs, the electrical energy accumulated in the power receiving side capacitor _C2 is transferred to the load element 6 as the charging current _IC indicated by the dashed line on the left side of FIG. 2(b). , the load element 6 is charged. Almost in synchronism with the increase in the charging current IC indicated by the _dashed line, the voltage _VCS across the terminals of the load element 6 indicated by the dotted line on the left side of FIG. It shows a transient response that decreases in synchronism with the decrease in _charging current IC. When the charging current _I.sub.C decreases to zero, the voltage _VCS across the terminals of the load element 6 stops decreasing, starts increasing, and the voltage across the terminals of the load element 6 becomes a steady value. A change in Vcs according to Ic occurs due to the presence of a series-parallel circuit of resistors and capacitors, the equivalent circuit of which is illustrated in FIG. 1(a).

Then, when the primary side drive switch SW1 is turned off at t=t _{i+1 already} described, the voltage VC1 across the terminals of the power transmission _side capacitor C1 becomes negative as indicated by the solid line in the center of FIG. 2(b). start decreasing towards the value. In this way, the electrical energy stored in the power transmission side coil L1 is doubled by the characteristic harmonic transmission between the _primary side circuit 2 and the secondary side circuit 3 between the primary side circuit 2 and the secondary side circuit 3. It is wirelessly transmitted to the power receiving coil L ₂ of the secondary circuit 3 . The electrical energy wirelessly transmitted to the power receiving side coil L2 of the _secondary side circuit 3 is accumulated in the power receiving side capacitor _C2 of the secondary side circuit 3 . As shown in FIG. 2(b), the step input and cutoff of the inter-terminal voltage E by the primary side drive switch SW1 causes the change in the inter-terminal voltage VC1 of the _primary side circuit 2 to be as follows: The voltage VC2 across the terminals of the _secondary circuit 3 exhibits a transient response of a repetitive waveform like a thinned triangular wave.

FIG. 3(a) is a circuit diagram for explaining the step response of the primary circuit 2 as a single circuit when there is no electromagnetic coupling between the primary circuit 2 and the secondary circuit 3. FIG. In FIG. 3A, the voltage across the terminals of the DC power supply is E, the voltage across the terminals of the transmission _side capacitor C1 is VC1, and the current flowing through the transmission _side coil _L1 is the transmission _side coil current IL1. When the primary side drive switch SW1 is turned on at t=0 ms, the inter-terminal voltage E of the DC power supply 5 is input stepwise. Due to the step input at t=0 ms, as shown in FIG. 3B, the time constant determined by the capacitance C ₁ of the power transmission side capacitor and the inductance L ₁ of the power transmission side coil is the rising waveform V _rise of the power transmission side capacitor C ₁ charged. , and the terminal voltage _VC1 increases with the rising waveform V _rise . At the same time, the power transmission side coil current _IL1 also starts to increase as shown in FIG. 3(b).

As shown in FIG. 3B, the inter-terminal voltage VC1 increases with the _rising waveform _Vrise , reaches a peak value at t=0.075 ms, and then starts decreasing. It can be seen that the power transmission side coil current _IL1 continues to increase even after the inter _- terminal voltage VC1 starts decreasing, and the electric energy of the power transmission _side capacitor C1 continues to move to the power transmission _side coil L1. When the primary-side drive switch SW1 is turned off at t=0.15 ms, the power-transmitting-side capacitor C1 starts full _- scale discharge and sharply decreases with the falling waveform V _fall . At this time, the power transmission side coil current _IL1 continues to increase, and until the peak value of the power transmission side coil current _IL1 is reached at t=0.17 ms, the electric energy stored in the power transmission _side capacitor C1 is Move to coil _L1 .

As _a result, after t=0.15 ms, as shown in FIG. ₃ (b), the voltage across the terminals of the power transmission side capacitor C1 is determined by the time constant determined by the capacitance C1 of the power transmission _side capacitor and the inductance L1 of the power transmission side coil. _VC1 decreases and the terminal voltage _VC1 becomes zero when the peak value of the sending coil current _IL1 is reached at t=0.17 ms. After t=0.17 ms, the voltage VC1 across the terminals of the power transmission _side capacitor C1 becomes _a negative value. When the voltage VC1 across the terminals of the power transmission _side capacitor C1 becomes _a negative value, the value of the power transmission side coil current IL1 begins to decrease, and the electrical energy stored in the power transmission _side coil _L1 flows back to the power transmission _side capacitor C1. Begin to. Then, as shown in FIG. 3(b), when the value of the power transmission _side coil current _IL1 becomes zero at t=0.29 ms, the terminal voltage VC1 of the power transmission _side capacitor C1 increases due to the circulating current. Start. When the value of the transmitting side coil current _{I L1} becomes the minimum negative value at t=0.4 ms, the terminal voltage VC1 becomes zero, and thereafter the terminal voltage VC1 becomes _a positive value. becomes and further increases.

The circuit shown in FIG. 3A does not turn on the primary drive switch SW1 again after turning off the primary drive switch SW1 at t=0.15 ms. In other words, in the case of the circuit shown in FIG. 3(a), since free vibration occurs in the shaded region on the right side of FIG. However, in the circuit shown in FIG. 1(a), the primary-side drive switch SW1 drives in a forced step response that periodically repeats ON/OFF. are outside the scope of the power transmission device according to the first embodiment. _In the case of forced step response, as shown in FIGS. Shows the transient response of the waveform. Further, the voltage VC2 across the terminals of the _secondary circuit 3 exhibits a transient response of a repetitive waveform such as a thinned triangular wave.

That is, in the power transmission device according to the first embodiment, the time constant inherent in the circuit characteristics of the primary side circuit 2 and the circuit of the secondary side circuit 3 are used instead of the resonance based on the sine wave in the normal AC theory. When the time constant inherent in the characteristics is in harmony, the electrical energy of the primary circuit 2 is efficiently transmitted to the secondary circuit 3 by characteristic harmonic transmission between the primary circuit 2 and the secondary circuit 3. . In order to harmonize the time constant inherent in the circuit characteristics of the primary side circuit 2 and the time constant inherent in the circuit characteristics of the secondary side circuit 3, the product of the power transmission side capacitor C1 and the power transmission side coil L1 and the power reception side capacitor Basically, the product of C2 and power receiving side coil L2 should be the same, and the time constants of the power transmitting side and power receiving side, including parasitic resistance and stray capacitance, should be matched or integral multiples to harmonize them. The simplest method is to equalize the capacities of the power transmitting side capacitor C1 and the power receiving side capacitor C2 including the parasitic resistance of the capacitors, and set the inductances of the power transmitting _side coil L1 and the power receiving _side coil L2 including the parasitic resistance of the coils to equality. It should be noted that when parasitic resistance exists in the coil and the capacitor, the transmission efficiency is maximized when the inductance L of the coil and the capacitance C of the capacitor satisfy equation (19), as will be described later. be.

As the primary side drive switch SW1 shown in FIG. 1(a), in addition to a mechanical switching element such as an electromagnetic contactor, as a more preferable embodiment, a power semiconductor switching element capable of high-speed switching is used. Examples of power semiconductor switching elements include field effect transistors (FET), static induction transistors (SIT), bipolar transistors (BJT), and thyristors such as gate turn-off thyristors (GTO) and static induction thyristors (SI thyristors). preferred. In particular, voltage-driven switching elements such as insulated gate field effect transistors (MISFET), insulated gate static induction transistors (MISSIT), insulated gate bipolar transistors (IGBT), and MOS-controlled SI thyristors consume less power. preferred. Based on availability in the market and evaluation of the internal resistance of semiconductor switching elements for electric power, MOS field effect transistors (MOSFETs), which are a type of MISFET, are currently adopted as shown in the circuit shown in Fig. 4(a). It is possible to

In a power transmission device that uses a rechargeable battery for an EV as a load element 6, a large amount of Joule heat is generated due to the flow of a large current, and heat generation of several hundred watts or more is accompanied, and the power transmission device is damaged. It becomes a heater. In the power transmission device according to the first embodiment, only one power semiconductor switching element is used as the primary side drive switch SW1. A structure that prevents this can be easily designed, and the generation of stray resistance, stray capacitance, and stray inductance can be minimized. Since the heat generated by the floating resistance (parasitic resistance) of the power transmission _side coil L1 and the power reception _side coil L2 is also large, measures such as cooling the power transmission _side coil L1 and the power reception _side coil L2 with air or water are preferable.

One method of suppressing the generation of Joule heat in a high-power power transmission device such as an EV vehicle is to increase the voltage of the primary side circuit ₂ so that the winding ratio of the power transmission side coil L1 and the power reception side coil L2 is _two . It is to set the voltage of the secondary circuit 3 to the optimum voltage of the load element 6 . When a power semiconductor switching element is employed as the primary side drive switch SW1, simple control of ON/OFF control of the power semiconductor switching element suffices. be.

As described above, according to the power transmission device according to the first embodiment, since the primary side drive switch SW1 is a simple design with only one piece, the voltage of the primary side circuit 2 is increased to increase the joule on the primary side circuit 2 side. It is easy to design to minimize power loss due to heat generation. Energy loss due to Joule heat generation can also be reduced, so according to the power transmission device according to the first embodiment, the loss of the power supply circuit (zero-order circuit) in the case of high-power power transmission such as for EV in-vehicle use can be reduced. Overall power transmission efficiency is increased, contributing to the resolution of human energy problems.

In the mounting circuit shown in FIG. 4( a ), considering the circulating current from the power transmission side coil L ₁ , the first free wheel diode (freewheel diode) FWD ₁ is the source of the MOSFET as the first semiconductor switching element Q1・Connected in parallel between the drains as a protective element. In order to prevent the circulating current from the power transmitting coil L1 from _circulating to the DC power supply 5, a power supply diode D1 is connected in series between the DC power supply 5 and the first semiconductor switching element Q1. In the mounting circuit shown in FIG. 4(a), the equivalent impedance X _Leq of the load element 6 is expressed by approximating it with the charge capacity _Cs .

In ac theory, which relies on a normal steady-state sinusoidal wave, the mutual inductance M between the sending coil L ₁ and the receiving coil L ₂ is given by the coupling coefficient K _AC :

M=K _AC (L1・^{L2) 1/2} _{( 1} )

can be shown. Then, the mutual induction between the series circuit of the power transmission side capacitor C1 and the power transmission _side coil L1 and the series circuit of the power reception _side capacitor _C2 and the power reception _side coil L2 is expressed by the following coupling equation using the mutual inductance M

_L1di1 /dt+( ₁ /C1)∫i1dt+ _Mdi2 /dt= ₀ ( ₂ )
L2di1/dt+( ₁ / _C2 ) _{∫i2dt + Mdi1} /dt=0 (3)

represented by In equations (2) and (3), ∫ is the integral symbol. That is, the mounting circuit shown in FIG. 4A can be expressed as shown in FIG.

However, since the power transmission device according to the first embodiment is a transient response transmission technology that does not rely on the sine wave AC theory, the representation of the equivalent circuit in FIG. It is only a schematic diagram in Maxwell's equations in the time-varying case can be solved analytically when the time-varying relies on a sine wave. However, when there is a sawtooth-like time change as shown in FIGS. 1(b) to 2(b), it is extremely difficult to analytically solve Maxwell's equations. Therefore, the mutual inductance M used in AC theory is a time-dependent parameter represented by a function M(t) with t as time in the present invention, and the equivalent circuit representation of FIG. need to be careful.

FIG. 5 shows a large-signal equivalent circuit of an nMOSFET used as an example of the first semiconductor switching element Q1 in the mounting circuit shown in FIG. 4(a). As shown in FIG. 5A, a typical nMOSFET has a p-type substrate 71, an n ⁺ -type source region 72 and an n ⁺ -type drain region 73 facing each other across the surface of the p-type substrate 71 serving as a channel region. I am letting A gate electrode 84 is provided on the channel regions of the source region 72 and the drain region 73 with a gate oxide film 81 having a thickness _{TOX interposed} therebetween. A source electrode 82 and a drain electrode 83 are in ohmic contact with the source region 72 and the drain region 73, respectively.

As shown in FIG. 5A, in a general nMOSFET, between the gate electrode 84 and the source region 72 is a gate-source capacitance C _GS , and between the gate electrode 84 and the drain region 73 is a gate-drain capacitance C GS . A capacitance _CGD exists between the gate electrode 84 and the substrate 71, and a gate-substrate capacitance _CGB exists. Furthermore, between the source region 72 and the substrate 71 there is a source-substrate capacitance _CBS , and between the drain region 73 and the substrate 71 there is a drain-substrate capacitance _CBD . In the equivalent circuit shown in FIG. 5B, the drain resistance R _D connected in series between the drain electrode and the channel region, and the source resistance R _S connected in series between the source electrode and the channel region are connected to the channel region. A configuration is shown connected in series with a constant current source of current I _DS provided at .

In the power transmission device according to the first embodiment shown in FIG. 4, the drain resistance R _D and the source resistance R _S of the MOSFET shown in FIG. Therefore, it is necessary to select an element with a small on-resistance for the first semiconductor switching element Q1. Therefore, in the implementation circuit shown in FIG. 4(a), the on-resistance of the _first semiconductor switching element Q1 is included in the equivalent internal resistance r1 of the DC power supply 5, and the time constant inherent in the circuit characteristics of the primary side circuit 2 must be determined.

When the primary side drive switch SW1 is turned off at t=t1 in FIG. ₁ (b), the primary side circuit 2 becomes an RLC series circuit. According to the AC theory, if the parasitic resistance of the power transmitting coil L1 is R _str ( _L1 ), then the damping coefficient _ζ1 of the _RLC series circuit is:

ζ ₁ = (R _str (L ₁ )/2) (C ₁ /L ₁ ) ^1/2 (4)

is represented. However, for the inductance of the power transmission _side coil L1, it is actually necessary to consider the mutual inductance M=M(t) shown in FIG. 4(b). It is difficult to explain.

Ignoring the load diode D2 on the load side of the secondary circuit 3, the load element 6, etc., it can be regarded as an RLC series circuit. If the AC theory is adopted ignoring the load diode D2 and the load element ₆ , the damping coefficient _ζ2 of the _secondary circuit ₃ is

ζ ₂ = (R _str (L ₂ )/2) (C ₂ /L ₂ ) ^1/2 (5)

is represented. In equation (5) as well, it is necessary to consider the mutual inductance M=M(t) shown in FIG. 4(b) for the inductance of the power receiving _side coil L2.

According to the AC theory, the resonance frequency of the RLC series circuit configured with the _primary -side drive switch SW1 turned off at t=t1 in FIG.

f _o1 = (1/2π) (C ₁ L ₁ ) - ^1/2 (6)

can be approximated as As described above, in equation (6), it should be noted that it is actually necessary to consider the mutual inductance M=M(t) shown in FIG. 4(b) as the inductance of the receiving _side coil L1. is. Similarly, the resonance frequency of the RLC series circuit when ignoring the load diode D2 on the load side of the secondary circuit 3, the load element 6, etc. is, according to the AC theory,

f _o2 =( ¹ /2π)( _C2 ·L2) _-1/2 (7)

can be approximated as In equation (7) as well, the inductance of the power receiving _side coil L2 must take into consideration the mutual inductance M=M(t) shown in FIG. 4(b).

The power transmitting side coil L ₁ and the power receiving side coil L ₂ of the power transmission device according to the first embodiment may be spiral planar coils as shown in FIGS. 6(a) to 7(c), for example. can. In the steady state where the AC theory holds, the primary side circuit 2 and the secondary side circuit 3 have an equivalent coupling coefficient K between the power transmitting side coil L ₁ and the power receiving side coil L ₂ as shown in FIG. 4(b). , can be approximated by a circuit magnetically coupled with mutual inductance M. Here, the "equivalent coupling coefficient K" is a pseudo coupling coefficient in an unsteady state defined during transient response, which is equivalent to the coupling coefficient K _AC derived from AC theory, and is strictly a time-dependent parameter. Therefore, even in characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3, the time constant inherent in the circuit characteristics of the primary side circuit 2 and the time constant inherent in the circuit characteristics of the secondary side circuit 3 It can be evaluated by the same "magnetic coupling degree" as the theoretical coupling coefficient _KAC .

6(a) to 7(c) are schematic diagrams specifically showing the structures of the power transmitting side coil L ₁ and the power receiving side coil L ₂ of FIG. 4(a). In the power transmission device according to the first embodiment, for example, a wiring cable having a conductor cross-sectional area of 16 mm ² is wound nine times to form a spiral planar coil having a diameter of approximately 30 cm. These two spiral planar coils with a diameter of about 30 cm are arranged in parallel and in a non-contact manner with a gap of d in between. The efficiency of characteristic harmonic transmission between the primary circuit 2 and the secondary circuit 3 from the primary circuit 2 to the secondary circuit 3 is the degree of magnetic coupling similar to the coupling coefficient K _AC defined in AC theory. Value dependent. The degree of magnetic coupling depends on the spacing d between the two spiral planar coils, and it is necessary to control the spacing d between the two spiral planar coils.

The degree of magnetic coupling is determined by mechanically adjusting the positional relationship between two spiral planar coils, inserting a magnetic material between the two spiral planar coils, or inserting a magnetic material around the two spiral planar coils. Adjustments can be made by placing, attaching to a preformed coupling utilizing the attraction or repulsion acting between two spiral planar coils, or the like.

As will be described in the fourth embodiment and the like, when the mutual relationship between the power transmitting side coil L ₁ and the power receiving side coil L ₂ has an equivalent coupling coefficient K that can be approximately approximated to the coupling coefficient K _AC =0.6 of the AC theory, It is suitable for characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 . In the case of a spiral planar coil with 9 turns of wiring cables each having a conductor cross-sectional area of 16 mm ² , the distance d needs to be about 0 Cm to 2.0 Cm in order to realize the equivalent coupling coefficient K = 0.6. . On the other hand, in order to realize the mutual relationship between the power transmitting side coil L ₁ and the power receiving side coil L ₂ under the condition of the equivalent coupling coefficient K that can be approximately approximated to the coupling coefficient K _AC =0.1 of the AC theory, the distance d must be 10 cm. degree. As schematically shown by exaggerating (enlarging) the power transmitting _side coil L1 and the power receiving side coil L2 in _FIG . In order to charge using the power transmission device according to No. 1, the distance d between the power transmission side coil L ₁ and the power reception side coil L ₂ is controlled to about 10 Cm using the rear wheel stop 33 as a magnetic coupling degree control mechanism, Efficient non-contact power supply can be performed.

As a magnetic coupling degree control mechanism for controlling the distance d between the power transmission _side coil L1 and the power reception _side coil _L2 , as shown in _FIG . By interposing the spacer 32 with a length d ₀ , it is possible to control the distance d=d ₀ between the power transmitting side coil L ₁ and the power receiving side coil L ₂ . In addition, the power transmission side coil L ₁ corresponding to the value of the degree of magnetic coupling that is important for the efficiency of characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 from the primary side circuit 2 to the secondary side circuit 3 and Regarding the interrelationship between the power receiving _side coil L2, the magnetic coupling degree control mechanism can be configured by parameters _{other than} the distance d between the power transmitting side coil L1 and the power receiving side coil L2.

For example, as shown in FIG. 7(b), a magnetic coupling degree control mechanism is configured by inserting a magnetic plate 31C such as ferrite having a magnetic permeability _μr between the power transmission _side coil L1 and the power reception side coil L2 _. good too. The degree of magnetic coupling between the power transmission side coil L ₁ and the power reception side coil L ₂ can be controlled by the vertical insertion position of the magnetic plate 31C or the insertion area of the magnetic plate 31C. The magnetic plate 31C may be inserted on the back side of the power transmission side coil L ₁ as shown in FIG. 7C instead of between the power transmission side coil L ₁ and the power reception side coil L ₂ . do not have. The degree of magnetic coupling between the power transmission _side coil L1 and the power reception _side coil L2 is determined by the insertion position of the magnetic plate 31b in the vertical direction or the ratio of the insertion area of the magnetic plate 31b to the area of the power transmission _side coil L1. You can control the value. Although not shown, it is possible to similarly control the value of the degree of magnetic coupling between the power transmission _side coil L1 and the power reception _side coil L2 by inserting a magnetic plate behind the power reception _side coil L2. is of course.

Specifically, in order to control the insertion position in the vertical direction of the magnetic plate 31C shown in FIG. 7B and the insertion position in the vertical direction of the magnetic plate 31b shown in FIG. ) may be provided on the side of the power supply device where the power transmission side coil L ₁ is provided. The distance measuring unit 41 may be provided with a magnetic coupling degree control mechanism having a light emitting section 411 and a light receiving section 412 . If the light receiving section 412 is an optical time-of-flight (TOF) distance measuring element d, the light emitting section 411 emits pulsed light. For example, a near-infrared LD (laser diode) or a near-infrared LED is used for pulsed light emission. Pulsed light reflected from the power receiving _side coil L2 or the rear part of the EV is irradiated onto the light receiving part 412 through a lens, a BPF (band pass filter), or the like. The distance measurement unit 41 may be configured as a laser interferometer or the like.

The light receiving section 412 of the distance measuring unit 41 is connected to the distance calculation section 421 of the logic operation control section 42 shown in FIG. 8(a). _The output of the light receiving unit 412 is input to the distance calculation unit 421 constituting the magnetic coupling degree control mechanism via an output buffer and an interface (not shown). The computations required for distance measurement between coils L2 _are performed. The logical operation control unit 42 stores data necessary for logical operations such as calculation of the degree of magnetic coupling in the logical operation control unit 42 and magnetic plates necessary for realizing a desired equivalent coupling coefficient (pseudo-coupling coefficient). A data storage device 45 is connected which stores data on the insertion position in the vertical direction of the . Although not shown, the logic operation control unit 42 may be connected to a program storage device or the like storing a program for instructing the operation of the logic operation control unit 42 .

The data of the distance between the power transmission side coil L ₁ and the power reception side coil L ₂ calculated by the distance calculation section 421 is transmitted to the coupling coefficient calculation section 422 of the logical operation control section 42 . The coupling coefficient calculation unit 422 calculates the current magnetic field between the power transmission side coil L ₁ and the power reception side coil L ₂ based on the distance data between the power transmission side coil L ₁ and the power reception side coil L ₂ calculated by the distance calculation unit 421 . Find the value of the degree of physical connectivity. The coupling coefficient calculation unit 422 further calculates the moving distance of the magnetic plate from the data of the insertion position in the vertical direction of the magnetic plate necessary for realizing the desired equivalent coupling coefficient stored in the data storage device 45. and output to the coupling coefficient adjustment driving device 43 . Coupling coefficient adjustment driving device 43 of the magnetic coupling degree control mechanism shown in FIG. Drive control is performed so that the insertion position in the vertical direction of the plate 31C and the insertion position in the vertical direction of the magnetic plate 31b shown in FIG. 7(c) are at desired positions. A well-known position control mechanism such as a step motor can be employed for the coupling coefficient adjusting drive device 43 . In this manner, feedback control can be performed based on the output of the distance measuring unit 41 so that the insertion positions in the vertical direction of the magnetic plates 31C and 31b are at desired positions.

In the computer system of the magnetic coupling degree control mechanism including the logical operation control unit 42 shown in FIG. It is also possible to use any combination appropriately selected from among. Also, the cache memory may be a combination of a primary cache memory and a secondary cache memory, or may have a hierarchy with a tertiary cache memory. The logical operation control unit 42 shown in FIG. 8A can constitute a computer system using a microprocessor (MPU) or the like implemented as a microchip. In addition, as a logic operation control unit 42 that constitutes the computer system of the magnetic coupling degree control mechanism, a digital signal processor (DSP) that has enhanced arithmetic operation functions and specializes in signal processing, and an embedded device equipped with a memory and peripheral circuits A microcontroller (microcomputer) or the like for the purpose of control may be used. Alternatively, the main CPU of current general-purpose computers may be used as the logical operation control unit 42 .

FIG. 9 is a diagram showing the operation of the mounted circuit shown in FIG. As shown in FIG. 9A, at the timing when the first semiconductor switching element Q1 as the primary side drive switch SW1 is turned on, electric charge is _first stored in the power transmission side capacitor C1. As shown in FIG. 9A, the internal resistance r _on1 of the first semiconductor switching element Q1 at this time. At the timing of FIG. 9( _a ), when the terminal voltage VC1 of the power transmission _side capacitor C1 begins to increase, part of the electrical energy accumulated in the power transmission _side capacitor C1 is transferred to the power transmission side as the power transmission side coil current _IL1 . It is transferred to the coil L ₁ and accumulated in the power transmission side coil L ₁ . The electrical energy of the power transmission side coil L ₁ is transmitted to the power reception side coil L ₂ of the secondary side circuit 3 although it is slight. The electrical energy transmitted to the power receiving side coil _L2 of the _secondary side circuit 3 is spent to charge the power receiving side capacitor _C2 of the secondary side circuit 3 as the power receiving side coil current IL2. However, at the timing of FIG. 9(a), the terminal voltage VC2 of the power receiving _side capacitor _C2 is a negative value.

Next, at the timing shown in FIG. 9B, when the first semiconductor switching element Q1 as the _primary side drive switch SW1 is turned off (off state), the voltage VC1 across the terminals of the power transmission _side capacitor C1 decreases. The electric energy accumulated in the power transmission _side capacitor C1 is transferred to the power transmission _side coil _L1 as the power transmission _side coil current IL1 and accumulated in the power transmission side coil L1. The electrical energy stored in the power transmission _side coil L1 is transferred to the secondary side circuit 3 by characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 between the primary side circuit 2 and the secondary side circuit 3. It is wirelessly transmitted to the power receiving _side coil L2. The electrical energy transmitted to the power receiving side coil _L2 of the _secondary side circuit 3 is stored in the power receiving side capacitor _C2 of the secondary side circuit 3 as the power receiving side coil current IL2. At the timing of FIG. 9( _b ), the terminal voltage VC2 of the power receiving side capacitor _C2 becomes a positive value. After the voltage VC2 across the terminals of the receiving _side capacitor _C2 reaches a peak value, it starts to decrease.

_When the terminal voltage VC2 decreases, the electrical energy stored in the power receiving capacitor _C2 flows into the load element 6 as the charging current _ICs , as shown in FIG. 9(b), and the load element 6 is charged. . However, as the terminal voltage VC2 decreases, part of the electrical energy stored in the power receiving _side capacitor _C2 flows into the power receiving side coil _L2 as shown in _FIG . , and electrical energy is also stored in the power receiving _side coil L2. The electrical energy accumulated in the power receiving side coil L2 is distributed between the primary side circuit ₂ and the secondary side circuit 3, as shown in FIG. 9(c). It is circulated to the power transmission side coil L1 of the _primary side circuit 2 by characteristic harmonic transmission.

As shown in FIG. ₉ (c), the electric energy stored in the power transmission _side coil L1 begins to circulate to the power transmission side capacitor C1 due to the power transmission side coil current IL1 circulated to the power transmission _side coil _L1 , and power transmission is performed. _The voltage VC1 across the _side capacitor C1 starts to increase due to the circulating current. Therefore, as shown in FIG. 9(c), an ammeter 461 for measuring the power transmission side coil current IL1 circulated in the power transmission side coil _L1 and _a voltmeter 462 for measuring the terminal voltage VC1 are connected to the _primary side. If provided in the circuit 2, the magnitude of the electric energy circulated from the secondary side circuit 3 can be measured. That is, the current is transmitted from the primary circuit 2 to the secondary circuit 3 by characteristic harmonic transmission between the primary circuit 2 and the secondary circuit 3 by the ammeter 461 and the voltmeter 462 provided in the primary circuit 2 . Efficiency of wireless transmission can be measured.

For this reason, a magnetic coupling degree control mechanism for controlling the insertion position in the vertical direction of the magnetic plate 31C shown in FIG. 7B and the insertion position in the vertical direction of the magnetic plate 31b shown in FIG. In order to configure, a transmission efficiency measuring unit 46 as shown in FIG. 8B may be provided on the side of the power supply device where the power transmission _side coil L1 is provided. The transmission efficiency measurement unit 46 is an ammeter 461 and a voltmeter 462 provided in the primary circuit 2, as shown in FIG. 9(c).

The ammeter 461 and the voltmeter 462 shown in FIG. 9(c) are connected to the transmission efficiency calculation unit 471 of the logical operation control unit 47 constituting the magnetic coupling degree control mechanism shown in FIG. 8(b). . Outputs of the ammeter 461 and the voltmeter 462 are input to the transmission efficiency calculation unit 471 via an output buffer and an interface (not shown), and the transmission efficiency calculation unit 471 calculates the power transmission _side coil L1 and the power reception _side coil L2. Arithmetic processing necessary for measuring the transmission efficiency between is performed. The logic operation control unit 47 stores data necessary for logic operations such as calculation of transmission efficiency in the logic operation control unit 47 and data on the insertion position in the vertical direction of the magnetic plate required to achieve desired transmission efficiency. A data storage device 45 is connected. Although not shown, the logic operation control unit 47 may be connected to a program storage device or the like that stores a program for instructing the operation of the logic operation control unit 47 .

The data of the transmission efficiency between the power transmission _side coil L1 and the power reception _side coil L2 calculated by the transmission efficiency calculation unit 471 is transmitted to the coupling coefficient calculation unit 472 of the logic operation control unit 47 that constitutes the magnetic coupling degree control mechanism. be done. The coupling coefficient calculation unit 472 calculates the current transmission efficiency between the power transmission side coil L ₁ and the power reception side coil L ₂ from the data of the transmission efficiency between the power transmission side coil L ₁ and the power reception side coil L ₂ calculated by the transmission efficiency calculation unit 471 . Find the value of the degree of magnetic coupling of The coupling coefficient calculation unit 472 further calculates the moving distance of the magnetic plate from the data of the insertion position in the vertical direction of the magnetic plate necessary for realizing the desired transmission efficiency, which is stored in the data storage device 45. , to the coupling coefficient adjustment driving device 43 . The coupling coefficient adjustment driving device 43 determines the insertion position in the vertical direction of the magnetic plate 31C shown in FIG. Drive control is performed so that the insertion position in the vertical direction of the magnetic plate 31b shown in 7(c) is at a desired position. A well-known position control mechanism such as a step motor can be employed for the coupling coefficient adjusting drive device 43 . In this manner, the magnetic coupling degree control mechanism shown in FIG. 8(b) can set the insertion position of the magnetic plate 31C or the magnetic plate 31b in the vertical direction to a desired position based on the output of the transmission efficiency measurement unit 46. It can be controlled by feedback.

As described in FIG. 8(a), the data storage device 45 forming part of the magnetic coupling degree control mechanism shown in FIG. 8(b) includes a plurality of registers, a plurality of cache memories, a main Any combination appropriately selected from a group including an auxiliary storage device is also possible. The logic operation control unit 47 shown in FIG. 8B can configure a computer system using an MPU or the like mounted as a microchip. Further, as the logical operation control unit 47 constituting the computer system, a DSP specialized for signal processing with enhanced arithmetic operation functions, a microcomputer equipped with a memory and peripheral circuits and intended for embedded device control, etc. may be used. . Alternatively, the main CPU of current general-purpose computers may be used as the logical operation control section 47 .

Conventionally known “resonance” means that the sinusoidal vibration of the primary circuit 2 is transmitted to the free-oscillating secondary circuit 3, causing the secondary circuit 3 to vibrate at the same frequency as the primary circuit 2. It is a vibrating concept. The power transmission device according to the first embodiment of the present invention includes a primary-side drive switch SW1 that limits the free oscillation of the primary-side circuit 2 and realizes a transient current-voltage change in the primary-side circuit 2. Therefore, the non-sinusoidal sawtooth transient response characteristic can be transmitted to the secondary side circuit 3 by the concept of "characteristic harmonic transmission" proposed for the first time by the present inventors. By using the non-sinusoidal sawtooth transient response characteristic, a complicated and expensive AC power supply circuit for generating sinusoidal vibration in the primary side circuit 2 as in the prior art is no longer required.

As already described, in the power transmission device according to the first embodiment, the time constant inherent in the primary circuit 2 and the time constant inherent in the secondary circuit 3 are harmonized to generate electric energy in the primary circuit 2. is transmitted to the secondary circuit 3 . The characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3, for example, equalizes the capacitance of the power transmission side capacitor C1 and the power reception _side capacitor _C2 including the parasitic resistance of the capacitors, and the power transmission side coil L ₁ and the inductance of the receiving side coil L ₂ should be made equal including the parasitic resistance of the coil. Therefore, if the ON/OFF repetition period of the primary side drive switch SW1 is set to 500 to 600 μs, for example, the capacities of the power transmission side capacitor C ₁ and the power reception side capacitor C ₂ are set to be the same within the range of 400 μF to 600 μF, for example. , the inductances of the power transmitting side coil L ₁ and the power receiving side coil L ₂ may be set to be the same within a range of 5 μH to 20 μH, for example.

A sawtooth-shaped transient response waveform as shown in FIGS. 1(b) and 2 includes a plurality of bumps in one cycle. It is extremely difficult to analytically solve the transient response waveform of the power transmission device according to the first embodiment. Therefore, although it is an approximation, the circuit shown in FIG. 28 is simulated by alternating current theory. In FIG. 28, the capacitance of the capacitor and the inductance of the coil of the primary side circuit 2 and the secondary side circuit 3 are set to the same value. That is, an approximate simulation is performed with C ₁ =C ₂ =500 μF and L ₁ =L ₂ =10 μH. At this time, the resonant frequency f _o1 of the RLC series circuit given by the equation (6) is 2.25 kHz, and the resonant frequency of the RLC series circuit given by the equation (7) is f ₀₂ =2.25 kHz. The corresponding repetition period is 444 μs.

When the primary side drive switch SW1 is turned on and there is a step input, current _first flows through the transmitting side capacitor C1. The power transmission side coil L ₁ originally has the property of preventing a sudden inflow of current. _The voltage of the power transmission side capacitor C1 gradually rises, and the current gradually begins to flow through the power transmission _side coil L1 as well. In the meantime, the charge accumulated in the power transmission _side capacitor C1 also begins to flow out to the power transmission side coil L1 _side . When this happens, the voltage of the power transmission side capacitor C1 drops. Ideally, the time until the primary drive switch SW1 is turned off is set so that the sum of the electrostatic energy (1/2) CV ² and the magnetic energy (1/2) LI ² is maximized. However, since the primary side drive switch SW1 is turned off with the current _I flowing through the power transmission side coil L1, _a back electromotive force is generated in the power transmission side coil L1.

Care must be taken so that the voltage of the back electromotive force generated in the power transmission side coil L1 does not exceed the withstand voltage of the _first semiconductor switching element Q1 used for the primary side drive switch SW1 shown in FIG. The ON/OFF repetition period of the primary side drive switch SW1 is slightly earlier than the peak timing, considering the time until the terminal voltage VC1 of the power transmission side capacitor C1 rises again and reaches the peak.

FIG. 29 shows changes in the voltage of the power transmission _side capacitor C1 as an approximate simulation result. In FIG. 29, as shown in FIGS. 1(b) and 2, a sawtooth transient response waveform is obtained in the change in the voltage of the power transmission _side capacitor C1. With reference to FIG. 30, it will be explained that sawtooth bumps occur because the current in the primary side circuit ₂ is reduced by the magnetic flux generated by the current induced in the power receiving side coil L2. As in the case of the mode using the four switches of the primary side drive switch SW1, the power transmission side switch SW2, the power reception side switch SW3, and the load control switch SW4 as described later in the fourth embodiment, the voltage of the secondary side circuit 3 is 34 corresponds to the phenomenon that the voltage of the primary side circuit 2 becomes zero when is maximized, and the central peak of the W-shaped transient response waveform shown in FIG. 34 becomes a sawtooth wave bump.

The W-shaped transient response waveform shown in FIG. 34 will be described later with reference to FIG. Using the circuit shown in FIG. 31, which is a circuit diagram simplified by omitting the parasitic capacitance and parasitic resistance from the circuit of FIG. A transient response waveform is obtained in which the voltage of the valley on the right side of is raised. The peaks in the middle of the W-shaped transient response waveforms enclosed by the dashed circles A ₁ and A ₂ in FIG. It can be seen that the sawtooth wave appears as a lump.

In the case of the mode with only the primary side drive switch SW1 in the first embodiment, as shown in FIGS. becomes smaller, gradually approaching a sawtooth wave response waveform as shown in FIG. 1(b) and FIG. FIG. 29 is an enlarged diagram showing the transient response waveform in the case of the repetition period of 575 μs shown in FIG. 30(c), which corresponds to the transient response waveforms shown in FIGS. .

In the case of the ON/OFF repetition period of the primary side drive switch SW1 of 565 μs, the W-shaped transient response waveform as shown by the dashed circle Aa in FIG. 30( _a ) is obtained. Since the repetition period obtained from the resonance frequency of the RLC series circuit defined by the formulas (6) and (7) is 444 μs, the repetition period in FIG. is longer than the repetition period required by AC theory. That is, it can be seen that the power transmission device according to the first embodiment oscillates at a repetition period different from the resonance frequency of the RLC series circuit determined by the AC theory.

In the power transmission device according to the first embodiment, the amplitude is reduced by transmitting electrical energy from the primary side circuit 2 to the secondary side circuit 3 by characteristic harmonic transmission. When the repetition period is slightly lengthened to 570 μs in _FIG . Lift up. Furthermore, bulges also begin to appear on the upper side of the sawtooth wave, as shown in FIGS. 1(b) and 2. FIG.

Further transmission of electrical energy from the primary side circuit 2 to the secondary side circuit 3 by characteristic harmonic transmission further reduces the amplitude. When the repetition period is further lengthened to 575 μs in FIG. 30( c ), the voltage at the valley on the right side of the shape of W is further lifted up as indicated by the dashed circle A _c , resulting in a bump-like shape. It becomes a shoulder and the shape of W disappears from the transient response waveform. Then, as shown in FIGS. 1(b) and 2, bumps also appear above the sawtooth wave.

Further transmission of electrical energy to the secondary side circuit 3 by characteristic harmonic transmission further reduces the amplitude, and when the repetition period is further lengthened to 580 μs in FIG. As indicated by box _d , the hump-shaped shoulder that exhibits the transient response waveform is further lifted. 1(b) and 2, the bumps on the upper side of the sawtooth wave also become conspicuous, and two steps of bumps are shown. In this way, in the shape of W showing the transient response waveform, the amplitude gradually decreases, and when the repetition period is lengthened, the voltage at the valley on the right side does not gradually decrease, and the valley on the right side of the shape of W rises, resulting in a two-step bump. It becomes a sawtooth transient response waveform with

When the ON/OFF repetition period of the primary side drive switch SW1 is lengthened, the voltage at the valley on the right side of the W shape rises because the electrical energy received by the secondary side circuit 3 moves to the load element 6 to be charged. it is conceivable that. The maximum value of the current when the primary side drive switch SW1 is turned on and the maximum value of the current flowing through the load element 6 are due to the parasitic inductance of the electric wire that constitutes the actual circuit of the power transmission device according to the first embodiment. Dependent. The parasitic inductance is estimated to be about 1 mH to 3 μH from analysis of waveforms measured in an actual circuit. That is, the sawtooth-shaped transient response waveform having a plurality of bumps shown in FIGS. I understand.

Next, using the resonance frequency f _o1 of the RLC series circuit given by the conventional AC theory (6), ω ₀ =2πf _o1 , ω1=ω ₀ /(1−k) ^1/2 , and FIG. As shown, the energy transfer function f1 is a sigmoid function that decays stepwise after 2π/ω1 seconds, which is the transfer timing:

f1=V3/[1+exp{10 ⁶ (t-2π/ω1)}]+V2 ……(8)

think of. Here, k defining ω1=ω ₀ /(1−k) ^1/2 is the coupling coefficient K _AC of the AC theory used to define the equation (1) (k=K _AC ).

Furthermore, for any decay function f2, a function that reduces with an appropriate decay constant τ:

f2=exp(-τt) ……(9)

think of. For example, in the power transmission device according to the first embodiment, the attenuation function f2 corresponds to the attenuation function of the maximum value of the voltage VC1 across the _primary capacitor C1 due to the time constant τ due to the parasitic resistance and the capacitor.

V1, V2 and V3 shown in FIG. 35 can all correspond to the voltage VC1 across the _primary side capacitor C1. V1 can correspond to the voltage initially charged on the _primary capacitor C1 and V2 to the voltage remaining on the _primary capacitor C1 after the electrical energy has been transferred to the secondary circuit 3. V3 shown in FIG. 35 corresponds to their difference. The energy transfer function f1 is a function intended to cause the voltage VC1 across the _primary side capacitor C1 to drop from V1 to V2.

Using the above ω ₀ = _{2πf 01} , ω2 = ω ₀ /(1+k) ^1/2 and the mutual induction function φ(k) is

φ(k)=cos(ω1t)+cos(ω2t) (10)

, the function (V1/2).phi.(k) shows a change as indicated by the thin broken line curve after the transfer timing of 2.pi./.omega.1 seconds in FIG. However, it should be noted that the coupling coefficient K of the power transmission device according to the first embodiment is a time-dependent parameter and is strictly different from the coupling coefficient K _AC of the AC theory. The thin dashed line in FIG. 35 is the waveform before the current is supplied to the load circuit 6, and corresponds to the waveform of the voltage VC1 across the _primary capacitor C1. A thin dashed line after the transfer timing of 2π/ω1 seconds can be considered as a waveform when the energy of the primary side capacitor C1 is not transferred to the secondary side.

The function (V2/2)φ(k) is the thin dashed curve up to the transfer timing 2π _/ ω1 seconds in FIG. is the waveform of This corresponds to the waveform when it is assumed that the energy of the primary side capacitor C1 has been transferred to the secondary side from the beginning by the amount transferred to the secondary side. It can be seen that the function _f1.f2..phi .(k) indicated by the solid line becomes the voltage VC1 across the capacitor C1, indicating the W type.

As described above, in the power transmission device according to the first embodiment, the amplitude is reduced by transmitting electrical energy to the secondary side circuit 3 by characteristic harmonic transmission. That is, as shown in FIG. 32, the trough on the right side of the W-shaped transient response waveform becomes smaller, gradually rises upward, and does not become depressed. This results in damping due to the dissipation of electrical energy through parasitic resistance, with a sawtooth-like overall RC time constant. An approximate simulation based on AC theory for the circuit shown in FIG. 28 is only an approximation, and there are limits to AC theory, but you should be able to understand the sawtooth-shaped transient response waveform having roughly two bumps. . Actually, only the experimental data shown in FIGS. 1B and 2 can explain the effect of the power transmission device according to the first embodiment.

That is, if the same capacitor is adopted for the power transmission side capacitor C1 and the power reception _side capacitor _C2 , and the same coil is adopted for the power transmission _side coil L1 and the power reception _side coil L2, the parasitic resistance is reduced to the coil and the capacitor. Including, the time constant inherent in the primary side circuit 2 and the time constant inherent in the secondary side circuit 3 can be harmonized.

As described above, the power transmission device according to the first embodiment of the present invention is a technology that does not rely on AC theory using the novel concept of characteristic harmonic transmission, so it is possible to use an inexpensive DC power supply 5. can. Therefore, the power transmission device according to the first embodiment does not require an expensive switching power supply, simplifies the circuit configuration, and minimizes power loss on the control circuit side. In particular, when a power semiconductor switching element is used as the primary side drive switch SW1, only a simple control of ON/OFF control of the power semiconductor switching element is required. Overall power transmission efficiency including (zero-order circuit) loss can be enhanced. In particular, since the circuit configuration is simplified, it is less likely to break and circuit design becomes easier. Also, the limit power of power transmission can be pushed up to a value that surpasses the limit power in the conventional AC theory. In principle, the limit power of power transmission can be increased to infinity, and in principle, the limit distance of power transmission can also be increased to infinity.

As a result, according to the power transmission device according to the first embodiment, the overall configuration of the power transmission device is simplified, the power loss on the control circuit side is minimized, and the weight, size, and efficiency are reduced. Therefore, it is possible to inexpensively manufacture a wireless power transmission device with improved overall power transmission efficiency due to power saving. In addition, since characteristic harmonic transmission can be realized with a repetition period longer than the repetition period required by the conventional AC theory, the frequency may be lower than in the case of multiple resonance in the conventional AC theory. Since a low-frequency circuit design is sufficient, it is easy to increase the voltage on the primary side circuit 2 side, and energy loss due to Joule heat generation can be reduced. A highly efficient power transmission device can be manufactured at low cost. By lowering the parasitic resistance, in principle, a power transmission device can be manufactured in which the power transmission efficiency exceeds 99% and is increased to a value close to 100%.

(Second embodiment)
As shown in FIG. 10A, the power transmission device according to the second embodiment of the present invention has a power transmission side switch SW2 added to the circuit configuration of the power transmission device according to the first embodiment shown in FIG. It is configured as The “power transmission side switch SW2” is also a circuit element that limits the free oscillation of the primary side circuit 2 and realizes a transient current-voltage change in the primary side circuit 2, like the primary side drive switch SW1.

As the primary side drive switch SW1 and power transmission side switch SW2 shown in FIG. A power semiconductor switching element is used. In particular, voltage-driven switching elements such as MISFETs, MISSITs, IGBTs, and MOS-controlled SI thyristors are suitable for the primary side drive switch SW1 and the power transmission side switch SW2 because they reduce power consumption. Based on availability in the market and evaluation of the internal resistance of the semiconductor switching element for electric power, it is currently possible to employ MOSFETs as the primary side drive switch SW1 and the power transmission side switch SW2 of the circuit shown in FIG. 10(b). It is possible.

As already described in the power transmission device according to the first embodiment, a large amount of Joule heat is generated in a high-power power transmission device in which a rechargeable battery for EV is used as the load element 6 . In the power transmission device according to the second embodiment, only two power semiconductor switching elements are used as the primary side drive switch SW1 and the power transmission side switch SW2. It can be designed, and the generation of stray resistance, stray capacitance, and stray inductance can be minimized. In addition, since a simple ON/OFF control of the primary side drive switch SW1 and the power transmission side switch SW2 is all that is required, it is possible to simply increase the voltage of the primary side circuit 2 and suppress the generation of Joule heat.

In the mounting circuit shown in FIG. 10(b), considering the circulating current from the power transmission side coil L ₁ , the first freewheeling diode FWD ₁ is connected between the source and drain of the MOSFET as the first semiconductor switching element Q1. _Two freewheeling diodes FWD2 are connected in parallel as protective elements between the source and drain of the MOSFET as the second semiconductor switching element Q2. As in the circuit shown in FIG. 4( a ), in order to prevent the circulating current from the power transmission side coil L ₁ from circulating to the DC power supply 5 , the power supply side diode D1 is connected between the DC power supply 5 and the first semiconductor switching element Q1. are connected in series between In the mounting circuit shown in FIG. 10(b), the equivalent impedance X _Leq of the load element 6 is also expressed by approximating it with the charging capacity C _s .

The wireless power transmission method according to the first embodiment will be described with reference to the timing diagram of FIG. 11 and the schematic time series diagrams of FIGS. 12(a) to 12(d). However, as in the first embodiment, it is assumed that the coupling coefficient K is equivalent to the coupling coefficient K _AC =0.6 derived from AC theory, and the value of the charging voltage _VCS in the initial state is close to full charge. Assume that the voltage is sufficiently high. First, at the timing shown in FIG. 12(a), the power transmission side switch SW2 is turned off, the primary side drive switch SW1 is turned on, and _an initial voltage is applied to the power transmission side capacitor C1 to store electric charge. In FIG. 12(a), since the first semiconductor switching element Q1 is used for the primary side drive switch SW1, the ON state of the primary side drive switch SW1 is indicated by the ON resistance r _on1 of the first semiconductor switching element Q1. . When the power transmission side switch SW2 is turned off, the _primary side circuit 2 is not formed yet, and as shown in FIG. A power feeding side circuit ₁ is composed of a parallel circuit of a semiconductor switching element Q1 and a _first freewheeling diode (freewheeling diode) FWD1, and a series circuit composed of a power transmission side capacitor C1.

As indicated by the thin dashed line in FIG. ₁₁ , the terminal voltage VC1 of the power transmission side capacitor C1 is charged to _a constant voltage while ringing. Although not shown in FIG. 11, FIG. 12(a) shows that the terminal voltage VC2 of the receiving side capacitor _C has a negative value at this timing. Next, at the timing shown in FIG. 12(b), the primary drive switch SW1 is turned off, and after a certain period of time, the power transmission _side switch SW2 is turned on. Energy is accumulated in the transmitting coil L ₁ via the transmitting coil current I _L1 , and characteristic harmonic transmission between the primary circuit 2 and the secondary circuit 3 occurs. At the timing of FIG. 12(b), since the second semiconductor switching element Q2 is used for the power transmission side switch SW2, the ON state of the power transmission side switch SW2 is indicated by the ON resistance r _on2 of the second semiconductor switching element Q2. . A primary side circuit 2 is formed by turning on the power transmission side switch SW2, and a DC power supply 5, an equivalent internal resistance r1, a _first semiconductor switching element Q1 and a _first freewheeling diode (freewheeling diode) FWD1 are connected in parallel. The feeding side circuit ₁ consisting of the circuit and the transmitting side capacitor C1 disappears.

When the electric energy stored in the power transmission side capacitor C1 moves to the power transmission _side coil L1, the inter _- terminal voltage VC1 indicated by the thin broken line in FIG. ₁₁ takes a negative maximum value and then becomes 0V. The electrical energy transmitted to the power receiving side coil _L2 by characteristic harmonic transmission from the primary side circuit ₂ to the secondary side circuit 3 charges the power receiving side capacitor _C2 with the power receiving side coil current IL2. When charging of the power receiving side capacitor _C2 is started, the terminal voltage VC2 of the power receiving side capacitor C takes a maximum negative value as indicated by the thick dashed line in _FIG . positive values as shown. It reaches its maximum value when the inter _- terminal voltage VC1 becomes 0V. As can be seen from the change of the thick dashed line in FIG. 11, the inter _- terminal voltage VC2 takes a negative maximum value and then becomes a positive value at the time when the inter _- terminal voltage VC1 indicated by the thin dashed line becomes 0V. takes the maximum value.

At the timing shown in FIG. 12(c), as the inter _- terminal voltage VC2 increases, part of the electrical energy accumulated in the power receiving side capacitor C generates the charging current _ICs indicated by the dashed-dotted line in FIG. Then, the charge is stored in the rechargeable battery as the load element 6 . Another part of the electric energy accumulated in the power receiving side capacitor C is returned to the power receiving _side coil _L2 as the power receiving side coil current IL2. At the timing shown in FIG. 12(d), when the charging current IC becomes ₀ , the inter _- terminal voltage _VC2 becomes the same value as the charging voltage VCS.

When the electric energy accumulated in the power receiving side capacitor C flows back to the power receiving side coil L2, characteristic harmonic transmission occurs between the primary side circuit ₂ and the secondary side circuit 3, and a part of the electric energy is transferred to the primary side circuit 2. return. When the electric energy stored in the power receiving capacitor C moves to the load element ₆ and the power receiving coil L2, the power receiving capacitor _C2 is discharged. When the receiving-side capacitor _C2 discharges, the inter-terminal voltage VC2 indicated by the thick dashed line on the right side of _FIG . 11 becomes 0V after taking a negative maximum value. At this time, the inter-terminal voltage VC1 indicated by the thin dashed line on the right side of FIG. ₁₁ assumes a negative maximum value, then increases to a positive value, and reaches a constant value when the power transmission side switch SW2 is turned off. maintained at _In the same manner as described in FIG. It can be seen that the charging status of the battery can be monitored. As can be seen from FIG. 11, the vibration waveform of the voltage VC1 between the terminals of the _primary side circuit ₂ and the vibration waveform of the voltage VC2 between the terminals of the secondary side circuit 3 are not mutually symmetrical.

As already mentioned, "resonance" is a concept that applies to systems in free vibration. On the other hand, in the power transmission device according to the second embodiment of the present invention, the power transmission side switch that limits the free oscillation of the primary side circuit 2 and realizes the transient current-voltage change in the primary side circuit 2 It has SW2 and a primary side drive switch SW1. Therefore, in the power transmission device according to the second embodiment, it is possible to transmit non-sinusoidal transient response characteristics to the secondary side circuit 3 by the new concept of “characteristic harmonic transmission”. . Since the configuration of the control circuit is simple and non-sinusoidal transient response characteristics based on the inexpensive DC power supply 5 can be used, the primary side circuit 2 is caused to generate sinusoidal vibration higher than the commercial frequency as in the conventional case. This eliminates the need for an expensive AC power supply circuit, making it less likely to break and easier to design.

Similar to the circuit shown in FIG. 28, the circuit shown in FIG. 33 is the circuit of the power transmission device according to the second embodiment in which an approximate simulation is performed by the AC theory. It has two switches, a switch SW1 and a power transmission side switch SW2. In FIGS. 33A to 33C, the capacitance of the capacitor and the inductance of the coil of the primary side circuit 2 and the secondary side circuit 3 are set to be the same. That is, C ₁ =C ₂ =500 μF and L ₁ =L ₂ =10 μH.

FIG. 33(a) shows a rechargeable battery that is used as a load circuit 36 by transmitting the electrical energy of the primary side circuit 2 to the secondary side circuit 3 by characteristic harmonic transmission at the voltage E ₀ =36 V of the DC power supply 5. 10 is a circuit of the power transmission device according to the second embodiment when charging voltage V _cs =24V. FIG. 33(b) uses the same voltage E ₀ =36 V of the DC power supply 5 as in FIG. 36 is the charging voltage of the rechargeable battery V _cs =100 V, and the circuit is set so that no current flows through the load circuit 36 . FIG. 33(b) shows a case where the charging voltage V _cs of the rechargeable battery employed as the load circuit 36 is set to 100 V, and the load circuit 36 is set so that no current flows, as in FIG. 33(b). However, this is the case where the voltage E ₀ of the DC power supply 5 is set to 26V by subtracting the amount transmitted to the secondary side circuit 3 in advance.

FIG. 34 shows changes in the voltage of the power transmission side capacitor C1 as an approximate simulation result. Similar to that shown in FIG. 32, in the approximate simulation based on the AC theory of the power transmission device according to the second embodiment, the change in the voltage of the power transmission side capacitor C1 is similar to that shown in FIG. A transient response waveform is shown. The solid line in FIG. 34 is the approximate simulation result for the circuit shown in FIG. 33(a), the dashed line in FIG. 34 is the approximate simulation result for the circuit shown in FIG. 33(c) is an approximate simulation result for the circuit shown in FIG.

At first, as indicated by the dashed line in FIG. 34, an attempt is made to cause a current to flow through the load circuit 36. However, since the charging voltage of the rechargeable battery is set high to V _cs =100 V, the current does not flow through the load circuit 36. It becomes a change like the curve shown by the dashed line. The timing at which the current is to flow through the load circuit 36 is presumed to be around the peak position in the center of the W-shaped transient response waveform in FIG.

As described above, according to the power transmission device according to the second embodiment, similarly to the power transmission device according to the first embodiment, it is possible to use the DC power supply 5 that has a simple control circuit and peripheral circuits and is inexpensive. Therefore, an expensive switching power supply is unnecessary. The circuit configuration of the power transmission device according to the second embodiment is simplified, the power loss on the control circuit side is minimized, the device is less likely to break, and circuit design is facilitated. As a result, the overall configuration of the power transmission device has been simplified, making it possible to make it lighter, smaller, and more efficient. A transmission device can be manufactured at low cost. As described in the power transmission device according to the first embodiment, the limit power of power transmission is raised to a value that surpasses the limit power in the conventional alternating current theory, and is pushed to infinity in principle, and power transmission can be extended infinitely in principle. Furthermore, it is possible in principle to increase the power transmission efficiency to a value close to 100%.

(Third embodiment)
As shown in FIG. 13A, the power transmission device according to the third embodiment of the present invention has a configuration in which a power receiving side switch SW3 is added to the power transmission device according to the second embodiment. Like the power transmission side switch SW2 and the primary side drive switch SW1, the "receiving side switch SW3" also limits the free oscillation of the secondary side circuit 3 and realizes transient current-voltage changes in the secondary side circuit 3. It is a circuit element that allows

As the primary side drive switch SW1, the power transmission side switch SW2, and the power reception side switch SW3 shown in FIG. Power semiconductor switching elements including thyristors such as thyristors and SI thyristors are used. Due to the requirement for low internal resistance and availability in the market, it is an industrial practice to employ MOSFETs as the primary side drive switch SW1, the power transmission side switch SW2, and the power reception side switch SW3 in the implementation circuit shown in FIG. 13(b), respectively. considered to be superior.

As described in the power transmission devices according to the first and second embodiments, a large amount of Joule heat is generated in the high-power power transmission device. In the power transmission device according to the third embodiment, only three power semiconductor switching elements are used as the primary side drive switch SW1, the power transmission side switch SW2, and the power reception side switch SW3. The cooling structure can be easily designed, and the generation of stray resistance, stray capacitance, and stray inductance can be minimized. In addition, since a simple ON/OFF control of the primary side drive switch SW1 and the power transmission side switch SW2 is all that is required, it is possible to simply increase the voltage of the primary side circuit 2 and suppress the generation of Joule heat.

In the implementation circuit shown in FIG. 13B, the first freewheeling diode FWD ₁ is between the source and drain of the MOSFET as the first semiconductor switching element Q1, and the second freewheeling diode FWD ₂ is the second semiconductor switching element. A third free wheel diode FWD ₃ is connected in parallel between the source and the drain of the MOSFET as the element Q2 and between the source and the drain of the MOSFET as the third semiconductor switching element Q3 as a protective element. As shown in FIG. 13(b), the third freewheeling diode FWD ₃ is provided in the direction in which the freewheeling current from the receiving side coil L ₂ flows, so that the second freewheeling diode FWD ₂ is placed in the opposite direction. is provided. 4(a) and 10(b), in order to prevent the circulating current from the power transmission _side coil L1 from circulating to the DC power supply 5, the power supply side diode D1 is connected to the DC power supply 5 and the second power supply. It is connected in series between one semiconductor switching element Q1. In the mounting circuit shown in FIG. 13(b), the equivalent impedance X _Leq of the load element 6 is also expressed by approximating it with the charge capacity C _s .

A wireless power transmission method according to the first embodiment will be described with reference to the flowchart shown in FIG. 14 and the timing chart shown in FIG. However, as in the first and second embodiments, characteristic harmonic transmission is assumed under conditions corresponding to the coupling coefficient K _AC =0.6 according to AC theory, and the value of the charging voltage VC _S in the initial state is full. Assume that the voltage is high enough to be close to charging.

First, in step S31 of the flowchart of FIG. 14, the power transmission side switch SW2 and the power reception side switch SW3 are turned off, and only the primary side drive switch SW1 is turned on. As indicated by the thin dashed line in FIG. ₁₅ , the terminal voltage VC1 of the power transmission side capacitor C1 is charged to _a constant voltage while ringing. Although not shown in FIG. 15, the terminal-to-terminal voltage VC2 of the power receiving side capacitor _C is a negative value at this timing. After an initial voltage is applied to the transmission _- side capacitor C1 to store electric charge, the primary-side drive switch SW1 is turned off as shown in FIG. As mentioned above, it is assumed that the charging voltage _VCS is high at this point.

As shown in FIG. 15, after the primary drive switch SW1 is turned off, the power transmission side switch SW2 and the power reception side switch SW3 are simultaneously turned on in step S32 after a certain period of time. When the power transmission side switch SW2 is turned on, the electric energy stored in the power transmission _side capacitor C1 is accumulated in the power transmission _side coil L1 via the power transmission side coil current, and furthermore, the primary side circuit 2 and the secondary side circuit. 3 results in characteristic harmonic transmission. When the electric energy stored in the power transmission side capacitor C1 moves to the power transmission _side coil L1, the inter _- terminal voltage VC1 indicated by the thin broken line in FIG. ₁₅ takes a negative maximum value and then becomes 0V. The electrical energy transmitted to the power receiving side coil L2 by the characteristic harmonic transmission from the primary side circuit ₂ to the secondary side circuit 3 is transferred to the power receiving side capacitor _C2 by the power receiving side coil current because the power receiving side switch SW3 is in the ON state. to charge. When the charging of the power receiving side capacitor _C2 starts, the voltage VC2 across the terminals of the power receiving side capacitor C starts to increase from a negative maximum value as indicated by the thick dashed line in _FIG . It will be a positive value as shown in the more position. While the terminal voltage _{VC2 takes} a negative value, the charging current _ICS does not flow. _ICS begins to rise.

At the timing when the charging current _ICs starts to rise, the power transmitting side switch SW2 and the power receiving side switch SW3 are turned off in step S33. In the OFF state of the power transmission side switch SW2 and the power reception side switch SW3, the terminal voltage VC2 indicated by the thick broken line in _FIG . This is also the time when the inter _- terminal voltage VC1 indicated by the thin dashed line becomes 0V. The charging current _ICs indicated by the dashed line in the center of FIG. 15 increases even after the power transmission side switch SW2 and the power reception side switch SW3 are turned off, reaches a peak value, then decreases, and becomes zero in step S34. Become.

As shown in the right side of the center of _FIG . 15, the maximum value of the terminal voltage _VC2 indicated by the thick dashed line becomes a slightly lower constant value when the charging current IC starts to decrease. It becomes a shoulder-shaped waveform. Even after the charging current IC becomes zero, the terminal voltage _VC2 indicated by the thick broken line in _FIG . 15 maintains a value lower than the maximum value when the power transmission side switch SW2 is turned off. After the power transmission _side switch SW2 is turned off, the maximum value of the terminal voltage _{VC2 decreases} after a certain period of _time has passed. The amount of decrease in the maximum value of is small, and the influence on the characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 is small.

In step S35 after the charging current _I.sub.C reaches 0 A, the power transmitting side switch SW2 and the power receiving side switch SW3 are turned on again at the same time. occurs. Due to the ON state of the power transmission side switch SW2 and the power reception side switch SW3 in step S35, the inter _- terminal voltage VC2 indicated by the thick dashed line on the right side of FIG. . At this time, the inter-terminal voltage VC1 indicated by a thin dashed line on the right side of FIG. ₁₅ also starts to decrease, reaches a maximum negative value, and then increases to a positive value.

Next, in step S36, the power transmitting side switch SW2 and the power receiving side switch SW3 are turned off when the terminal voltage _VC1 becomes maximum and the terminal voltage _VC2 becomes 0V. As shown in _FIG . ₁₅ , the terminal voltage VC1 indicated by the thin dashed line at step S36 is the same value as the terminal voltage VC2 indicated by the thick dashed line at step S34. After the switch SW3 is turned off, the constant value is maintained. be. Although not shown, the terminal voltage VC1 and the terminal voltage of the load element ₆ have the same value at step S36. For this _reason , in the same way as described in FIG. It can be seen that the charging status of the rechargeable battery can be monitored. Comparing the timing chart of the power transmission device according to the second embodiment shown in FIG. 11 with the timing chart of the power transmission device according to the third embodiment shown in FIG. Also, waveforms showing temporal changes (transient responses) such as terminal voltage VC1 and terminal voltage VC2 in characteristic harmonic transmission between _primary side circuit ₂ and secondary side circuit 3 are almost the same. I understand.

"Resonance" is a concept used in an AC circuit that vibrates freely. It has a primary side drive switch SW1, a power transmission side switch SW2, and a power reception side switch SW3 that realize a transient current-voltage change in the primary side circuit 2 and the secondary side circuit 3. FIG. Therefore, in the power transmission device according to the third embodiment, the transient response characteristics of the primary side circuit 2 can be transmitted to the secondary side circuit 3 by the new concept of "characteristic harmonic transmission." be. The configuration of the control circuit is simple, and electrical energy can be transmitted using a non-sinusoidal transient response characteristic based on the inexpensive DC power supply 5. Expensive AC power circuits that generate vibrations are no longer required.

Therefore, according to the power transmission device according to the third embodiment of the present invention, similarly to the power transmission devices according to the first and second embodiments, the direct-current power supply 5 having a simple control circuit and peripheral circuits and being inexpensive is provided. Since it can be used, an expensive switching power supply is not required, the circuit configuration is simplified, and the power loss on the control circuit side is also minimized. As a result, according to the power transmission device according to the third embodiment, the overall configuration of the power transmission device can be simplified, the weight and size can be reduced, and the efficiency can be improved. It is possible to inexpensively manufacture a wireless power transmission device with improved overall power transmission efficiency including As in the power transmission devices according to the first and second embodiments, the power transmission device according to the third embodiment simplifies the circuit configuration, making it difficult to break and easy to design the circuit. . In addition, it is possible to raise the limit power of power transmission to infinity in principle, extend the limit distance of power transmission to infinity in principle, and increase the power transmission efficiency to a value close to 100% in principle. is.

(Fourth embodiment)
As shown in FIG. 16A, the power transmission device according to the fourth embodiment of the present invention has a configuration in which a load control switch SW4 is added to the power transmission device according to the third embodiment. The “load control switch SW4” is a circuit element that limits the free oscillation of the secondary circuit 3 and realizes a transient current-voltage change in the secondary circuit 3, like the power receiving switch SW3.

As the primary side drive switch SW1, the power transmission side switch SW2, the power reception side switch SW3, and the load control switch SW4 shown in FIG. , BJTs, power semiconductor switching elements including thyristors such as GTO thyristors and SI thyristors can be used. Considering the requirement for low internal resistance, current market availability allows the MOSFET to be the primary side drive switch SW1, the power transmission side switch SW2, the power reception side switch SW3, and the load switch of the implementation circuit shown in FIG. 16(b). It is preferable to employ each of them as the control switch SW4.

In the power transmission device according to the fourth embodiment, only four power semiconductor switching elements may be used as the primary side drive switch SW1, the power transmission side switch SW2, the power reception side switch SW3, and the load control switch SW4. It is possible to easily design a cooling structure that prevents the generation of stray resistance, stray capacitance, and stray inductance. In addition, since simple ON/OFF control of the primary side drive switch SW1 and the power transmission side switch SW2 is sufficient, it is possible to increase the voltage of the primary side circuit 2 to suppress the generation of Joule heat.

In the implementation circuit shown in FIG. 16(b), the first freewheeling diode FWD ₁ is between the source and drain of the MOSFET as the first semiconductor switching element Q1, and the second freewheeling diode FWD ₂ is the second semiconductor switching element. Between the source and drain of the MOSFET as the element Q2, the _third freewheeling diode FWD3 is connected between the source and the drain of the MOSFET as the third semiconductor switching element Q3, and the fourth freewheeling diode FWD4 is the _fourth semiconductor switching element. They are connected in parallel as protection elements between the source and drain of the MOSFET as the element Q4. As shown in FIG. 16(b), the third freewheeling diode FWD ₃ is provided in the direction in which the freewheeling current from the receiving side coil L ₂ flows, so that the second freewheeling diode FWD ₂ is placed in the opposite direction. The arrangement is the same as in FIG. 13(b). 4(a), 10(b) and 13(b), in order to prevent the circulating current from the power transmission _side coil L1 from circulating to the DC power supply 5, the power supply side diode D1 are connected in series between the DC power supply 5 and the first semiconductor switching element Q1. In the mounting circuit shown in FIG. 16(b), the equivalent impedance X _Leq of the load element 6 is also expressed by approximating it with the charging capacity C _s .

As already described, when comparing the timing chart of the power transmission device according to the second embodiment shown in FIG. 11 and the timing chart of the power transmission device according to the third embodiment shown in FIG. , the waveforms showing temporal changes (transient response) such as terminal voltage VC1 and terminal voltage VC2 in the characteristic harmonic transmission between the _primary side circuit ₂ and the secondary side circuit 3 are almost are the same. As shown in FIG. 16(a), even if the load control switch SW4 is added to the power transmission device according to the third embodiment, the essence of the characteristic harmonic transmission does not change. Waveforms showing temporal changes (transient responses) of terminal voltage VC1, terminal voltage VC2, etc. in characteristic harmonic transmission between _primary side circuit ₂ and secondary side circuit 3 are almost the same.

However, as shown in FIG. 16(a), in a configuration having four switches, that is, the primary side drive switch SW1, the power transmission side switch SW2, the power reception side switch SW3, and the load control switch SW4, the primary side drive switch SW1 and the load control switch At the timing when the switch SW4 is turned off and the power transmission side switch SW2 and the power reception side switch SW3 are turned on, the circuit on the DC power supply 5 side and the circuit on the load element 6 side are connected to the primary side circuit 2 and the secondary side circuit 3, respectively. Since they are separated, the primary side circuit 2 and the secondary side circuit 3 can oscillate freely. That is, since the LC resonant circuit of the primary side circuit 2 and the LC resonant circuit of the secondary side circuit 3 can be treated as a circuit coupled by mutual inductance M, the idea of multiple resonance in AC theory can be adopted. That is, at the timing when the primary-side drive switch SW1 and the load control switch SW4 are in the disconnected state, and the power-transmitting-side switch SW2 and the power-receiving-side switch SW3 are in the conductive state, the above-described coupled equations (2) and (3) give: The efficiency of characteristic harmonic transmission can be studied.

However, in the implementation circuit, it is necessary to consider the on-resistance of the power semiconductor switching elements used in the four switches of the primary side drive switch SW1, the power transmission side switch SW2, the power reception side switch SW3, and the load control switch SW4. Therefore, it cannot be described by the coupled equations of formulas (2) and (3). Therefore, when the primary side drive switch SW1 and the load control switch SW4 are cut off and the power transmission side switch SW2 and the power receiving side switch SW3 are turned on, the LCR resonance circuit of the primary side circuit 2 and the secondary side circuit 3 It is necessary to consider the circuit in which the LCR resonant circuit is coupled with the mutual inductance M.

In addition, since it is necessary to consider energy transmission in a transient response such as a step response when the primary side drive switch SW1 and the load control switch SW4 are made conductive, the power transmission device according to the fourth embodiment of the present invention is Not everything can be interpreted by the conventional exchange theory. That is, in the region of free vibration already shown by hatching in FIG. In the operating environment of the power transmission device according to the fourth embodiment, in which the four switches of the control switch SW4 are used to change the boundary conditions of the circuit from moment to moment, a sawtooth-shaped rise as illustrated in FIG. It is necessary to analyze including transient response such as characteristics.

(Simulation of characteristic harmonic transmission waveform)
The waveforms of the terminal voltage VC ₁ and the terminal voltage VC ₂ of the power transmission side capacitor C ₁ and the power reception side capacitor C ₂ in the configuration illustrated in FIG. 2 and the secondary side circuit 3 to confirm the characteristic harmonic transmission waveform. The coupling coefficient K _AC between the primary side circuit 2 and the secondary side circuit 3 is 0.6, the capacitance of the power transmission side capacitor C ₁ and the power reception side capacitor C ₂ are both 65 μF, the power transmission side coil L ₁ and the power reception side coil L ₂ Both inductances are assumed to be 60 μH.

First, the power transmission side switch SW2, the power reception side switch SW3, and the load control switch SW4 are turned off, the _primary side drive switch SW1 is turned on, and an initial voltage of 20 V is applied to the power transmission side capacitor C1 to turn off the power transmission _side capacitor C1. stores an electric charge in Next, when the primary side drive switch SW1 is turned off and the power transmitting side switch SW2 and the power receiving side switch SW3 are turned on, it can be expected that characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 will occur. _FIG . 17A shows the waveforms of the terminal voltage _VC1 and the terminal voltage VC2 obtained by simulation. _Both the waveform of the terminal voltage VC1 and the waveform of the terminal voltage VC2 are waveforms in which _a large-amplitude sine wave and a small-amplitude sine wave are synthesized, and are different from the sine wave in normal AC theory. .

In FIG. 17A, the power transmission side switch SW2 and the power reception side switch SW3 are turned on at 0.2 ms. _{At 0.45ms} , the voltage across terminal VC1 becomes 0V and the voltage across terminal VC2 becomes 20V. This indicates that all the energy on the power transmission side is transmitted to the power reception side. transmission can take place.

(Measurement of characteristic harmonic transmission waveform by mounting circuit)
Subsequently, the waveform of the characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 is measured using the mounted circuit having the configuration illustrated in FIG. 16(a). The equivalent coupling coefficient K of the primary side circuit 1 and the secondary side circuit 2 is 0.6, the capacitance of the power transmission side capacitor C ₁ and the power reception side capacitor C ₂ are both 65 μF, the power transmission side coil L ₁ and the power reception side coil L ₂ Assume that the inductance is 60 μH and the initial voltage applied to the transmission _side capacitor C1 is 20V. These are the same values as in the simulation. _FIG . 17B shows the waveforms of the terminal voltage _VC1 and the terminal voltage VC2 of the power transmission _side capacitor C1 and the power receiving _side capacitor _C2 obtained by measurement. It can be seen that there is no symmetry between the waveforms of the terminal voltage _VC2 .

Since there is a parasitic resistance in the mounted circuit, the waveform attenuates with time, unlike the result of the simulation based on the normal AC theory shown in FIG. 17(a). In FIG. 17(b), transmission starts at 0.2 ms, and the inter _- terminal voltage VC2 reaches the maximum of 15 V at 0.45 ms. At this time, the terminal voltage VC1 is -3V, not _all the energy on the power transmission side is transmitted to the power receiving side, and some energy remains on the power transmission side, but the energy on the power transmission side is the power receiving side. It can be confirmed that it is transmitted to the side. As described above, even if the load control switch SW4 is added to the power transmission device according to the _third embodiment, temporal changes ₍ transient response ) are almost identical. That is, the waveform of the terminal voltage _VC1 and the waveform of the terminal voltage VC2 shown in _FIG . 17(b) are diagrams showing macro changes, and in the macro, a sine wave with a large amplitude and a sine wave with a small amplitude are synthesized. Although the waveform looks like a sine wave, it is similar to the waveforms shown in FIGS. 11 and 15 when the time axis is lengthened and viewed in detail, and does not indicate the change of a sine wave.

(Change in equivalent coupling coefficient and change in characteristic harmonic transmission)
In the configuration illustrated in FIG. 16(a), after charging the power transmission side capacitor C1, the _primary side drive switch SW1 and the load control switch SW4 are turned off, and the power transmission side switch SW2 and the power reception side switch SW3 are turned on. Characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 occurs when . According to conventional AC theory, this operation is represented by the coupling equations of equations (2) and (3) already mentioned.

In the conventional AC theory, the voltage VC2 across the terminals of the receiving side capacitor _C2 generated by the characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 by solving the coupled equations of formulas ( ₂ ) and (3) is obtained, using the mutual induction function φ(k) defined by equation (10),

VC2 ₌ VC0/ ₂ x (L2 _/ L1) ( _1/2 ⁾ x φ(k) (11)

becomes. Here, VC ₀ is the initial voltage across the terminals of the power transmitting capacitor C ₁ , ω ₀ is the resonance angular frequency, and ω ₀ =L ₁ ×C ₁ =L ₂ ×C ₂ . When the voltage VC ₁ between the terminals of the capacitor on the power transmission side is 0, the formula (11) becomes the maximum value VC ₀ ×(L ₂ /L ₁ ) ^(1/2) . All the energy has been transferred to the receiving side.

In the configuration illustrated in FIG. 16(a), the change in the waveform of the characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 when the coupling coefficient K _AC according to the normal AC theory is changed is Obtained by simulation based on AC theory. The capacitance of the power transmission side capacitor C1 and the power reception _side capacitor _C2 are both 500 μF, the inductance of the power transmission _side coil L1 and the power reception side coil L2 are _{both 10} μH, and the initial voltage applied to the power transmission side capacitor C1 is 25 V. . In the following explanation, it is approximated that the coupling coefficient K _AC according to AC theory is equal to the equivalent coupling coefficient, and the equivalent coupling coefficient K is set to 0.00, 0.1, 0.6, 0.8, 0.88, and normal A simulation based on AC theory was performed. FIG. 18(a) to FIG. 19(c) show the waveforms of the terminal _- to-terminal voltage VC1 and the terminal-to _- terminal voltage VC2 obtained as a result of simulation based on the ordinary AC theory. As shown in FIG. 18(a), when the coupling coefficient K=0.00 according to the normal AC theory, the primary side circuit 1 and the secondary side circuit do not interact with each other, and the primary side circuit 2 and the secondary side circuit No characteristic harmonic transmission between 3 occurs.

When the equivalent coupling coefficient K=0.1, 0.6, 0.8 and 0.88, characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 occurs. 2.2 ms when the equivalent coupling coefficient K=0.1 as shown in FIG. 18(b), 0.28 ms when the equivalent coupling coefficient K=0.6 as shown in FIG. As shown in 19(c), when the equivalent coupling coefficient K is 0.88, the voltage VC1 across the terminals of the capacitor on the power transmission side becomes 0 V in 0.3 ms, and the voltage VC2 across the terminals of the capacitor on the power reception _side becomes ₀ V on the power transmission side. It has the same _{value as the initial voltage VC 0 across} the terminals of the capacitor C1, and all the energy on the transmitting side is transferred to the receiving side. As shown in FIG. 19(b), when the equivalent coupling coefficient K ₌ 0.8, the maximum value of the voltage VC2 between the terminals of the receiving side capacitor is smaller _than the initial voltage VC0 between the terminals of the transmitting _side capacitor C1. , and power transmission cannot be efficiently performed when the equivalent coupling coefficient K is 0.8 compared to when the equivalent coupling coefficient K is 0.1, 0.6, and 0.88.

Also, compared to the equivalent coupling coefficients K = 0.6 and 0.88, the time required for the terminal voltage VC2 of the receiving _side capacitor to reach its maximum value when the equivalent coupling coefficient K = 0.1 is is long. Equation (11) is expressed as the sum of two modes,

(1+k) ^(1/2) /(1-k) ^(1/2) =2 ……(12)

, that is, when the equivalent coupling coefficient K=0.6, the time until the voltage VC2 across the terminals of the receiving _side capacitor _C2 takes the maximum value is the shortest, and the next shortest is

(1+k) ^(1/2) /(1-k) ^(1/2) = 4 ……(13)

, that is, when the equivalent coupling coefficient K=0.88. In the implementation circuit, the waveform is attenuated with time due to the parasitic resistance r _L =R _str (L ₁ )=R _str (L ₂ ) of the coil and the parasitic resistance r _C of the capacitor. At 88, power transmission can be performed most efficiently. Also, when the parasitic resistances r _L and r _C are low, power can be efficiently transmitted even when the equivalent coupling coefficient K=0.1. As the distance d increases, the equivalent coupling coefficient K decreases. Therefore, if the parasitic resistances r _L and r _C are sufficiently low, it can be said that they can be sent over a long distance.

_FIG . 20( _a ) shows enlarged waveforms of the terminal voltage VC1 and the terminal voltage VC2 when the equivalent coupling coefficient K=0.6 shown in FIG. 19(a). _FIG . 20(b) shows the waveforms of the currents I1 and _I2 of the power transmission _side coil L1 and the power reception _side coil L2 at this time. _At 0.28 ms, the voltage VC1 across the terminals becomes _0V , and the voltage across the terminals VC2 becomes the same value as the initial voltage _VC0 across the terminals of the transmission _side capacitor C1, and at the same time, the currents _I1 and _I2 become 0A. ing. When the power transmitting _side switch SW2 and the power receiving _side switch SW3 _are turned off at 0.28 ms, the power can be transmitted with maximum efficiency. Since the back electromotive force generated in the power receiving side coil L2 becomes ₀ , it is possible to prevent damage to the power transmitting side switch SW2 and the power receiving side switch SW3.

(Optimal combination of inductance L and capacitance C)
In the configuration illustrated in FIG. 16A, the transmission efficiency is the highest when transmitting the energy stored on the power transmission side to the power reception side by characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3. A combination of the inductance L of the coil and the capacitance C of the capacitor is obtained by the following procedure. The transmission efficiency P is defined by P _on C _e-tr for the energy to be transmitted at one time, P _on C _e- loss for the energy lost at one time due to the parasitic resistance r _L of the coil and the parasitic resistance r _C of the capacitor. Assuming that the time required for the cycle is τ _on C _e , it is represented by Equation (14).

P=( _{PonCe -tr} - _{PonCe -loss} )/ _{τonCe (} 14)

The energy P _on C _e-tr to be transmitted at one time is 1/2×CV ₀ ² , and the energy P _on C _e-loss lost at one time is (r _L +r _C )/2×C/L×V. ₀ ² , the time τ _on C _e required for one transmission is 2(1.6) ^1/2 π(LC) ^1/2 , so the transmission efficiency P is expressed by Equation (15).

P=(1/2×CV ₀ ² −(r _L +r _C )/2×C/L×V ₀ ² )/(2(1.6) ^1/2 π(LC) ^1/2 ) ( 15)

Here, V ₀ is the initial voltage applied to the power transmission side capacitor C ₁ .

If K _L =r _L /L and K _C =r _C ×C for the parasitic resistance r _L of the coil and the parasitic resistance r _C of the capacitor, and let I _max be the maximum current flowing through the coil, then I _max =(C/L ) ^1/2 × V, so

P=I _max V {(1.6) ^−1/2 π ⁻¹ −(K _L (LC) ^1/2 +K _C (LC) ^−1/2 )}
…………(16)

becomes,

K _L (LC) ^1/2 + K _C (LC) ^−1/2 >= 2(K _L K _C ) ^1/2 (17)

is.

When the transmission efficiency P is maximized

K _L (LC) ^1/2 + K _C (LC) ^−1/2 = 2(K _L K _C ) ^1/2 (18)

and at this time

LC= _KC / _KL (19)

becomes. The transmission efficiency is maximized when the inductance L of the coil and the capacitance C of the capacitor satisfy the expression (19).

In the configuration exemplified in FIG. 16A, the transmission efficiency when changing the inductance of the coil and the capacitance of the capacitor is obtained by simulation based on ordinary AC theory. Assume that V=36 V, coupling coefficient K _L =600/H, and coupling coefficient K _C =3.00×10 ⁻⁶ ΩF. FIG. 21 shows changes in transmission efficiency with respect to the capacity of the capacitor when the inductance of the coil is 1, 2, 5, 10, 20 and 50 μH, obtained as a result of a simulation based on ordinary alternating current theory. As shown in FIG. 21, the combination of the coil inductance L and the capacitor capacity C that maximizes the transmission efficiency is They are 5000, 2500, 1000, 500, 250, and 100 μF, and satisfy equation (19).

(Power transmission when the voltage between the terminals of the load element is low)
See the flowchart shown in FIG. 23 and the timing chart shown in FIG. and explain. However, when the characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 occurs, such as the equivalent coupling coefficient K equivalent to the coupling coefficient K _AC = 0.6, 0.88 defined in the AC theory , the maximum value of the voltage across the terminals _VC2 is the same as the initial voltage _VC0 across the terminals of the transmitting capacitor C1, and the equivalent coupling _coefficient K is adjusted so that the voltage across the terminals _VC1 is 0V at that time. shall be

First, in step S11, the power transmission side switch SW2, the power reception side switch SW3, and the load control switch SW4 are turned off, and the primary side drive switch SW1 is turned on. After an initial voltage is applied to the transmission _- side capacitor C1 to store electric charge, the primary-side drive switch SW1 is turned off. It is assumed that the voltage across the terminals of the load element 6 is sufficiently low at this point. Next, in step S12, when the power transmitting side switch SW2 and the power receiving side switch SW3 are turned on, characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 occurs. Next, in step S13, the absolute value of the terminal voltage VC2 becomes maximum due to characteristic harmonic transmission between the primary side circuit ₂ and the secondary side circuit 3, and when the terminal voltage VC1 becomes ₀ V, the power transmission side The switch SW2 and the power receiving side switch SW3 are turned off.

Next, in step S14, when the load control switch SW4 is turned on, the charging current _ICS is generated and the inter _- terminal voltage VC2 is reduced. Next, in step S15, the load control switch SW4 is turned off when the charging current _ICS becomes zero. At this time, the terminal voltage VC2 and the terminal voltage of the load element ₆ have the same value. If the voltage across the terminals of the load element 6 is sufficiently low at step S11, the voltage VC2 across the terminals of the power receiving side capacitor at step S15 is ₀ V, or sufficiently low to the extent that it can be regarded as 0 V, and the power receiving side capacitor _C2 is discharged. Without doing so, the process can return to step S11.

(Power transmission method when the voltage between the terminals of the load element is not low)
The second wireless power transmission method according to the fourth embodiment when the voltage across the terminals of the load element 6 is not low will be described with reference to the flowchart shown in FIG. 25 and the timing chart shown in FIG. 24(b). . However, it is assumed that the equivalent coupling coefficient K is adjusted in the same manner as when the voltage across the terminals of the load element 6 is low.

Steps S21 to S24 are the same as steps S11 to S14. In step S25, the load control switch SW4 is turned off when the charging current _ICS becomes zero. The inter _- terminal voltage _VC2 at this time becomes the same value as the charging voltage VCS. In the next step S26, the power receiving side capacitor is discharged.

In step S26, when the power transmission side switch SW2 and the power reception side switch SW3 are turned on, characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 occurs again. In step S27, the absolute value of the terminal voltage VC1 becomes maximum due to characteristic harmonic transmission between the _primary side circuit ₂ and the secondary side circuit 3, and when the terminal voltage VC2 becomes 0 V, the power transmission side switch SW2 and The power receiving side switch SW3 is turned off. Since the terminal voltage _VC1 at the time of step S27 has the same value as the terminal voltage VC2 at the time of step S25, the terminal voltage _VC1 and the charging voltage _VCS have the same value at the _time of step S27. Therefore, in this case, the charging voltage VC _S can be monitored with the voltage VC ₁ between the terminals.

As described above, according to the power transmission device according to the fourth embodiment of the present invention, like the power transmission devices according to the first to third embodiments, the control circuit and peripheral circuits are simple and inexpensive. Since the DC power supply 5 can be used, an expensive switching power supply is not required, the circuit configuration is simplified, the power loss on the control circuit side is minimized, the circuit is less likely to break, and circuit design is facilitated. As a result, according to the power transmission device according to the fourth embodiment, the overall configuration of the power transmission device is simplified, making it possible to reduce the weight, size, and efficiency, and reduce the loss of the power supply circuit (zero-order circuit). In principle, the overall power transmission efficiency including power transmission efficiency is increased to a value close to 100%, the limit power of power transmission is pushed up to infinity in principle, and the limit distance of power transmission is extended to infinity in principle. Therefore, the wireless power transmission device can be manufactured at low cost.

(Fifth embodiment)
In the power transmission device according to the fifth embodiment of the present invention, as shown in FIG. 26, the primary side circuit 2C and the secondary side circuit 3C have a first mutual coupling capacitor _C23 and a second mutual coupling capacitor C23. It is electrostatically coupled at C ₂₄ , and has a configuration in which the coil of the power transmission device according to the first embodiment is replaced with a capacitor, and the capacitor is replaced with a coil. According to the law of electromagnetic induction and Maxwell's equations, such replacement of coils and capacitors is possible. That is, as shown in FIG. 26, the power transmission device according to the fifth embodiment has a power transmission side capacitor for accumulating electrostatic energy, similar to the power transmission device according to the first embodiment shown in FIG. C ₂₁ has a power transmission side coil L ₂₁ that is connected in parallel with the power transmission side capacitor C ₂₁ and accumulates the electrostatic energy sent from the power transmission side capacitor C ₂₁ as magnetic energy and circulates this magnetic energy to the power transmission side capacitor C ₂₁ . A primary side circuit 2 is provided.

Then, as shown in FIG. 26, the power transmission device according to the fifth embodiment has a first electrode in which one electrode is connected to one node that connects the power transmission side capacitor _C21 and the power transmission side coil _L21 in parallel. 1 further includes a mutual coupling capacitor _C23 and a second mutual coupling capacitor _C24 having one electrode connected to the other node connecting the power transmission side capacitor _C21 and the power transmission side coil _L21 in parallel. It is different from the power transmission device according to the first embodiment shown. In the power transmission device according to the fifth embodiment, one electrode is connected to the other electrode of the first mutual coupling capacitor _C23 , and the other electrode is connected to the other electrode of the second mutual coupling capacitor _C24 . A power receiving side capacitor _C22 connected in parallel to receive electrostatic energy from the primary side circuit 2, and a power receiving side coil L that is connected in parallel to the power receiving side capacitor _C22 and accumulates the electrostatic energy accumulated in the power receiving side capacitor _C22 as magnetic energy. It further comprises a secondary side circuit 3 with ₂₂ .

Furthermore, as shown in FIG. 26, the power transmission device according to the fifth embodiment includes a DC power supply 5 that constitutes a circuit that connects between one terminal and the other terminal of the power transmission side coil L ₂₁ , and a power transmission side coil A primary side drive switch SW1 is connected in series between one terminal of _L21 and the DC power supply 5, and step-inputs an intermittent DC voltage to the power transmission side coil _L21 . Also, a circuit connecting between one terminal and the other terminal of the power receiving side coil _L22 is configured, and a load element 6 that receives magnetic energy from the power receiving side coil _L22 and an anode of one side of the power receiving side coil _L22 . A load-side diode D2 having a cathode connected to the load element 6 is provided on the terminal side. Even with electrostatic coupling as shown in FIG. 26, the power transmission device according to the fifth embodiment can transmit electrical energy from the primary side circuit 2 to the secondary side circuit 3 without contact. . The waveforms of the currents flowing through the power transmitting side coil _L21 and the power receiving side coil _L22 are obtained by simulation based on ordinary alternating current theory, and the characteristic harmonic transmission waveforms between the primary side circuit 2 and the secondary side circuit 3 are confirmed.

The equivalent coupling coefficient of the primary side circuit 21 and the secondary side circuit 22 is 0, the inductance of the power transmission side coil L ₂₁ and the power reception side coil L ₂₂ is 0.1 μH, and the capacitance of the power transmission side capacitor C ₂₁ and the power reception side capacitor C ₂₂ are both 400 pF, and the capacitances of the first mutual coupling capacitor _C23 and the second mutual coupling capacitor _C24 are both 500 pF. The DC power supply 5 is a constant voltage source as in the first embodiment. FIG. 27 ₍ a) shows waveforms of currents flowing through the power transmission side coil _L21 and the power reception side coil L22. FIG. 27(b) shows waveforms of voltages V21 and V22 between the terminals of the power transmission side capacitor _C21 and the power reception side capacitor _C22 , respectively. _At 0 ns, the currents flowing through the power transmitting side coil L21 and the power receiving side coil L22 are ₃₀ A and 0 A, respectively, and at 60 ns, the currents flowing through the power transmitting side coil L21 and the power receiving side coil _L22 are ₃₀ A and 0 A, respectively. Characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 occurs.

As described above, according to the power transmission device according to the fifth embodiment of the present invention, even if electrostatic coupling is used, magnetic coupling in the power transmission devices according to the first to fourth embodiments As in the case of coupling, an inexpensive DC power supply 5 having simple control circuits and peripheral circuits can be used, eliminating the need for an expensive switching power supply. Also, even if the electrostatic coupling is used, the circuit configuration is simplified as in the power transmission devices according to the first to fourth embodiments, and the circuit design is simple and hard to break, and the power loss on the control circuit side is also minimized. As a result, according to the power transmission device according to the fifth embodiment of the present invention, the overall configuration of the power transmission device is simplified, making it possible to reduce the weight, size, and efficiency of the power transmission device. The overall power transmission efficiency, including the loss of power, has been increased to a value close to 100%, the limit power of power transmission has been pushed to infinity in principle, and the limit distance of power transmission has been extended to infinity in principle. A wireless power transmission device can be manufactured at low cost.

(Other embodiments)
As described above, the present invention has been described by the first to fifth embodiments, but the statements and drawings forming part of this disclosure should not be understood to limit the present invention. Various alternative embodiments, implementations and operational techniques will become apparent to those skilled in the art from this disclosure. For example, in the power transmission device according to the fifth embodiment of the present invention, the circuit configuration including only one primary side drive switch SW1 has been described as the electrostatic coupling method. It is only an example. As described in the power transmission device according to the second embodiment of the present invention, even in the case of the circuit configuration of the electrostatic coupling method, the configuration includes the primary side drive switch SW1 and the power transmission side switch SW2. It is possible.

Similarly, even in the case of the circuit configuration of the electrostatic coupling method as described in the power transmission device according to the third embodiment of the present invention, the primary side drive switch SW1, the power transmission side switch SW2, and the power receiving side A configuration including the switch SW3 is possible. Further, the configuration may be the same as that of the power transmission device according to the fourth embodiment of the present invention, including the primary side drive switch SW1, the power transmission side switch SW2, the power reception side switch SW3, and the load control switch SW4.

That is, the power transmission device according to the present invention is configured by combining the technical ideas of the respective embodiments shown in FIGS. You can also Further, the magnetic coupling degree control mechanism described with reference to FIGS. It may be applied to a power transmission device. As described above, the present invention includes various embodiments and the like that are not described in the specification and drawings, and the technical scope of the present invention is is defined only.

DESCRIPTION OF SYMBOLS 1... Power supply side circuit 2, 2C... Primary side circuit 3, 3C... Secondary side circuit 5... DC power supply 6... Load element 32... Spacer 33... Car stop 41... Ranging unit 411... Light emission Part 412... Light receiving part 42, 47... Logical operation control part 421... Distance calculation part 422, 472... Coupling coefficient calculation part 43... Coupling coefficient adjustment drive device 45... Data storage device 46... Transmission efficiency measurement Unit 461 Ammeter 462 Voltmeter 471 Transmission efficiency calculator 71 Substrate 72 Source region 73 Drain region 81 Gate oxide film 82 Source electrode 83 Drain electrode 84 … Gate electrode

Claims (8)

A power transmission device that does not use a switching power supply as an AC power supply circuit that generates a steady-state sinusoidal oscillation ,
a primary side circuit having a power transmission side capacitor, a power transmission side coil connected in parallel to the power transmission side capacitor, accumulating electrostatic energy sent from the power transmission side capacitor as magnetic energy, and circulating the magnetic energy to the power transmission side capacitor; ,
A DC power supply that supplies a DC voltage of a constant potential, wherein a high-voltage side terminal is connected to a connection node between one terminal of the power transmission side capacitor and one terminal of the power transmission side coil, and the other terminal of the power transmission side capacitor is connected to a connection node. a DC power supply that forms a circuit that connects a terminal on the low voltage side to the terminal and connects between one terminal and the other terminal of the power transmission side capacitor;
A primary side drive connected between said one terminal of said transmission side capacitor and said DC power supply by connecting one terminal to said high voltage side terminal of said DC power supply and connecting the other terminal to said connection node. a switch, wherein the intermittent DC voltage is step-inputted to the power transmission side capacitor for a predetermined period of time , and the LC includes the power transmission side coil and the power transmission side capacitor between the high voltage side terminal and the low voltage side terminal; From a single power semiconductor switching element, wherein parallel circuits are connected in parallel, the connection node is separated from the high-voltage side terminal in an interrupted state, and an LC series circuit including the power transmission side coil and the power transmission side capacitor is configured. a primary side drive switch,
a power receiving side coil that faces the power transmitting side coil and receives the magnetic energy from the power transmitting side coil; and a power receiving side capacitor that is connected in parallel to the power receiving side coil and stores the magnetic energy accumulated in the power receiving side coil as electrostatic energy. a secondary circuit having
a load element forming a circuit connecting between one terminal and the other terminal of the power receiving side capacitor and receiving the electrostatic energy from the power receiving side capacitor;
a load-side diode having an anode connected to the one terminal of the power-receiving-side capacitor and a cathode connected to the load element to transmit electric energy from the primary-side circuit to the secondary-side circuit without contact. A power transmission device characterized by: 2. The power transmission device according to claim 1, further comprising a power supply side diode having an anode connected to the DC power supply and a cathode connected to the one terminal of the power transmission side capacitor. 3. The power transmission device according to claim 1, further comprising a protective element connected in parallel with said power semiconductor switching element. 4. The power transmission device according to any one of claims 1 to 3, further comprising a magnetic coupling degree control mechanism that controls the degree of magnetic coupling between the power transmitting side coil and the power receiving side coil. . 5. The power transmission device according to claim 1, further comprising a power transmission side switch connected in series between said one terminal of said power transmission side capacitor and said power transmission side coil. . 6. The power transmission device according to claim 5, further comprising a power receiving side switch connected in series between said one terminal of said power receiving side capacitor and said power receiving side coil. 7. The power transmission device according to claim 6, further comprising a load control switch connected in series between said one terminal of said power receiving side capacitor and said load side diode. a primary side circuit having a power transmission side capacitor, a power transmission side coil connected in parallel to the power transmission side capacitor, storing electrostatic energy sent from the power transmission side capacitor as magnetic energy, and circulating the magnetic energy to the power transmission side capacitor; ,
a DC power supply that constitutes a circuit that connects between one terminal and the other terminal of the power transmission side coil;
a primary side drive switch connected between the one terminal of the power transmission side coil and the DC power supply for step-inputting an intermittent DC voltage to the power transmission side coil;
a first mutual coupling capacitor having one electrode connected to one node connecting the power transmission side capacitor and the power transmission side coil in parallel;
a second mutual coupling capacitor having one electrode connected to the other node connecting the power transmission side capacitor and the power transmission side coil in parallel;
one electrode connected to the other electrode of the first mutual coupling capacitor and the other electrode connected to the other electrode of the second mutual coupling capacitor for receiving the electrostatic energy from the primary side circuit a secondary side circuit having a side capacitor, a power receiving side coil connected in parallel to the power receiving side capacitor and storing electrostatic energy accumulated in the power receiving side capacitor as magnetic energy;
a load element forming a circuit connecting between one terminal and the other terminal of the power receiving coil and receiving the magnetic energy from the power receiving coil;
a load-side diode having an anode connected to the one electrode side of the power-receiving-side coil and a cathode connected to the load element, and transmitting electric energy from the primary-side circuit to the secondary-side circuit without contact. A power transmission device characterized by:

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