JP7314355B2 - Power transmission device and power transmission method - Google Patents

Description

The present invention relates to a power transmission device and a power transmission method using this power transmission device, and more particularly to a new circuit design for a wireless power transmission device and a power transmission method.

Karalis et al. of the Massachusetts Institute of Technology (MIT) requested a proposal in 2005, and research on wireless power transmission has become active (see Patent Documents 1, 2 and 3). The inventions described in Patent Documents 1, 2, and 3 are power transmission technologies that use an AC power source as a power source. However, as described in Patent Document 2, etc., in a wireless power transmission system based on AC theory, it is said that it is necessary to match the resonance frequency 2π√LC of the resonance circuit (LC circuit) on the power supply side with the resonance frequency 2π√LC of the resonance circuit (LC circuit) on the power reception side. However, in practice, the power-supplying-side resonant circuit and the power-receiving-side resonant circuit interact with each other, thereby causing new resonance. It is technically common knowledge that it is better not to cause resonance (multiple resonance) between the power feeding side resonance circuit and the power receiving side resonance circuit, including this new resonance.

According to Patent Document 2, a power-supply-side resonant circuit and a power-receiving-side resonant circuit are coupled by a magnetic field component (magnetic field coupling), and a mutual inductance M is formed between the power supply coil and the power reception coil. A new resonant circuit formed by the mutual inductance M, the power-supply-side resonant circuit, and the power-receiving-side resonant circuit has a resonant frequency _fr2 different from the resonant frequency _fr1 , and multiple resonance occurs. When trying to make the drive frequency fo of the AC power supplied to the power supply side resonance circuit follow the resonance frequency _fr1 , there is a possibility that the drive frequency fo will follow the resonance frequency fr2 instead of the resonance frequency _fr1 , which is the original target, and the resonance frequency _fr2 is an undesirable resonance point, and it is desirable to remove it. In the conventional AC theory, multiple _resonance has been avoided.

Moreover, the invention described in Patent Document 1 requires an AC power supply of 10 kHz to 50 GHz, the invention described in Patent Document 2 requires an AC power supply with a driving frequency fo = about 100 kHz, and the invention described in Patent Document 3 requires an AC power supply of several hundred kHz to several MHz. In particular, Patent Document 1 reports experimental data in a frequency band around 10 MHz. A frequency band power supply circuit (zero-order circuit) such as those described in Patent Documents 1 to 3 is created by using an expensive switching power supply from a commercial power supply to create a highly accurate direct current, then performing complex and precise switching of a large number of power semiconductor elements, and converting the DC pulse cut out on a rectangular wave into an alternating current in a pseudo or equivalent manner using PWM (Pulse Width Modulation) or the like. At this time, power loss such as resistance loss that occurs in the power semiconductor element and switching loss that rapidly increases with an increase in frequency occurs. In addition, switching elements are likely to be destroyed by back electromotive force induced in the coil or by excessive voltage rise due to resonance, and the higher the frequency and the greater the power, the more difficult the circuit design becomes. On the other hand, if you want to send power over long distances, you have to raise the frequency. As described above, the conventional wireless power transmission device based on the alternating current theory has problems that the device is complicated, the overall power transmission efficiency is low, the device is fragile, the reliability is low, and the cost is high. For these reasons, the conventional technology cannot realize wireless power transmission that efficiently transmits power over long distances, which will be required in the future. In other words, the circuit design itself of the power transmission device based on the AC theory is required to be studied.

U.S. Patent Application Publication No. 2008/0278264 Japanese Patent No. 5549745 Japanese Patent No. 5462953

In view of the above problems, the present invention focuses on transient response that is not the conventional AC theory, simplifies the circuit configuration, improves power transmission efficiency, and provides an inexpensive power transmission device and a power transmission method using this power transmission device.

A first aspect of the present invention includes (a) a primary side circuit that has a power transmission side capacitor and a power transmission side coil that accumulates electrostatic energy sent from the power transmission side capacitor connected to the power transmission side capacitor as magnetic energy and circulates this magnetic energy to the power transmission side capacitor, and that exhibits rising and falling transient response with a hump sawtooth wave due to accumulation and circulation; (b) a power receiving side coil that faces the power transmission side coil and receives magnetic energy from the power transmission side coil; is stored as electrostatic energy, and the power transmission device includes a secondary circuit exhibiting an oscillating waveform having no symmetry between rising and falling transient responses. In the power transmission device according to the first aspect, the time constant of the rising/falling transient response and the time constant inherent in the circuit characteristics of the secondary side circuit are set so as to harmonize, and electric energy can be transmitted from the primary side circuit to the secondary side circuit without contact.

A second aspect of the present invention is summarized in that it is a power transmission method for contactlessly transmitting electrical energy from a primary side circuit including a power transmission side capacitor and a power transmission side coil connected in parallel to the power transmission side capacitor to a secondary side circuit including a power reception side coil and a power reception side capacitor connected in parallel to the power reception side coil. The power transmission method according to the second aspect includes: (p) a step in which electrostatic energy accumulated in the power transmission side capacitor is sent to the power transmission side coil and stored as magnetic energy in the power transmission side coil, and the accumulated magnetic energy is circulated to the power transmission side capacitor; (q) a step in which a part of the magnetic energy accumulated in the power transmission side coil is received by the power reception side coil, and a part of the received magnetic energy is stored; A portion of this stored electrostatic energy is circulated back to the receiving coil.

According to the present invention, which is based on a circuit design that transcends conventional alternating current theory, it is possible to provide an inexpensive power transmission device and a power transmission method using this power transmission device by simplifying the circuit configuration and increasing power transmission efficiency by using characteristic harmonic transmission, which is a phenomenon during transient response.

FIG. 1(a) is a circuit diagram showing an outline of an example of the power transmission device according to the first embodiment of the present invention, and FIG. 1(b) is a waveform diagram of the voltage between the terminals of the power transmission side capacitor in the circuit shown in FIG. 1(a). FIG. 2(a) is a waveform diagram of voltages between terminals of the power transmission side capacitor and the power reception side capacitor of the power transmission device according to the first embodiment, and FIG. 2(b) is a detailed waveform diagram following the waveform of FIG. 2(a). It is a figure explaining the transient response at the time of step-inputting to an LC parallel circuit. FIG. 2 is a circuit diagram showing a mounted circuit of the power transmission device according to the first embodiment; FIG. 4 is a diagram illustrating a large-signal equivalent circuit of a MOSFET used in the power transmission device according to the first embodiment; FIG. 6(a) is a schematic diagram illustrating the importance of the inter-coil spacing of the power transmission device according to the first embodiment, and FIG. 6(b) is a bird's-eye view illustrating a magnetic coupling degree control mechanism that adjusts the inter-coil spacing when applied to charging the battery of an electric vehicle (EV). FIG. 7A is a bird's-eye view illustrating a mechanism for adjusting the magnetic coupling of the coils of the power transmission device according to the first embodiment, in which FIG. 7A shows a case in which the spacing between the coils is controlled using spacers, and FIGS. FIG. 8(a) is a block diagram explaining an example of the hardware configuration of the magnetic coupling degree control mechanism of the power transmission device according to the first embodiment, and FIG. 8(b) is a block diagram explaining another example. FIG. 4 is a schematic diagram for explaining the operation of the power transmission device according to the first embodiment divided for each timing in chronological order; FIG. 10(a) is a circuit diagram showing an outline of an example of a power transmission device according to a second embodiment of the present invention, and FIG. 10(b) is a circuit diagram showing a specific implementation circuit of the circuit of FIG. 10(a). FIG. 10 is a timing chart for explaining a power supply method of the power transmission device according to the second embodiment; 4A and 4B are schematic diagrams for explaining a power supply method of the power transmission device according to the second embodiment, (a) during charging, (b) during characteristic harmonic transmission between the primary circuit 2 and the secondary circuit 3, (c) during transfer, and (d) during characteristic harmonic transmission between the primary circuit 2 and the secondary circuit 3. FIG. 13(a) is a circuit diagram showing an outline of an example of a power transmission device according to a third embodiment of the present invention, and FIG. 13(b) is a circuit diagram showing a specific implementation circuit of the circuit shown in FIG. 13(a). 10 is a flowchart for explaining a power supply method for a power transmission device according to a third embodiment; FIG. 11 is a timing chart for explaining a method of supplying power to a power transmission device according to a third embodiment; FIG. 16(a) is a circuit diagram showing an outline of an example of a power transmission device according to a fourth embodiment of the present invention, and FIG. 16(b) is a circuit diagram showing a specific implementation circuit of the circuit shown in FIG. 16(a). FIG. 17A is a waveform diagram obtained by simulating the voltage across the terminals of the power transmission side capacitor and the power reception side capacitor in the power transmission device according to the fourth embodiment, and FIG. FIG. 18(a) is a waveform chart when the equivalent coupling coefficient K of the voltage between the terminals of the power transmitting side capacitor and the power receiving side capacitor in the power transmission device according to the fourth embodiment is 0.00, and FIG. 18(b) is a waveform chart when the equivalent coupling coefficient K is 0.1. FIG. 19A is a waveform chart when the equivalent coupling coefficient K of the terminal voltage of the power transmission side capacitor and the power reception side capacitor in the power transmission device according to the fourth embodiment is 0.6, FIG. 19B is a waveform chart when the equivalent coupling coefficient K is 0.8, and FIG. 19C is a waveform chart when the equivalent coupling coefficient K is 0.88. FIG. 20A is a waveform diagram when the equivalent coupling coefficient K of the voltage between the terminals of the power transmission side capacitor and the power reception side capacitor in the power transmission device according to the fourth embodiment is 0.6, and FIG. FIG. 11 is a graph showing changes in transmission efficiency with respect to capacitance of a capacitor of the power transmission device according to the fourth embodiment; FIG. It is a graph which shows the efficiency of the power transmission apparatus which concerns on 4th Embodiment. FIG. 11 is a flow chart illustrating a first power supply method for a power transmission device according to a fourth embodiment; FIG. FIG. 24(a) is a timing chart for explaining the first power supply method of the power transmission device according to the fourth embodiment, and FIG. 24(b) is a timing chart for explaining the second power supply method. FIG. 14 is a flowchart for explaining a second power supply method of the power transmission device according to the fourth embodiment; FIG. FIG. 11 is a circuit diagram showing an outline of an example of a power transmission device according to a fifth embodiment of the present invention; FIG. 27(a) is a waveform chart of currents flowing through the power transmission side coil and the power reception side coil of the power transmission device according to the fifth embodiment, and FIG. FIG. 4 is a circuit diagram used in approximate simulation for explaining sawtooth waves with knobs in the power transmission device according to the first embodiment of the present invention; FIG. 29 is a diagram showing transient response characteristics of a sawtooth wave with knobs obtained by approximate simulation for the circuit of FIG. 28; FIG. 29 is a diagram for explaining how the bumps of the sawtooth wave obtained by approximate simulation using the circuit of FIG. 28 change with repetition periods; 29 is a simplified circuit diagram in which parasitic capacitance, parasitic resistance, etc. are omitted from the circuit of FIG. 28; FIG. FIG. 32 is a diagram for explaining that a W-type transient response characteristic is obtained by approximate simulation for the simplified circuit of FIG. 31; FIG. 8 is a circuit diagram used when approximating the transient response characteristics of the power transmission device according to the second embodiment of the present invention by changing the power supply voltage and the load voltage; FIG. 34 is a diagram illustrating transfer of electric energy with transient response characteristics obtained by approximate simulation for the three circuits of FIG. 33; FIG. 4 is a schematic diagram illustrating W-type transient response characteristics of the power transmission device according to the first embodiment of the present invention;

Next, first to fifth embodiments of the present invention will be described with reference to the drawings. In the following description of the drawings, the same or similar parts are denoted by the same or similar reference numerals. However, it should be noted that the drawings are schematic, and that the relationship between thickness and planar dimension, the ratio of thickness of each member, and the like are different from the actual ones. Therefore, specific thicknesses and dimensions should be determined with reference to the following description. In addition, it is a matter of course that there are portions with different dimensional relationships and ratios between the drawings.

Further, the first to fifth embodiments shown below are examples of apparatuses and methods for embodying the technical idea of the present invention, and the technical idea of the present invention does not specify the material, shape, structure, arrangement, etc. of the component parts as follows. Various modifications can be made to the technical idea of the present invention within the technical scope defined by the claims. Furthermore, the directions of "left and right" and "up and down" in the following description are merely definitions for convenience of description, and do not limit the technical idea of the present invention. Therefore, for example, if the page is rotated 90 degrees, "left and right" and "up and down" are read interchangeably, and if the page is rotated 180 degrees, "left" becomes "right" and "right" becomes "left." Similarly, the direction of the helix of the spiral as shown in FIGS. 6(a) to 7(c) is merely a selection for convenience of explanation, and it is also possible to select left-handed for right-handed and right-handed for left-handed according to actual design circumstances.

(First embodiment)
The power transmission device according to the first embodiment of the present invention includes a primary circuit 2 and a secondary circuit 3, as shown in FIG. 1(a). The primary side circuit 2 is an LC resonance circuit having a power transmission side capacitor C ₁ that stores electrostatic energy, a power transmission side coil _{L 1} that is connected in parallel to the power transmission side capacitor C ₁ and that is connected in parallel to the power transmission side capacitor C ₁ and that accumulates the electrostatic energy sent from the power transmission side capacitor C ₁ as magnetic energy. A DC power supply 5 and a primary side drive switch SW1, which are connected in series, are connected in parallel to a transmission side capacitor _C1 . A DC power supply 5 supplies a DC voltage to the transmission side capacitor _C1 .

As will be described later, the “primary side drive switch SW1” is a circuit element that limits free vibration of the primary side circuit 2 . By limiting free oscillation, the primary side drive switch SW1 implements transient current-voltage changes in the primary side circuit 2. FIG. The DC power supply 5 may be a pseudo constant voltage source, or may be a DC power supply having a simple structure with only rectification and a power supply containing a large ripple component, so that a control circuit and peripheral circuits are simple, hard to break, circuit design is easy, and inexpensive. The secondary circuit 3 is an LC resonance circuit having a power receiving side coil L ₂ which is spaced apart from the power transmitting side coil L 1 and receives magnetic energy from the power transmitting side coil L ₁ , and a power receiving side capacitor C ₂ which is connected in parallel to the power receiving side coil L ₂ and stores the magnetic energy accumulated in the power receiving side coil _{L 2 as} electrostatic energy.

A load-side diode D2 and a load element 6, which are connected in series with each other, are connected in parallel to a receiving-side capacitor _C2 . A rechargeable battery such as a lithium (Li) ion battery for use in an electric vehicle (EV) can be used as the load element 6, for example. FIG. 1A schematically shows an equivalent circuit of a lithium-ion battery by way of a series-parallel circuit of resistors and capacitors. Lithium-ion batteries include current collectors, electrolyte resistors, and electrical double layer capacitors and resistors formed at interfaces within the battery. The load-side diode D2 is connected so that its anode faces the receiving-side resonator 2 and its cathode faces the load element 6, limiting the flow direction of the charging current _IC to one direction.

In FIG. 1(a) _{, let E be the voltage across the DC power supply 5 and the equivalent internal resistance r1, VC1 be the voltage across the terminals of the transmitting capacitor C1, VC2 be the voltage across the terminals of the receiving capacitor C2, VCS be the charging voltage measured across the terminals of the load element 6, and IC be the current flowing through the} equivalent floating resistance _rL on the load side. The equivalent internal resistance _r1 approximately indicates the internal impedance of the DC power supply 5 with a resistance value. FIG. 1(b) shows the transient response waveform of the inter-terminal voltage _VC1 obtained by actual measurement when the primary side drive switch SW1 is turned on and off.

In the first embodiment, it is assumed that the value of the charging voltage _VCS of the load element 6 in the initial state is a high value close to the charge completion voltage (close to full charge). At time t=0, the transmitting capacitor _C1 is not charged and the terminal voltage _VC1 =0. When the primary side drive switch SW1 is turned on at t=0, the inter-terminal voltage E of the DC power supply 5 is input stepwise. A step input at t=0 initially causes a charging current to capacitor C1, whose value is E/r1. At this time coil L1 generates a back EMF to block the incoming current, so the current into L1 is zero. As shown in FIG. 1(b), the power transmission side capacitor C1 is charged with a time constant _τ1 determined by the equivalent internal resistance _r1 , the power transmission side capacitor capacitance _C1 , _the power transmission side coil inductance _L1 , and the mutual inductance M, and the terminal voltage _VC1 increases. The time constant τ ₁ is mainly a value that depends on parameters related to the product C 1 · _r 1 of _the capacitance C ₁ of the transmission side capacitor C ₁ and the resistance value C ₁ of the equivalent internal resistance r ₁ .

As the voltage of C1 gradually rises and the current becomes smaller, the back electromotive force of L1 becomes smaller and the current begins to flow to L1. This causes the voltage across C1 to drop slightly. At this point, SW1 is closed (t=t1). The closing time t1 of this switch should be such that SW1 is not destroyed by the voltage applied to SW1 caused by the back electromotive force generated by turning off the coil when the current starts to flow. When the primary side drive switch _SW1 is turned off at t= _t1 , current flows from the power transmission side capacitor _C1 to the power transmission coil _L1 , and full-scale discharge starts. The electric energy stored in the power transmission side capacitor _C1 tries to move to the power transmission side coil _L1 . At this time, due to the magnetic field generated around _L1 due to the current flowing through the power transmission side coil _L1 , an electromotive force is generated in the power reception side coil _L2 coupled by the mutual inductance M, and current flows. As will be described later, if the characteristics of the primary side circuit 2 and the secondary side circuit 3 are in harmony at this time, the power receiving side capacitor _C2 is most efficiently charged by this transmitted power. That is, power is most efficiently transmitted from the primary side circuit 2 to the secondary side circuit 3 . An electromotive force is generated in L1 by a magnetic field generated around _the power receiving side coil _L2 due to the current flowing through the power receiving side coil L2. Due to the original voltage and this electromotive force, the voltage of the power transmission side coil _L1 becomes a sawtooth wave that deviates from a sine wave that is not a normal alternating current waveform. Slightly past the lowest voltage portion of the sawtooth wave, the sawtooth wave characteristic rises while rising like a bump.

The time constant τ ₂ until this bump-like swelling occurs mainly depends on parameters expressed in a relationship similar to equation (4) described later, with R _str (L ₁ ) representing the capacitance C _{1 of the transmission side capacitor C 1 ,} the inductance L ₁ of the transmission side coil L ₁ and the parasitic resistance of the transmission side coil L ₁ . However, it is necessary to consider mutual inductance M=M(t) between the inductance of the power transmission side coil L ₁ and the power reception side coil L ₂ , which is a value dependent on time t.

At t=t ₂ , the voltage of the primary side capacitor becomes a maximum peak again, and at this time the voltage of the secondary side capacitor becomes a value close to a minimum value. When the primary side drive switch SW1 is turned on again in accordance with this peak, the inter-terminal voltage E of the DC power supply 5 is input stepwise again. As before, a charging current flows to the capacitor _C1 at first, and its value is the value obtained by subtracting the voltage across the capacitor _C1 at t= _t2 from the voltage E across the terminals of the DC power supply 5 and dividing it by _r1 . At this time coil _L1 generates a back EMF to block the incoming current, so the current into _L1 is zero. As the voltage of C1 gradually rises and the current becomes smaller, the back electromotive force of L1 becomes smaller and the current begins to flow to L1. This causes the voltage across _C1 to drop slightly. At this point, SW1 is closed (t= _t3 ). The closing time _t3 of this switch should be such that SW1 is not destroyed by the voltage applied to _SW1 by the back electromotive force generated in the transmitting coil _L1 by turning off the coil when the current starts to flow. After the primary side drive switch SW1 is turned off at t= _t3 , current starts to flow from the power transmission side capacitor _C1 to the power transmission coil _L1 , and full-scale discharge starts. The electric energy stored in the power transmission side capacitor _C1 tries to move to the power transmission side coil _L1 . At this time, due to the magnetic field generated around _L1 due to the current flowing through the power transmission side coil _L1 , an electromotive force is generated in the power reception side coil _L2 coupled by mutual inductance M, and current flows. As will be described later, if the characteristics of the primary side circuit 2 and the secondary side circuit 3 are in harmony at this time, the power receiving side capacitor _C2 is most efficiently charged by this transmitted power. That is, power is most efficiently transmitted from the primary side circuit 2 to the secondary side circuit 3 . Electromotive force is generated in L1 by a magnetic field generated around the power receiving side coil _L2 due to the current flowing through _the power receiving side coil L2. Due to the original voltage and this electromotive force, the voltage of the power transmission side coil _L1 becomes a sawtooth wave that deviates from a sine wave that is not a normal alternating current waveform. Slightly past the lowest voltage portion of the sawtooth wave, the sawtooth wave characteristic rises while rising like a bump.

As a result, after t= _t3 , as shown on the right side of FIG. 1(b), the voltage _VC1 across the terminals of the power transmission side capacitor _C1 decreases and becomes a negative value again. When the voltage _{VC 1} across the terminals of the power transmission side capacitor _{C 1} becomes a negative value, the electrical energy stored in the power transmission side coil L ₁ begins to circulate to the power transmission side capacitor C ₁ , and as shown in the right end of FIG. The time up to this point is determined by the time constant τ ₂ determined by the capacitance C ₁ of the power transmission side capacitor, the inductance L ₁ of the power transmission side coil, and the mutual inductance M. As shown in FIG. 1(b), when the voltage E across the terminals of the DC power supply 5 is stepped in and cut off by the primary-side drive switch SW1, the change in the voltage across the terminals _VC1 is not a sinusoidal waveform in the usual AC theory, but a transient response of a repetitive rising/falling waveform with dull sawtooth waves.

When the time constant inherent in the circuit characteristics of the primary circuit 2 and the time constant inherent in the circuit characteristics of the secondary circuit 3 are in harmony, the electrical energy of the primary circuit 2 is most efficiently transmitted to the secondary circuit 3, resulting in characteristic harmonious transmission between the primary circuit 2 and the secondary circuit 3. FIG. 2A shows the transient response waveforms of the voltage _VC1 across the terminals of the primary side circuit 2 and the voltage VC2 across the terminals of the secondary side circuit 3 when the characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit ₃ occurs. In FIG. 2(a), as in FIG. 1(b), due to the step input and interruption of the voltage E between the terminals of the DC power supply 5 by the primary side drive switch SW1, the voltage _VC1 between the terminals shows a repetitive transient response waveform with rising and falling sawtooth waves with bumps. The transient response waveform shown by the solid line in FIG. 2(a) shows how the voltage _VC1 across the terminals of the power transmission side capacitor _C1 starts increasing from a negative value, becomes a positive value, and further increases until the primary side drive switch SW1 turns on at t= _ti .

At this time, as shown by the dashed line on the left side of FIG. 2(a), the power receiving side capacitor _C2 of the secondary side circuit 3 is charged by the characteristic harmonic transmission between the primary side circuit ₂ and _{the secondary side circuit 3, and after the voltage between the terminals of the power receiving side capacitor C2 reaches the peak value, the power receiving side capacitor C2 starts discharging, and the voltage between the terminals of the power receiving side capacitor C2 starts to} decrease. When the primary side drive switch SW1 is turned on at t=t _i , the inter-terminal voltage E of the DC power supply 5 is input stepwise. The terminal voltage _VC2 of the receiving side capacitor _C2 of the secondary side circuit 3 decreases to a negative value at t= _ti .

_Due to the step input at t= _ti , the power transmission side capacitor _C1 of the primary circuit 2 is charged with a time constant _τ determined by the equivalent internal resistance _r1 , the power transmission side capacitor capacitance _C1 , the power transmission side coil inductance _L1 , and the mutual inductance M, as shown near the center left of FIG. As shown in FIG. 2(a), the voltage _VC1 across the terminals of the primary circuit 2 reaches a peak value and then begins to decrease. When the primary side drive switch SW1 is turned off at t=t _i+1 , the power transmission side capacitor _C1 of the primary side circuit 2 starts full-scale discharge, and the electric energy stored in the power transmission side capacitor _C1 moves to the power transmission side coil _L1 . The terminal voltage _VC2 of the receiving side capacitor _C2 of the secondary side circuit 3 has a negative value between t=t _i and t=t _i+1 as indicated by the dashed line.

After t=t _i+1 , as indicated by the solid line near the center of FIG. 2(a), the voltage _VC1 across the terminals of the power transmission capacitor C1 decreases to a negative value due to the time constant _τ2 determined by the capacitance _C1 of the power transmission side capacitor, the inductance _L1 of the power transmission side coil, and the _mutual inductance M, and the electrical energy accumulated in the power transmission side capacitor _C1 is transferred to the power transmission side coil _L1 . The electrical energy stored in the power transmission side coil _L1 is wirelessly transmitted to the power reception side coil L2 of the secondary side circuit 3 by characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 between the primary side circuit 2 and the secondary side circuit ₃ . The electric energy wirelessly transmitted to the power receiving side coil L2 of the secondary side circuit ₃ is stored in the power receiving side capacitor _C2 of the secondary side circuit 3 .

As a result, after t=t _i+1 , the terminal voltage _VC2 of the receiving side capacitor _C2 of the secondary circuit 3 starts to increase, as indicated by the dashed line near the center of FIG. 2(a). After reaching a peak value, the terminal voltage _VC2 of the receiving side capacitor _C2 of the secondary side circuit 3 starts to decrease and becomes a negative value as indicated by the dashed line on the right side of FIG. 2(a). As the conditions in this embodiment, since the value of the charging voltage _VCS of the load element 6 in the initial state is close to the charging completion voltage (close to full charge) and high, the voltage _VC2 across the terminals of the power receiving side capacitor _C2 exceeds the charging voltage VCS of the load element ₆ near the peak value, so that the current flows through the load element 6, the electric energy accumulated in the power receiving side capacitor C2 moves to the load element 6, and the rechargeable battery, which is the load element 6, is charged.

After t ₌ t _i+1 , when the voltage VC ₁ across the terminals of _the power transmission side _{capacitor C 1 reaches} the minimum negative _value as indicated by the solid line in the center of FIG. As indicated by the dashed line on the right side of FIG. 2(a), when the terminal voltage _VC1 takes a positive value, the terminal voltage _VC2 of the receiving-side capacitor _C2 takes a negative value.

When the voltage _VC1 across the terminals of the power transmission side capacitor _C1 increases to a positive value and the primary side drive switch SW1 is turned on again at t=t _i+2 , the voltage E across the terminals of the DC power supply 5 is stepped in. Due to the step input at t=t _i+2 , the power transmission side capacitor _C1 is charged as indicated by the solid line on the right side of FIG. 2(a), and the inter-terminal voltage _VC1 increases. As shown by the solid line in FIG. 2(a), the terminal voltage _VC1 reaches the peak value _E0 and then starts to decrease again. When the primary side drive switch SW1 is turned off at t=t _i+3 , the power transmission side capacitor _C1 starts discharging again, and the electric energy stored in the power transmission side capacitor _C1 moves to the power transmission side coil _L1 . The terminal-to-terminal voltage _VC2 of the power receiving side capacitor _C2 of the secondary side circuit 3 is a negative value between t=t _i+2 and t=t _i+3 as indicated by the dashed line.

The electrical energy stored in the power transmission side coil _L1 is wirelessly transmitted to the power reception side coil L2 of the secondary side circuit 3 by characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 between the primary side circuit 2 and the secondary side circuit ₃ . The electrical energy wirelessly transmitted to the power receiving side coil L2 of the secondary side circuit ₃ is stored in the power receiving side capacitor _C2 of the secondary side circuit 3 . As shown in FIG. 2(a), due to the step input and cutoff of the inter-terminal voltage E by the primary side drive switch SW1, the change in the inter-terminal voltage _VC1 of the primary side circuit 2 is not a sinusoidal waveform in the usual AC theory, but a transient response of a repetitive rising/falling characteristic with a rounded sawtooth wave. On the other hand, the change in the inter-terminal voltage _VC2 of the secondary circuit 3 exhibits a transient response of a repetitive waveform like a thinned-out triangular wave, but it is not a sinusoidal waveform in normal AC theory. As can be seen from FIG. 2(a), the "decimated triangular wave" can be interpreted as a waveform obtained by reversing the polarity of the trapezoidal wave. In any case, the vibration waveform of the primary circuit 2 and the vibration waveform of the secondary circuit 3 are not mutually symmetrical.

FIG. 2(b) shows measured transient response waveforms obtained by adding the voltage E across the terminals of the DC power supply ₅ , the voltage _VCS across the terminals of the load element 6, and the charging current _IC to the rechargeable battery that is the load element 6 to the transient response waveforms of the voltage across the terminals VC1 and voltage _VC2 shown in FIG. 2(a). At t=t _i , the primary side drive switch SW1 is turned on to store electric charge in the power transmission side capacitor _C1 , and then at t=t _i+1 , the primary side drive switch SW1 is turned off. When the primary side drive switch SW1 is turned on, the equivalent internal resistance _r1 of the DC power supply 5 is small, so the inter-terminal voltage E of the DC power supply ₅ indicated by the thick solid line in FIG.

Description will be made focusing on the transient response after t=t _i+1 in FIG. 2(b). As indicated by the solid line near the center of FIG. 2(b), the voltage _VC1 across the terminals of the power transmission side capacitor _C1 decreases after t=t _i+1 and becomes a negative value. The electric energy accumulated in the power transmission side capacitor _C1 is transferred to the power transmission side coil _L1 and accumulated in the power transmission side coil _L1 . The electrical energy stored in the power transmission side coil _L1 is transmitted to the power reception side coil L2 of the secondary side circuit 3 by characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 between the primary side circuit 2 and the secondary side circuit ₃ . Since the electrical energy transmitted to the power receiving coil _L2 of the secondary circuit 3 is stored in the power receiving capacitor C2 of the secondary circuit ₃ , the voltage _VC2 across the terminals of the power receiving capacitor _C2 starts increasing from a negative value after t=t _i+1 , as indicated by the dashed line near the center of FIG. 2(b).

The terminal-to-terminal voltage _VC2 of the power receiving capacitor _C2 of the secondary circuit 3 has a negative value between t=t _i and t=t _i+1 as indicated by the dashed line from the left side of the center. The voltage _VC2 across the terminals of the receiving side capacitor _C2 increases from a negative value at t=t _i+1 , becomes a positive value, further increases, reaches a peak value, and then starts decreasing as indicated by the broken line on the right side of FIG. 2(b). When the terminal voltage _VC2 decreases, the electrical energy accumulated in the power receiving capacitor _C2 flows into the load element 6 as the charging current _IC indicated by the dashed line on the right side of FIG. 2(b), and the load element 6 is charged. Almost in synchronism with the increase in the charging current _IC indicated by the dashed line, the voltage _VCS across the terminals of _the load element 6 indicated by the dotted line on the right side of FIG. When the charging current _I.sub.C decreases to zero, the voltage _VCS across the terminals of the load element 6 stops decreasing, starts increasing, and the voltage across the terminals of the load element 6 becomes a steady value. A change in Vcs according to Ic occurs due to the presence of a series-parallel circuit of resistors and capacitors, the equivalent circuit of which is illustrated in FIG. 1(a).

After t ₌ t _i+1 , when the voltage _{VC 1} across the terminals _of the power transmission side _{capacitor C 1 reaches} the minimum negative value as shown by the solid line in the center of FIG. As indicated by the dashed line on the right side of FIG. 2(b), when the terminal voltage _VC1 takes a positive value, the terminal voltage _VC2 of the power receiving capacitor _C2 takes a negative value.

When the voltage _VC1 across the terminals of the power transmission side capacitor _C1 increases to a positive value and the primary side drive switch SW1 is turned on again at t=t _i+2 , the voltage E across the terminals of the DC power supply 5 is stepped in. Due to the step input at t=t _i+2 , the power transmission side capacitor _C1 is charged as indicated by the solid line on the right side of FIG. 2(b), and the inter-terminal voltage _VC1 increases. As described above, since the equivalent internal resistance _r1 of the DC power supply 5 is small, when the primary-side drive switch SW1 is turned on, the inter-terminal voltage E of the DC power supply ₅ indicated by the thick solid line in FIG. As indicated by the solid line superimposed on the terminal voltage E on the right end of FIG. 2(b), the terminal voltage _VC1 reaches its peak value and then starts to decrease again. When the primary side drive switch SW1 is turned off at t=t _i+3 , the power transmission side capacitor _C1 starts discharging again, and the electric energy stored in the power transmission side capacitor _C1 moves to the power transmission side coil _L1 . The terminal-to-terminal voltage _VC2 of the receiving side capacitor _C2 of the secondary side circuit 3 has a negative value between t=t _i+2 and t=t _i+3 as indicated by the dashed line.

The behavior before t=t _i is similar, and the voltage _VC2 across the terminals of the receiving side capacitor _C2 starts decreasing after reaching a peak value, as indicated by the dashed line on the left side of FIG. 2(b). When the terminal voltage _VC2 increases and exceeds Vcs, the electrical energy accumulated in the power receiving side capacitor _C2 flows into the load element 6 as the charging current _IC indicated by the dashed line on the left side of FIG. 2(b), and the load element 6 is charged. Almost in synchronism with the increase in the charging current _IC indicated by the dashed-dotted line, the voltage _VCS across the terminals of _the load element 6 indicated by the dotted line on the left side of FIG. When the charging current _IC decreases to zero, the voltage _VCS across the terminals of the load element 6 stops decreasing and starts to increase, and the voltage across the terminals of the load element 6 becomes a steady value. The change in Vcs according to Ic is caused by the existence of a series-parallel circuit of resistors and capacitors, the equivalent circuit of which is illustrated in FIG. 1(a).

Then, when the primary side drive switch SW1 is turned off at t=t _i+1 already described, the terminal voltage _VC1 of the power transmission side capacitor _C1 starts to decrease toward a negative value as indicated by the solid line in the center of FIG. 2(b). In this way, the electrical energy stored in the power transmission side coil _L1 is wirelessly transmitted to the power reception side coil L2 of the secondary side circuit 3 by characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 between the primary side circuit 2 and the secondary side circuit ₃ . The electrical energy wirelessly transmitted to the power receiving side coil L2 of the secondary side circuit ₃ is stored in the power receiving side capacitor _C2 of the secondary side circuit 3 . As shown in FIG. 2(b), due to the step input and cutoff of the inter-terminal voltage E by the primary-side drive switch SW1, the change in the inter-terminal voltage _VC1 of the primary-side circuit 2 is not a sinusoidal waveform in normal AC theory, but a repetitive transient response waveform with rising and falling characteristics with a rounded sawtooth wave _.

FIG. 3(a) is a circuit diagram for explaining the step response of the primary circuit 2 as a single circuit when there is no electromagnetic coupling between the primary circuit 2 and the secondary circuit 3. FIG. In FIG. 3A, the voltage across the terminals of the DC power supply is E, the voltage across the terminals of the transmitting capacitor _C1 is _VC1 , and the current flowing through the power transmitting coil _L1 is the power transmitting coil current _IL1 . When the primary side drive switch SW1 is turned on at t=0 ms, the inter-terminal voltage E of the DC power supply 5 is input stepwise. Due to the step input at t=0 ms, as shown in FIG. 3B, the time constant determined by the capacitance C ₁ of the power transmission side capacitor and the inductance L ₁ of the power transmission side coil defines the rising waveform V _rise of the charging of the power transmission side capacitor C ₁ , and the terminal voltage VC ₁ increases with the rising waveform V _rise . At the same time, the transmission side coil current _IL1 also starts to increase as shown in FIG. 3(b).

As shown in FIG. 3B, the inter-terminal voltage _VC1 increases with the rising waveform _Vrise , reaches a peak value at t=0.075 ms, and then starts decreasing. It can be seen that the power transmission side coil current _IL1 continues to increase even after the inter-terminal voltage _VC1 starts decreasing, and the electric energy of the power transmission side capacitor _C1 continues to move to the power transmission side coil _L1 . When the primary-side drive switch SW1 is turned off at t=0.15 ms, the power-transmitting-side capacitor _C1 starts full-scale discharge and sharply decreases with the falling waveform V _fall . At this time, the power transmission side coil current _IL1 continues to increase, and the electric energy stored in the power transmission side capacitor _C1 moves to the power transmission side coil _L1 until the power transmission side coil current _IL1 reaches the peak value at t=0.17ms.

As a result, after t=0.15 ms, as _{shown in FIG. 3(b), the voltage VC1 across the terminals of the power transmission side capacitor C1 decreases with a time constant determined by the capacitance C1 of the power transmission side capacitor and the inductance L1 of the power transmission side} coil. After t=0.17 ms, the terminal voltage _VC1 of the power transmission side capacitor _C1 becomes a negative value. When the terminal voltage _VC1 of the power transmission side capacitor _C1 becomes a negative value, the value of the power transmission side coil current _IL1 begins to decrease, and the electrical energy stored in the power transmission side coil _L1 begins to circulate to the power transmission side capacitor _C1 . Then, as shown in FIG. 3(b), when the value of the power transmission side coil current _IL1 becomes zero at t=0.29 ms, the terminal voltage _VC1 of the power transmission side capacitor _C1 starts to increase due to the circulating current. When the value of the transmitting side coil current _IL1 reaches the minimum negative value at t=0.4 ms, the terminal voltage _VC1 becomes zero, after which the terminal voltage _VC1 becomes a positive value and further increases.

The circuit shown in FIG. 3(a) does not turn on the primary drive switch SW1 again after turning off the primary drive switch SW1 at t=0.15 ms. In other words, in the case of the circuit shown in FIG. 3(a), since free vibration occurs in the shaded region on the right side of FIG. However, in the circuit shown in FIG. 1(a), the primary-side drive switch SW1 performs forced step-response driving that periodically repeats ON/OFF. Therefore, the region of free vibration indicated by hatching in FIG. 3(b) is out of the scope of the power transmission device according to the first embodiment. In the forced step response, as shown in FIGS. 1(b) and 2(b), the voltage VC ₁ across the terminals of the primary circuit 2 shows a transient response of a repetitive rising/falling characteristic with a rounded sawtooth wave. Further, the voltage _VC2 across the terminals of the secondary circuit 3 exhibits a transient response of a repetitive waveform such as a thinned triangular wave.

That is, in the power transmission device according to the first embodiment, when the time constant inherent in the circuit characteristics of the primary circuit 2 and the time constant inherent in the circuit characteristics of the secondary circuit 3 are harmonized, the electrical energy of the primary circuit 2 is efficiently transmitted to the secondary circuit 3 by characteristic harmonic transmission between the primary circuit 2 and the secondary circuit 3, instead of resonance based on a sine wave in ordinary AC theory. In order to harmonize the time constant inherent in the circuit characteristics of the primary side circuit 2 and the time constant inherent in the circuit characteristics of the secondary side circuit 3, the product of the power transmission side capacitor C1 and the power transmission side coil L1 and the product of the power reception side capacitor C2 and the power reception side coil L2 should basically be the same, and the respective time constants, which take into account the parasitic resistance and stray capacitance of the power transmission side and the power reception side, should be matched or integrally multiplied. The simplest method is to equalize the capacitances of the power transmission side capacitor C1 and the power reception side capacitor C2 including the parasitic resistance of the capacitors, and to make the inductances of the power transmission side coil _L1 and the power reception side coil _{L2 the same} including the parasitic resistance of the coils. It should be noted that when parasitic resistance exists in the coil and the capacitor, the transmission efficiency is maximized when the inductance L of the coil and the capacitance C of the capacitor satisfy equation (19), as will be described later.

As the primary side drive switch SW1 shown in FIG. 1(a), in addition to a mechanical switching element such as an electromagnetic contactor, as a more preferable embodiment, a power semiconductor switching element capable of high-speed switching is used. As power semiconductor switching elements, thyristors such as field effect transistors (FET), static induction transistors (SIT), bipolar transistors (BJT), gate turn-off thyristors (GTO) and static induction thyristors (SI thyristors) are suitable. In particular, voltage-driven switching elements such as an insulated gate field effect transistor (MISFET), an insulated gate static induction transistor (MISSIT), an insulated gate bipolar transistor (IGBT), and a MOS-controlled SI thyristor are preferable due to their low power consumption. Based on availability in the market and evaluation of the internal resistance of semiconductor switching elements for electric power, it is currently possible to adopt a MOS field effect transistor (MOSFET), which is a type of MISFET, as shown in the circuit shown in FIG. 4(a).

In a power transmission device that uses a rechargeable battery for an EV as a load element 6, a large amount of Joule heat is generated due to the flow of a large current, resulting in heat generation of several hundred watts or more, and the power transmission device becomes a heating device (heater). In the power transmission device according to the first embodiment, only one power semiconductor switching element is required to be used as the primary side drive switch SW1. Therefore, it is possible to easily design a structure in which the element is covered with a heat sink such as a copper block to increase thermal conductivity and prevent the element from being destroyed due to heat generation, while also minimizing the occurrence of stray resistance, stray capacitance, and stray inductance. Since the heat generated by the floating resistance (parasitic resistance) of the power transmission side coil _L1 and the power reception side coil _L2 is also large, measures such as cooling the power transmission side coil _L1 and the power reception side coil _L2 with air or water are preferable.

One method of suppressing the generation of Joule heat in a high-power power transmission device such as an EV in-vehicle is to increase the voltage of the primary side circuit 2 and set the voltage of the secondary side circuit 3 to the optimum voltage of the load element 6 by the winding ratio of the power transmission side coil L ₁ and the power reception side coil L ₂ . When a power semiconductor switching element is employed as the primary side drive switch SW1, simple control of ON/OFF control of the power semiconductor switching element is sufficient, so it is easy to design a circuit that increases the voltage of the primary side circuit 2.

As described above, according to the power transmission device according to the first embodiment, since the primary side drive switch SW1 is a simple design with only one piece, it is easy to increase the voltage of the primary side circuit 2 and minimize the power loss due to the generation of Joule heat in the primary side circuit 2 side. Energy loss due to Joule heat generation can also be reduced, so according to the power transmission device according to the first embodiment, the overall power transmission efficiency including the loss of the power supply circuit (zero-order circuit) in the case of power transmission for high power such as in-vehicle use of EV is increased, and it can contribute to solving the energy problem of mankind.

In the mounting circuit shown in FIG. 4(a), considering the circulating current from the power transmission side coil _L1 , the first free wheel diode (freewheel diode) _FWD1 is connected in parallel as a protective element between the source and drain of the MOSFET as the first semiconductor switching element Q1. In order to prevent the circulating current from the power transmitting coil _L1 from circulating to the DC power supply 5, a power supply diode D1 is connected in series between the DC power supply 5 and the first semiconductor switching element Q1. In the mounting circuit shown in FIG. 4(a), the equivalent impedance X _Leq of the load element 6 is expressed by approximating it with the charge capacity _Cs .

In ac theory, which relies on normal steady-state sinusoidal waves, the mutual inductance M between the sending coil L ₁ and the receiving coil L ₂ is given by the coupling coefficient K _AC :

M=K _AC ( _L1・_L2 ) ^1/2 (1)

can be shown. Then, the mutual induction between the series circuit of the power transmission side capacitor _C1 and the power transmission side coil _L1 and the series circuit of the power reception side capacitor _C2 and the power reception side coil _L2 is expressed by the following coupling equation using the mutual inductance M

_L1di1 /dt+(1/ _C1 ) _∫i1dt + _{Mdi2 /} dt=0 (2)
_L2di1 /dt+(1/ _C2 ) _∫i2dt + _{Mdi1 /} dt=0 (3)

represented by In equations (2) and (3), ∫ is the integral symbol. That is, the mounting circuit shown in FIG. 4A can be expressed as shown in FIG.

However, since the power transmission device according to the first embodiment is a transient response transmission technology that does not rely on the sine wave AC theory, the representation of the equivalent circuit in FIG. 4B is merely a schematic diagram for considering an approximate physical model. Maxwell's equations in the time-varying case can be solved analytically when the time-varying relies on a sine wave. However, when there is a sawtooth-like time change as shown in FIGS. 1(b) to 2(b), it is extremely difficult to analytically solve Maxwell's equations. Therefore, the mutual inductance M used in AC theory is a time-dependent parameter represented by a function M(t) with t as time in the present invention, and attention must be paid to the expression of the equivalent circuit in FIG. 4(b).

FIG. 5 shows a large-signal equivalent circuit of an nMOSFET used as an example of the first semiconductor switching element Q1 in the mounting circuit shown in FIG. 4(a). As shown in FIG. 5(a), a general nMOSFET has a p-type substrate 71, an n ⁺ -type source region 72 and an n ⁺ -type drain region 73 facing each other across the surface of the p-type substrate 71 serving as a channel region. A gate electrode 84 is provided on the channel regions of the source region 72 and the drain region 73 with a gate oxide film 81 having a thickness _TOX interposed therebetween. A source electrode 82 and a drain electrode 83 are in ohmic contact with the source region 72 and the drain region 73, respectively.

As shown in FIG. 5(a), a general nMOSFET has a gate-source capacitance _CGS between the gate electrode 84 and the source region 72, a gate-drain capacitance _CGD between the gate electrode 84 and the drain region 73, and a gate-substrate capacitance _CGB between the gate electrode 84 and the substrate 71. Furthermore, between the source region 72 and the substrate 71 there is a source-substrate capacitance _CBS , and between the drain region 73 and the substrate 71 there is a drain-substrate capacitance _CBD . The equivalent circuit shown in FIG. 5B shows a configuration in which a drain resistor R _D connected in series between the drain electrode and the channel region, and a source resistor R _S connected in series between the source electrode and the channel region are connected in series to a constant current source of current I _DS provided in the channel region.

In the power transmission device according to the first embodiment shown in FIG. 4, the drain resistance R _D and the source resistance R _S of the MOSFET shown in FIG. 5, which serve as the on-resistance of the first semiconductor switching element Q1, have an important meaning, and it is necessary to select an element with a small on-resistance for the first semiconductor switching element Q1. Therefore, in the implementation circuit shown in FIG. 4(a), it is necessary to include the on-resistance of the first semiconductor switching element _Q1 in the equivalent internal resistance r1 of the DC power supply 5 to determine the time constant inherent in the circuit characteristics of the primary side circuit 2.

When the primary side drive switch SW1 is turned off at t= _t1 in FIG. 1(b), the primary side circuit 2 becomes an RLC series circuit. According to AC theory, if the parasitic resistance of the power transmission side coil _L1 is R _str ( _L1 ), then the damping coefficient _ζ1 of the RLC series circuit is

ζ ₁ = (R _str (L ₁ )/2) (C ₁ /L ₁ ) ^1/2 (4)

is represented. However, although it is necessary to consider the mutual inductance M=M(t) shown in FIG. 4(b) for the inductance of the power transmission side coil _L1 , it is difficult to analytically explain the mutual inductance M=M(t).

Ignoring the load diode D2 on the load side of the secondary circuit 3, the load element 6, etc. at this time, it can be regarded as an RLC series circuit. If the AC theory is adopted ignoring the load-side diode D2, the load element 6, etc., and the parasitic resistance of the power-receiving-side coil _L2 is R _str ( _L2 ), the damping coefficient _ζ2 of the secondary-side circuit 3 is:

ζ ₂ = (R _str (L ₂ )/2) (C ₂ /L ₂ ) ^1/2 (5)

is represented. In equation (5) as well, the inductance of the power receiving side coil _L2 must take into consideration the mutual inductance M=M(t) shown in FIG. 4(b).

According to AC theory, the resonance frequency of the RLC series circuit configured with the primary-side drive switch SW1 turned off at t= _t1 in FIG.

f _o1 = (1/2π) (C ₁ L ₁ ) - ^1/2 (6)

can be approximated as As described above, it should be noted that in equation (6), it is actually necessary to consider the mutual inductance M=M(t) shown in FIG. 4(b) as the inductance of the power receiving side coil _L1 . Similarly, according to AC theory, the resonance frequency of the RLC series circuit when the load diode D2 on the load side of the secondary circuit 3 and the load element 6, etc. are ignored is

f _o2 =(1/2π)( _C2 · _L2 ) ^-1/2 (7)

can be approximated as In equation (7) as well, it is necessary to consider the mutual inductance M=M(t) shown in FIG. 4(b) for the inductance of the power receiving side coil _L2 .

The power transmitting side coil L ₁ and the power receiving side coil L ₂ of the power transmission device according to the first embodiment can be, for example, spiral planar coils as shown in FIGS. 6(a) to 7(c). The primary side circuit 2 and the secondary side circuit 3 can be approximated by a circuit magnetically coupled with an equivalent coupling coefficient K and a mutual inductance M between the power transmission side coil L ₁ and the power reception side coil L ₂ in a steady state where the AC theory holds, as shown in FIG. 4(b). Here, the "equivalent coupling coefficient K" is a pseudo coupling coefficient in an unsteady state defined during transient response, which is equivalent to the coupling coefficient K _AC derived from AC theory, and is strictly a time-dependent parameter. Therefore, even in characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3, the time constant inherent in the circuit characteristics of the primary side circuit 2 and the time constant inherent in the circuit characteristics of the secondary side circuit 3 can be evaluated by the "magnetic coupling degree" similar to the coupling coefficient K _AC of the AC theory.

FIGS. 6(a) to 7(c) are schematic diagrams specifically showing the structures of the power transmission side coil L ₁ and the power reception side coil L ₂ of FIG. 4(a). In the power transmission device according to the first embodiment, for example, a wiring cable having a conductor cross-sectional area of 16 mm ² is wound nine times to form a spiral planar coil having a diameter of approximately 30 cm. These two spiral planar coils with a diameter of about 30 cm are arranged in parallel and in a non-contact manner with a gap of d in between. The efficiency of characteristic harmonic transmission between the primary circuit 2 and the secondary circuit 3 from the primary circuit 2 to the secondary circuit 3 depends on the value of the degree of magnetic coupling, which is similar to the coupling coefficient K _AC defined in AC theory. The degree of magnetic coupling depends on the spacing d between the two spiral planar coils, and it is necessary to control the spacing d between the two spiral planar coils.

The degree of magnetic coupling can be adjusted by mechanically adjusting the positional relationship between the two spiral planar coils, inserting a magnetic material between the two spiral planar coils, arranging a magnetic material around the two spiral planar coils, or attaching a preformed coupling using the attraction or repulsion acting between the two spiral planar coils.

As will be described in the fourth embodiment and the like, the mutual relationship between the power transmitting side coil L ₁ and the power receiving side coil L ₂ that has an equivalent coupling coefficient K that can be approximately approximated to the coupling coefficient K _AC = 0.6 in AC theory is suitable for characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3. In the case of spiral planar coils each having 9 windings of a wiring cable with a conductor cross-sectional area of 16 mm ² , the distance d needs to be about 0 Cm to 2.0 Cm in order to realize the equivalent coupling coefficient K=0.6. On the other hand, the distance d is about 10 cm in order to realize the mutual relationship between the power transmitting side coil L ₁ and the power receiving side coil L ₂ under the condition of the equivalent coupling coefficient K which can be approximated to the coupling coefficient K _AC =0.1 of the AC theory. As schematically shown by exaggerating (enlarging) the power transmission side coil L ₁ and the power reception side coil L ₂ in FIG. 6B, in order to charge the load element 6, which is a rechargeable battery for EV, using the power transmission device according to the first embodiment, the distance d between the power transmission side coil L ₁ and the power reception side coil L ₂ can be controlled to about 10 cm using the rear wheel stop 33 as a magnetic coupling degree control mechanism, and efficient wireless power supply can be performed.

As a magnetic coupling degree control mechanism for controlling the distance d between the power transmission side coil _L1 and the power reception side coil _L2 , as shown in FIG. 7A, by inserting a spacer ₃₂ having a thickness of d0 between the power transmission side coil _L1 and the power reception side coil _L2 , the distance d between the power transmission side coil _L1 and the power reception side coil _L2 can be controlled to be d= _d0 . Note that the interrelationship between the power transmission side coil L ₁ and the power reception side coil L ₂ , which corresponds to the value of the degree of magnetic coupling that is important for the efficiency of the characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 from the primary side circuit ₂ to the secondary side circuit 3, can configure the magnetic coupling degree control mechanism by parameters other than the distance d between the power transmission side coil L 1 and the power reception side coil L ₂ .

For example, as shown in FIG. 7B, a magnetic coupling degree control mechanism may be configured by inserting a magnetic plate 31C such as ferrite having a magnetic permeability μ _r between the power transmission side coil L ₁ and the power reception side coil L ₂ . The degree of magnetic coupling between the power transmission side coil L ₁ and the power reception side coil L ₂ can be controlled by the vertical insertion position of the magnetic plate 31C or the insertion area of the magnetic plate 31C. The magnetic plate 31C may be inserted not between the power transmission side coil L ₁ and the power reception side coil L ₂ but on the back side of the power transmission side coil L ₁ as shown in FIG. 7(c). The degree of magnetic coupling between the power transmission side coil _L1 and the power reception side coil _L2 can be controlled by the insertion position of the magnetic plate _31b in the vertical direction or the ratio of the insertion area of the magnetic plate 31b to the area of the power transmission side coil L1. Although illustration is omitted, it goes without saying that the value of the degree of magnetic coupling between the power transmission side coil _L1 and the power reception side coil _L2 can be similarly controlled by inserting a magnetic plate behind the power reception side coil _L2 .

Specifically, in order to control the insertion position in the vertical direction of the magnetic plate 31C shown in FIG. 7B and the insertion position in the vertical direction of the magnetic plate 31b shown in FIG. 7C, an optical distance measuring unit ₄₁ as shown in FIG. The distance measuring unit 41 may be provided with a magnetic coupling degree control mechanism having a light emitting section 411 and a light receiving section 412 . If the light receiving section 412 is an optical time-of-flight (TOF) distance measuring element d, the light emitting section 411 emits pulsed light. For example, a near-infrared LD (laser diode) or a near-infrared LED is used for pulsed light emission. The pulsed light reflected from the power receiving side coil _L2 and the rear part of the EV is irradiated onto the light receiving part 412 through a lens, a BPF (band pass filter), or the like. The distance measurement unit 41 may be configured as a laser interferometer or the like.

The light receiving section 412 of the distance measurement unit 41 is connected to the distance calculation section 421 of the logic operation control section 42 shown in FIG. 8(a). The output of the light receiving unit 412 is input to the distance calculation unit 421 that constitutes the magnetic coupling degree control mechanism via an output buffer and an interface (not shown), and the distance calculation unit 421 performs the calculation processing necessary for measuring the distance between the power transmission side coil L ₁ and the power reception side coil L ₂ . Connected to the logical operation control unit 42 is a data storage device 45 that stores data necessary for logical operations such as calculation of the degree of magnetic coupling in the logical operation control unit 42 and data on the insertion position in the vertical direction of the magnetic plate required for realizing a desired equivalent coupling coefficient (pseudo-coupling coefficient). Although not shown, the logic operation control unit 42 may be connected to a program storage device or the like storing a program for instructing the operation of the logic operation control unit 42 .

The data of the distance between the power transmission side coil L ₁ and the power reception side coil L ₂ calculated by the distance calculation section 421 is transmitted to the coupling coefficient calculation section 422 of the logical operation control section 42 . The coupling coefficient calculator 422 obtains the current magnetic coupling between the power transmitter coil L ₁ and the power receiver coil L ₂ from the data on the distance between the power transmitter coil L ₁ and the power receiver coil L ₂ calculated by the distance calculator 421. The coupling coefficient calculation unit 422 further calculates the movement distance of the magnetic plate from the data of the insertion position in the vertical direction of the magnetic plate necessary for realizing a desired equivalent coupling coefficient, which is stored in the data storage device 45, and outputs it to the coupling coefficient adjustment driving device 43. The coupling coefficient adjustment driving device 43 of the magnetic coupling degree control mechanism shown in FIG. 8(a) drives and controls the insertion position in the vertical direction of the magnetic plate 31C shown in FIG. 7(b) and the insertion position in the vertical direction of the magnetic plate 31b shown in FIG. A well-known position control mechanism such as a step motor can be employed for the coupling coefficient adjusting drive device 43 . In this manner, feedback control can be performed based on the output of the distance measuring unit 41 so that the insertion positions in the vertical direction of the magnetic plate 31C and the magnetic plate 31b are at desired positions.

In the computer system of the magnetic coupling degree control mechanism including the logical operation control unit 42 shown in FIG. 8(a), the data storage device 45 can be any combination appropriately selected from a group including multiple registers, multiple cache memories, a main storage device, and an auxiliary storage device. Also, the cache memory may be a combination of a primary cache memory and a secondary cache memory, or may have a hierarchy with a tertiary cache memory. The logical operation controller 42 shown in FIG. 8A can constitute a computer system using a microprocessor (MPU) or the like implemented as a microchip. In addition, as the logical operation control unit 42 constituting the computer system of the magnetic coupling degree control mechanism, a digital signal processor (DSP) with enhanced arithmetic operation function and specializing in signal processing, or a microcontroller (microcomputer) equipped with memory and peripheral circuits and intended for embedded device control may be used. Alternatively, the main CPU of current general-purpose computers may be used as the logical operation control unit 42 .

FIG. 9 is a diagram showing the operation of the mounted circuit shown in FIG. As shown in FIG. 9A, at the timing when the first semiconductor switching element Q1 as the primary side drive switch SW1 is turned on, electric charge is first stored in the power transmission side capacitor _C1 . As shown in FIG. 9A, the internal resistance r _on1 of the first semiconductor switching element Q1 at this time. At the timing of FIG. 9(a), when the terminal voltage _VC1 of the power transmission side capacitor _C1 starts to increase, part of the electric energy accumulated in the power transmission side capacitor _C1 is transferred to the power transmission side coil _L1 as the power transmission side coil current _IL1 and accumulated in the power transmission side coil _L1 . The electrical energy of the power transmission side coil L ₁ is transmitted to the power reception side coil L ₂ of the secondary side circuit 3 although it is slight. The electrical energy transmitted to the power receiving side coil _L2 of the secondary side circuit 3 is spent to charge the power receiving side capacitor C2 of the secondary side circuit ₃ as the power receiving side coil current _IL2 . However, at the timing of FIG. 9(a), the terminal voltage _VC2 of the power receiving side capacitor _C2 is a negative value.

Next, at the timing shown in FIG. 9B, when the first semiconductor switching element Q1 as the primary drive switch SW1 is turned off (OFF state), the _{voltage VC1} across the terminals of the power transmission side capacitor _C1 begins to decrease, and the electric energy accumulated in the power transmission side capacitor C1 is transferred to the power transmission side coil _L1 as the power transmission side coil current _IL1 , and accumulated in the power transmission side coil _L1 . The electrical energy stored in the power transmission side coil _L1 is wirelessly transmitted to the power reception side coil L2 of the secondary side circuit 3 by characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 between the primary side circuit 2 and the secondary side circuit ₃ . The electrical energy transmitted to the power receiving side coil L2 of the secondary side circuit ₃ is stored in the power receiving side capacitor _C2 of the secondary side circuit ₃ as the power receiving side coil current IL2. At the timing of FIG. 9(b), the terminal voltage _VC2 of the power receiving side capacitor _C2 becomes a positive value. After the voltage _VC2 across the terminals of the receiving side capacitor _C2 reaches a peak value, it starts to decrease.

When the terminal voltage _VC2 decreases, the electric energy accumulated in the power receiving capacitor _C2 flows into the load element 6 as the charging current _ICS , and the load element 6 is charged, as shown in FIG. 9(b). However, as the terminal voltage _VC2 decreases, part of the electrical energy stored in the power receiving side capacitor _C2 circulates in the power receiving side coil _L2 as a power receiving side coil current _IL2 , as _shown in FIG. The electrical energy accumulated in the power receiving side coil _L2 is circulated to the power transmitting side coil _L1 of the primary side circuit 2 by characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3, as shown in FIG. 9(c).

As shown in FIG. 9(c), the electric energy stored in the power transmission side coil _L1 begins to circulate to the power transmission side capacitor _C1 due to the power transmission side coil current _IL1 circulated to the power transmission side coil _L1 , and the terminal voltage _VC1 of the power transmission side capacitor _C1 starts to increase due to the circulating current. Therefore, as shown in FIG. 9(c), if an ammeter 461 for measuring the power transmission side coil current I _L1 circulated in the power transmission side coil _L1 and a voltmeter 462 for measuring the inter-terminal voltage _VC1 are provided in the primary side circuit 2, the magnitude of the electrical energy circulated from the secondary side circuit 3 can be measured. That is, the efficiency of wireless transmission from the primary circuit 2 to the secondary circuit 3 can be measured by the characteristic harmonic transmission between the primary circuit 2 and the secondary circuit 3 by using the ammeter 461 and the voltmeter 462 provided in the primary circuit 2.

Therefore, in order to configure a magnetic coupling degree control mechanism that controls the insertion position in the vertical direction of the magnetic plate 31C shown in FIG. 7B and the insertion position in the vertical direction of the magnetic plate 31b shown in FIG. 7C, a transmission efficiency measurement unit ₄₆ as shown in FIG. The transmission efficiency measurement unit 46 is an ammeter 461 and a voltmeter 462 provided in the primary circuit 2, as shown in FIG. 9(c).

The ammeter 461 and the voltmeter 462 shown in FIG. 9(c) are connected to the transmission efficiency calculator 471 of the logical operation controller 47 constituting the magnetic coupling degree control mechanism shown in FIG. 8(b). The outputs of the ammeter 461 and the voltmeter 462 are input to the transmission efficiency calculation unit 471 via an output buffer and an interface (not shown), and the transmission efficiency calculation unit 471 performs calculation processing necessary for measuring the transmission efficiency between the power transmission side coil L ₁ and the power reception side coil L ₂ . Connected to the logic operation control unit 47 is a data storage device 45 that stores data necessary for logic operations such as calculation of transmission efficiency in the logic operation control unit 47 and data on the insertion position in the vertical direction of the magnetic plate required to achieve desired transmission efficiency. Although not shown, the logic operation control unit 47 may be connected to a program storage device or the like that stores a program for instructing the operation of the logic operation control unit 47 .

The data of the transmission efficiency between the power transmission side coil L ₁ and the power reception side coil L ₂ calculated by the transmission efficiency calculation section 471 is transmitted to the coupling coefficient calculation section 472 of the logic operation control section 47 that constitutes the magnetic coupling degree control mechanism. The coupling coefficient calculator 472 obtains the current magnetic coupling between the power transmitter coil L ₁ and the power receiver coil L ₂ from the data of the transmission efficiency between the power transmitter coil L ₁ and the power receiver coil L ₂ calculated by the transmission efficiency calculator 471. The coupling coefficient calculation unit 472 further calculates the movement distance of the magnetic plate from the data of the insertion position in the vertical direction of the magnetic plate necessary for realizing the desired transmission efficiency, which is stored in the data storage device 45, and outputs it to the coupling coefficient adjustment driving device 43. The coupling coefficient adjustment driving device 43 drives and controls the insertion position in the vertical direction of the magnetic plate 31C shown in FIG. 7B and the insertion position in the vertical direction of the magnetic plate 31b shown in FIG. A well-known position control mechanism such as a step motor can be employed for the coupling coefficient adjusting drive device 43 . In this way, the magnetic coupling degree control mechanism shown in FIG. 8B can feedback-control the insertion positions in the vertical direction of the magnetic plate 31C and the magnetic plate 31b from the output of the transmission efficiency measurement unit 46 so that they are at desired positions.

As described in FIG. 8(a), the data storage device 45 forming part of the magnetic coupling degree control mechanism shown in FIG. 8(b) can be any combination appropriately selected from a group including multiple registers, multiple cache memories, a main storage device, and an auxiliary storage device. The logic operation control unit 47 shown in FIG. 8B can configure a computer system using an MPU or the like mounted as a microchip. Further, as the logical operation control unit 47 constituting the computer system, a DSP specialized for signal processing with enhanced arithmetic operation functions, a microcomputer equipped with a memory and peripheral circuits and intended for embedded device control, etc. may be used. Alternatively, the main CPU of current general-purpose computers may be used as the logical operation control section 47 .

Conventionally known “resonance” is a concept in which sinusoidal vibration of the primary side circuit 2 is transmitted to the freely vibrating secondary side circuit 3, and the secondary side circuit 3 vibrates at the same frequency as the primary side circuit 2. The power transmission device according to the first embodiment of the present invention includes the primary side drive switch SW1 that limits the free oscillation of the primary side circuit 2 and realizes a transient current-voltage change in the primary side circuit 2. Therefore, it is possible to transmit the non-sinusoidal sawtooth transient response characteristic to the secondary side circuit 3 by the concept of "characteristic harmonic transmission" proposed for the first time by the present inventors. By using a non-sinusoidal sawtooth transient response characteristic, a complicated and expensive AC power supply circuit for generating sinusoidal vibrations on the primary side circuit 2 side as in the prior art is no longer required.

As already described, in the power transmission device according to the first embodiment, the time constant inherent in the primary circuit 2 and the time constant inherent in the secondary circuit 3 are harmonized to transmit the electrical energy of the primary circuit 2 to the secondary circuit 3. For the characteristic harmonic transmission between the primary circuit 2 and the secondary circuit 3, for example, the capacities of the power transmission side capacitor _C1 and the power reception side capacitor _C2 may be equalized including the parasitic resistance of the capacitors, and the inductances of the power transmission side coil _L1 and the power reception side coil _L2 may be equalized including the parasitic resistance of the coils. Therefore, for example, if the ON/OFF repetition period of the primary side drive switch SW1 is set to 500 to 600 μs, the capacitance of the power transmission side capacitor C ₁ and the power reception side capacitor C ₂ may be set to be the same within the range of 400 μF to 600 μF, for example, and the inductance of the power transmission side coil L ₁ and the power reception side coil L ₂ may be set to be the same within the range of, for example, 5 μH to 20 μH.

A sawtooth-shaped transient response waveform as shown in FIGS. 1(b) and 2 includes a plurality of bumps in one cycle. It is extremely difficult to analytically solve the transient response waveform of the power transmission device according to the first embodiment. Therefore, although it is an approximation, the circuit shown in FIG. 28 is simulated by alternating current theory. In FIG. 28, the capacitance of the capacitor and the inductance of the coil of the primary side circuit 2 and the secondary side circuit 3 are set to the same value. That is, an approximate simulation is performed with C ₁ =C ₂ =500 μF and L ₁ =L ₂ =10 μH. At this time, the resonant frequency f _o1 of the RLC series circuit given by the equation (6) is 2.25 kHz, and the resonant frequency of the RLC series circuit given by the equation (7) is f ₀₂ =2.25 kHz. The corresponding repetition period is 444 μs.

When the primary side drive switch SW1 is turned on and there is a step input, current first flows through the transmitting side capacitor _C1 . The power transmission side coil L ₁ originally has the property of preventing a sudden inflow of current. The voltage of the power transmission side capacitor _C1 gradually rises, and the current gradually begins to flow through the power transmission side coil _L1 as well. In the meantime, the charge accumulated in the power transmission side capacitor _C1 also begins to flow out to the power transmission side coil _L1 side. When this happens, the voltage of the power transmission side capacitor C1 drops. Ideally, _the time until the primary drive switch SW1 is turned off is set so that the sum of the electrostatic energy (1/2) CV ² and the magnetic energy (1/2) LI ² is maximized _.

Care must be taken so that the voltage of the back electromotive force generated in the power transmission side coil _L1 does not exceed the withstand voltage of the first semiconductor switching element Q1 used for the primary side drive switch SW1 shown in FIG. The ON/OFF repetition period of the primary side drive switch SW1 is slightly earlier than the peak timing, considering the time until the terminal voltage VC1 of the power transmission side capacitor C1 rises again and reaches the peak.

FIG. 29 shows changes in the voltage of the power transmission side capacitor _C1 as an approximate simulation result. In FIG. 29, as shown in FIGS. 1(b) and 2, a sawtooth transient response waveform is obtained in the change in the voltage of the power transmission side capacitor _C1 . With reference to FIG. 30, it will be explained that sawtooth bumps occur because the current in the primary side circuit ₂ is reduced by the magnetic flux generated by the current induced in the power receiving side coil L2. As in the case of the mode using the four switches of the primary side drive switch SW1, the power transmission side switch SW2, the power reception side switch SW3, and the load control switch SW4, which will be described later in the fourth embodiment, when the voltage of the secondary side circuit 3 becomes maximum, the peak in the middle of the W-shaped transient response waveform shown in FIG.

The W-type transient response waveform shown in FIG. 34 will be described later with reference to FIG. An approximate simulation is performed using the circuit shown in FIG. 31, which is a circuit diagram simplified by omitting the parasitic capacitance and parasitic resistance from the circuit of FIG. It can be seen that the peak in the middle of the W-shaped transient response waveform enclosed by the dashed circles _A1 and _A2 in FIG. 32 appears as a bump in the sawtooth wave shown in FIGS.

In the case of the mode in which only the primary side drive switch SW1 is provided in the first embodiment, as shown in FIGS. 30(a) to 30(d), when the on/off repetition period of the primary side drive switch SW1 is lengthened, the transient response waveform becomes less bumpy and gradually approaches the sawtooth wave response waveform shown in FIGS. 1(b) and 2. FIG. 29 is an enlarged diagram showing the transient response waveform in the case of the repetition period of 575 μs shown in FIG. 30C, which corresponds to the transient response waveforms shown in FIGS. 1B and 2. FIG.

When the ON/OFF repetition period of the primary side drive switch SW1 is 565 μs, the waveform is a W-shaped transient response waveform as shown by the dashed circle _Aa in FIG. 30(a). Since the repetition period obtained from the resonance frequency of the RLC series circuit defined by the formulas (6) and (7) is 444 μs, the repetition period in FIG. That is, it can be seen that the power transmission device according to the first embodiment oscillates at a repetition period different from the resonance frequency of the RLC series circuit determined by the AC theory.

In the power transmission device according to the first embodiment, the amplitude is reduced by transmitting electrical energy from the primary side circuit 2 to the secondary side circuit 3 by characteristic harmonic transmission. When the repetition period is slightly increased to 570 μs in FIG. 30(b), the voltage at the valley on the right side of the shape of W showing the transient response waveform rises as indicated by the dashed circle A _b . Furthermore, bulges also begin to appear on the upper side of the sawtooth wave, as shown in FIGS. 1(b) and 2. FIG.

Further transmission of electrical energy from the primary side circuit 2 to the secondary side circuit 3 by characteristic harmonic transmission further reduces the amplitude. When the repetition period is further lengthened to 575 μs in FIG. 30(c), the voltage of the valley on the right side of the shape of W is further lifted as indicated by the dashed circle _Ac , resulting in a bumpy shoulder, and the shape of W disappears from the transient response waveform. Then, as shown in FIGS. 1(b) and 2, bumps also appear above the sawtooth wave.

Further transmission of electrical energy to the secondary side circuit 3 by characteristic harmonic transmission further reduces the amplitude, and when the repetition period is further lengthened to 580 _μs in FIG. 1(b) and 2, the bumps on the upper side of the sawtooth wave also become conspicuous, and two steps of bumps are shown. In this way, in the shape of W showing the transient response waveform, the amplitude gradually decreases, and when the repetition period is lengthened, the voltage in the valley on the right side does not gradually decrease, and the valley on the right side of the shape of W rises, resulting in a sawtooth-shaped transient response waveform having two bumps.

The reason why the voltage at the valley on the right side of the W shape rises when the ON/OFF repetition period of the primary side drive switch SW1 is lengthened is considered to be that the electrical energy received by the secondary side circuit 3 moves to the load element 6 to be charged. The maximum value of the current when the primary-side drive switch SW1 is turned on and the maximum value of the current flowing through the load element 6 depend on the parasitic inductance of the wires forming the actual circuit of the power transmission device according to the first embodiment. From analysis of waveforms measured in an actual circuit, the parasitic inductance is estimated to be about 1 mH to 3 μH. That is, it can be seen that the sawtooth-shaped transient response waveform having a plurality of bumps shown in FIGS.

Next, using the resonance frequency f _o1 of the RLC series circuit given by the conventional AC theory (6), ω ₀ =2πf ₀₁ , ω1=ω ₀ /(1−k) ^1/2 , and as shown in FIG.

f1=V3/[1+exp{10 ⁶ (t-2π/ω1)}]+V2 ……(8)

think of. Here, k defining ω1=ω ₀ /(1−k) ^1/2 is the coupling coefficient K _AC of the AC theory used to define the equation (1) (k=K _AC ).

Furthermore, for any decay function f2, a function that reduces with an appropriate decay constant τ:

f2=exp(-τt) ……(9)

think of. For example, in the power transmission device according to the first embodiment, the attenuation function f2 corresponds to the attenuation function of the maximum value of the voltage VC1 across the primary capacitor _C1 due to the time constant τ due to the parasitic resistance and the capacitor.

V1, V2 and V3 shown in FIG. 35 can all correspond to the voltage VC1 across the primary side capacitor _C1 . V1 can correspond to the voltage initially charged on the primary capacitor _C1 and V2 to the voltage remaining on the primary capacitor _C1 after the electrical energy has been transferred to the secondary circuit 3. V3 shown in FIG. 35 corresponds to their difference. The energy transfer function f1 is a function intended to cause the voltage VC1 across the primary side capacitor _C1 to drop from V1 to V2.

Using the above ω ₀ =2πf ₀₁ , ω2 = ω ₀ /(1+k) ^1/2 and the mutual induction function φ(k) is

φ(k)=cos(ω1t)+cos(ω2t) (10)

, the function (V1/2).phi.(k) shows a change as indicated by a thin broken line curve after the transfer timing of 2.pi./.omega.1 seconds in FIG. However, it should be noted that the coupling coefficient K of the power transmission device according to the first embodiment is a time-dependent parameter and is strictly different from the coupling coefficient K _AC of the AC theory. The thin dashed line in FIG. 35 is the waveform before the current is supplied to the load circuit 6, and corresponds to the waveform of the voltage VC1 across the primary capacitor _C1 . A thin dashed line after the transfer timing of 2π/ω1 seconds can be considered as a waveform when the energy of the primary side capacitor C1 is not transferred to the secondary side.

_The function (V2/2)φ(k) is the thin dashed curve up to the transfer timing of 2π/ω1 seconds in FIG. This corresponds to the waveform when it is assumed that the energy of the primary side capacitor C1 has been transferred to the secondary side from the beginning by the amount transferred to the secondary side. It can be seen that the function f1.f2..phi.(k) indicated by the solid line becomes the voltage VC1 across the capacitor _C1 , indicating the W type.

As described above, in the power transmission device according to the first embodiment, the amplitude is reduced by transmitting electrical energy to the secondary side circuit 3 by characteristic harmonic transmission. That is, as shown in FIG. 32, the trough on the right side of the W-shaped transient response waveform becomes smaller, gradually rises upward, and does not become depressed. This results in damping due to the dissipation of electrical energy through parasitic resistance, with a sawtooth-like overall RC time constant. An approximate simulation based on AC theory for the circuit shown in FIG. 28 is only an approximation, and there are limits to AC theory. Actually, only the experimental data shown in FIGS. 1B and 2 can explain the effect of the power transmission device according to the first embodiment.

That is, if the same capacitor is used for the power transmitting side capacitor _C1 and the power receiving side capacitor _C2 , and the same coil is used for the inductance of the power transmitting side coil _L1 and the power receiving side coil _L2 , the time constant inherent in the primary side circuit 2 and the time constant inherent in the secondary side circuit 3 can be harmonized by including the parasitic resistance in the coil and the capacitor.

As described above, the power transmission device according to the first embodiment of the present invention is a technology that does not rely on AC theory using the novel concept of characteristic harmonic transmission, so an inexpensive DC power supply 5 can be used. Therefore, the power transmission device according to the first embodiment does not require an expensive switching power supply, simplifies the circuit configuration, and minimizes power loss on the control circuit side. In particular, when a power semiconductor switching element is employed as the primary side drive switch SW1, only a simple ON/OFF control of the power semiconductor switching element is required, so power loss on the control circuit side is also reduced, and overall power transmission efficiency including loss in the power supply circuit (zero-order circuit) can be increased. In particular, since the circuit configuration is simplified, it is less likely to break and circuit design becomes easier. Also, the limit power of power transmission can be pushed up to a value that surpasses the limit power in the conventional AC theory. In principle, the limit power of power transmission can be increased to infinity, and in principle, the limit distance of power transmission can also be increased to infinity.

As a result, according to the power transmission device according to the first embodiment, the overall configuration of the power transmission device is simplified, the power loss on the control circuit side is minimized, the weight and size can be reduced, and the efficiency can be improved. Thus, the wireless power transmission device can be manufactured at low cost with improved overall power transmission efficiency due to power saving. In addition, since characteristic harmonic transmission can be realized with a repetition period longer than the repetition period required by the conventional AC theory, the frequency may be lower than in the case of multiple resonance in the conventional AC theory. Since a low-frequency circuit design is sufficient, it is easy to increase the voltage on the primary side circuit 2 side, and energy loss due to Joule heat generation can be reduced, so that the power transmission device according to the first embodiment can be manufactured at low cost with high overall power transmission efficiency. By lowering the parasitic resistance, in principle, a power transmission device can be manufactured in which the power transmission efficiency exceeds 99% and is increased to a value close to 100%.

(Second embodiment)
As shown in FIG. 10A, the power transmission device according to the second embodiment of the present invention has a configuration in which a power transmission side switch SW2 is added to the circuit configuration of the power transmission device according to the first embodiment shown in FIG. The “power transmission side switch SW2” is also a circuit element that limits the free oscillation of the primary side circuit 2 and realizes a transient current-voltage change in the primary side circuit 2, like the primary side drive switch SW1.

As the primary side drive switch SW1 and the power transmission side switch SW2 shown in FIG. 10(a), in addition to FETs, SITs, and BJTs similar to those of the power transmission device according to the first embodiment, power semiconductor switching elements including thyristors such as GTO thyristors and SI thyristors are used. In particular, voltage-driven switching elements such as MISFETs, MISSITs, IGBTs, and MOS-controlled SI thyristors are suitable for the primary side drive switch SW1 and the power transmission side switch SW2 because they reduce power consumption. Based on market availability and evaluation of the internal resistance of power semiconductor switching elements, it is currently possible to employ MOSFETs as the primary side drive switch SW1 and the power transmission side switch SW2 in the circuit shown in FIG. 10(b).

As already described in the power transmission device according to the first embodiment, a large amount of Joule heat is generated in a high-power power transmission device in which a rechargeable battery for EV is used as the load element 6 . In the power transmission device according to the second embodiment, only two power semiconductor switching elements are used as the primary side drive switch SW1 and the power transmission side switch SW2. Therefore, a cooling structure that prevents the elements from being destroyed by heat generation can be easily designed, and the occurrence of stray resistance, stray capacitance, and stray inductance can be minimized. In addition, since a simple ON/OFF control of the primary side drive switch SW1 and the power transmission side switch SW2 is all that is required, it is possible to simply increase the voltage of the primary side circuit 2 and suppress the generation of Joule heat.

In the implementation circuit shown in FIG. 10(b), a first freewheeling diode _FWD1 is connected in parallel between the source and drain of the MOSFET as the first semiconductor switching element Q1, and a second freewheeling diode _FWD2 is connected in parallel between the source and the drain of the MOSFET as the second semiconductor switching element Q2, as protective elements, in consideration of the circulating current from the power transmission side coil _L1 . As in the circuit shown in FIG. 4(a), a power supply diode D1 is connected in series between the DC power supply 5 and the first semiconductor switching element Q1 in order to prevent the circulating current from the power transmitting coil _L1 from flowing back to the DC power supply 5. In the mounting circuit shown in FIG. 10(b), the equivalent impedance X _Leq of the load element 6 is also expressed by approximating it with the charging capacity C _s .

The power transmission method according to the first embodiment will be described with reference to the timing diagram of FIG. 11 and the schematic time series diagrams of FIGS. 12(a) to 12(d). However, as in the first embodiment, it is assumed that the coupling coefficient K is equivalent to the coupling coefficient K _AC =0.6 derived from AC theory, and that the charging voltage _VCS in the initial state is a sufficiently high voltage close to full charge. First, at the timing shown in FIG. 12(a), the power transmission side switch SW2 is turned off, the primary side drive switch SW1 is turned on, and an initial voltage is applied to the power transmission side capacitor _C1 to store electric charge. Since the first semiconductor switching element Q1 is used as the primary side drive switch SW1 in FIG. 12(a), the ON state of the primary side drive switch SW1 is indicated by the ON resistance r _on1 of the first semiconductor switching element Q1. When the power transmission side switch SW2 is turned off, the primary side circuit 2 is not yet formed. As shown in FIG. 12(a), when the primary side drive switch SW1 is turned on, the power feeding side circuit 1 is formed by the DC power source 5, the equivalent internal resistance _r1 , the parallel circuit of the first semiconductor switching element Q1 and the first freewheeling diode (freewheeling diode) _FWD1 , and the series circuit of the power transmission side capacitor _C1 .

As indicated by the thin dashed line in FIG. 11, the terminal voltage _VC1 of the power transmission side capacitor _C1 is charged to a constant voltage while ringing. Although not shown in FIG. 11, FIG. 12(a) shows that the terminal voltage _VC2 of the receiving side capacitor C has a negative value at this timing. Next, at the timing shown in FIG. 12(b), the primary drive switch SW1 is turned off, and after a certain period of time, the power transmission side switch _SW2 is turned on. Then, the electric energy stored in the power transmission side capacitor C1 is accumulated in the power transmission side coil _L1 via the power transmission side coil current _IL1 , and characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 occurs. At the timing of FIG. 12(b), since the second semiconductor switching element Q2 is used for the power transmission side switch SW2, the ON state of the power transmission side switch SW2 is indicated by the ON resistance r _on2 of the second semiconductor switching element Q2. By turning on the power transmission side switch SW2, the primary side circuit 2 is formed, and the power feeding side circuit 1 consisting of the DC power supply 5, the equivalent internal resistance _r1 , the parallel circuit of the first semiconductor switching element Q1 and the first freewheeling diode (freewheeling diode) _FWD1 , and the power transmission side capacitor _C1 disappears.

When the electric energy stored in the power transmission side capacitor _C1 moves to the power transmission side coil _L1 , the inter-terminal voltage _VC1 indicated by the thin broken line in FIG. 11 takes a negative maximum value and then becomes 0V. The electrical energy transmitted to the power receiving side coil _L2 by characteristic harmonic transmission from the primary side circuit 2 to the secondary side circuit 3 charges the power receiving side capacitor _C2 with the power receiving side coil current _IL2 . When charging of the power receiving side capacitor _C2 starts, the voltage _VC2 across the terminals of the power receiving side capacitor C takes a maximum negative value as indicated by the thick dashed line in FIG. 11, and then becomes a positive value as shown in FIG. 12(c). It reaches its maximum value when the inter-terminal voltage _VC1 becomes 0V. As can be seen from the change of the thick dashed line in FIG. 11, the inter-terminal voltage _VC2 takes a negative maximum value, then becomes a positive value, and reaches its maximum value when the inter-terminal voltage _VC1 indicated by the thin dashed line reaches 0V.

At the timing shown in FIG. 12(c), as the inter-terminal voltage _VC2 increases, part of the electrical energy accumulated in the power receiving side capacitor C generates a charging current I _CS indicated by the dashed dotted line in FIG. Another part of the electric energy accumulated in the power receiving side capacitor C is returned to the power receiving side coil _L2 as the power receiving side coil current _IL2 . At the timing shown in FIG. 12(d), when the charging current _IC becomes 0, the inter-terminal voltage _VC2 becomes the same value as the charging voltage _VCS .

When the electrical energy stored in the power receiving capacitor C flows back to the power receiving coil _L2 , characteristic harmonic transmission occurs between the primary side circuit 2 and the secondary side circuit 3, and part of the electrical energy returns to the primary side circuit 2. When the electrical energy stored in the power receiving capacitor C moves to the load element 6 and the power receiving coil _L2 , the power receiving capacitor _C2 is discharged. When the receiving-side capacitor _C2 discharges, the inter-terminal voltage _VC2 indicated by the thick broken line on the right side of FIG. 11 becomes 0V after taking a negative maximum value. At this time, the inter-terminal voltage _VC1 indicated by the thin dashed line on the right side of FIG. 11 takes a negative maximum value, then increases to a positive value, and is maintained at a constant value when the power transmission side switch SW2 is turned off. 9(d), by measuring the inter-terminal voltage _VC1 , it is possible to monitor the transmission efficiency of the characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 and the charging status of the rechargeable battery as the load element 6. As can be seen from FIG. 11, the vibration waveform of the voltage _VC1 between the terminals of the primary side circuit 2 and the vibration waveform of the voltage _VC2 between the terminals of the secondary side circuit 3 are not symmetrical with each other.

As already mentioned, "resonance" is a concept that applies to systems in free vibration. In contrast, the power transmission device according to the second embodiment of the present invention includes a power transmission side switch SW2 and a primary side drive switch SW1 that limit the free oscillation of the primary side circuit 2 and realize transient current-voltage changes in the primary side circuit 2. Therefore, in the power transmission device according to the second embodiment, it is possible to transmit non-sinusoidal transient response characteristics to the secondary side circuit 3 by the new concept of “characteristic harmonic transmission”. Since the configuration of the control circuit is simple and non-sinusoidal transient response characteristics based on the inexpensive DC power supply 5 can be used, an expensive AC power supply circuit for generating sinusoidal vibration higher than the commercial frequency for the primary side circuit 2 as in the conventional art is not required, and the circuit design is difficult to break and easy.

Like the circuit shown in FIG. 28, the circuit shown in FIG. 33 is the circuit of the power transmission device according to the second embodiment in the case of approximate simulation based on alternating current theory, but it includes two switches, a primary side drive switch SW1 and a power transmission side switch SW2. In FIGS. 33A to 33C, the capacitance of the capacitor and the inductance of the coil of the primary side circuit 2 and the secondary side circuit 3 are set to be the same. That is, C ₁ =C ₂ =500 μF and L ₁ =L ₂ =10 μH.

FIG. 33(a) is a circuit of the power transmission device according to the second embodiment in which the voltage E ₀ of the DC power supply 5 is 36 V, the electrical energy of the primary side circuit 2 is transmitted to the secondary side circuit 3 by characteristic harmonic transmission, and the charging voltage of the rechargeable battery employed as the load circuit 36 is V _cs =24 V. FIG. 33(b) shows a circuit in which the same voltage E ₀ =36 V of the DC power supply 5 as in FIG. 33(a) is used, and the electrical energy _of the primary side circuit 2 is transmitted to the secondary side circuit 3 by characteristic harmonic transmission. FIG. 33(b) shows a case where the charging voltage V _cs of the rechargeable battery employed as _the load circuit 36 is set to 100 V, similar to FIG.

FIG. 34 shows changes in the voltage of the power transmission side capacitor C1 as an approximate simulation result. As shown in FIG. 32, in the approximate simulation based on the AC theory of the power transmission device according to the second embodiment, the change in the voltage of the power transmission side capacitor C1 shows a W-shaped transient response waveform similar to that shown in FIG. The solid line in FIG. 34 is the approximate simulation result for the circuit shown in FIG. 33(a), the dashed line in FIG. 34 is the approximate simulation result for the circuit shown in FIG. 33(b), and the one-dot chain line in FIG.

At first, as indicated by the dashed line in FIG. 34, an attempt is made to pass a current through the load circuit 36, but since the charging voltage of the rechargeable battery is set to V _cs =100 V, the current does not flow through the load circuit 36, resulting in a change like the curve indicated by the dashed line. The timing at which the current is to flow through the load circuit 36 is presumed to be around the peak position in the center of the W-shaped transient response waveform in FIG.

As described above, according to the power transmission device according to the second embodiment, similarly to the power transmission device according to the first embodiment, it is possible to use the DC power supply 5 that has a simple control circuit and peripheral circuits and is inexpensive. Therefore, an expensive switching power supply is unnecessary. The circuit configuration of the power transmission device according to the second embodiment is simplified, the power loss on the control circuit side is minimized, the device is less likely to break, and circuit design is facilitated. As a result, the overall configuration of the power transmission device is simplified, making it possible to make it lighter, smaller, and more efficient, and it is possible to inexpensively manufacture a wireless power transmission device with improved overall power transmission efficiency including the loss of the power supply circuit (zero-order circuit). As described in the power transmission device according to the first embodiment, the limit power of power transmission is raised to a value that surpasses the limit power in the conventional AC theory, and is pushed to infinity in principle, and the limit distance of power transmission can also be extended to infinity in principle. Furthermore, it is possible in principle to increase the power transmission efficiency to a value close to 100%.

(Third Embodiment)
As shown in FIG. 13A, the power transmission device according to the third embodiment of the present invention has a configuration in which a power receiving side switch SW3 is added to the power transmission device according to the second embodiment. The “receiving side switch SW3” is also a circuit element that limits the free oscillation of the secondary side circuit 3 and realizes a transient current-voltage change in the secondary side circuit 3, like the power transmitting side switch SW2 and the primary side drive switch SW1.

As the primary side drive switch SW1, the power transmission side switch SW2, and the power reception side switch SW3 shown in FIG. 13A, power semiconductor switching elements including thyristors such as GTO thyristors and SI thyristors are used in addition to FETs, SITs, and BJTs similar to those of the power transmission devices according to the first and second embodiments. From the demand for low internal resistance and availability in the market, it is industrially advantageous to adopt MOSFETs as the primary side drive switch SW1, the power transmission side switch SW2, and the power reception side switch SW3 of the mounting circuit shown in FIG. 13(b).

As described in the power transmission devices according to the first and second embodiments, a large amount of Joule heat is generated in the high-power power transmission device. In the power transmission device according to the third embodiment, only three power semiconductor switching elements are used as the primary side drive switch SW1, the power transmission side switch SW2, and the power reception side switch SW3. Therefore, a cooling structure that prevents the elements from being destroyed by heat generation can be easily designed, and the occurrence of floating resistance, floating capacitance, and floating inductance can be minimized. In addition, since a simple ON/OFF control of the primary side drive switch SW1 and the power transmission side switch SW2 is all that is required, it is possible to simply increase the voltage of the primary side circuit 2 and suppress the generation of Joule heat.

In the implementation circuit shown in FIG. 13(b), a first freewheeling diode FWD ₁ is connected in parallel between the source and drain of the MOSFET as the first semiconductor switching element Q1, a second freewheeling diode FWD ₂ is connected in parallel between the source and the drain of the MOSFET as the second semiconductor switching element Q2, and a third freewheeling diode FWD ₃ is connected in parallel between the source and the drain of the MOSFET as the third semiconductor switching element Q3. As shown in FIG. 13(b), the third freewheeling diode _FWD3 is provided in the direction in which the freewheeling current flows from the receiving side coil _L2 , so the second freewheeling diode _FWD2 is provided in the opposite direction. As in the circuits shown in FIGS. 4A and 10B, a power supply diode D1 is connected in series between the DC power supply 5 and the first semiconductor switching element Q1 in order to prevent the circulating current from the power transmission coil _L1 from circulating to the DC power supply 5. In the mounting circuit shown in FIG. 13(b), the equivalent impedance X _Leq of the load element 6 is also expressed by approximating it with the charging capacity C _s .

A wireless power transmission method according to the first embodiment will be described with reference to the flowchart shown in FIG. 14 and the timing chart shown in FIG. However, as in the first and second embodiments, characteristic harmonic transmission is assumed under conditions corresponding to the coupling coefficient K _AC =0.6 according to AC theory, and the value of the charging voltage _VCS in the initial state is a sufficiently high voltage close to full charge.

First, in step S31 of the flowchart of FIG. 14, the power transmission side switch SW2 and the power reception side switch SW3 are turned off, and only the primary side drive switch SW1 is turned on. As indicated by the thin dashed line in FIG. 15, the terminal voltage _VC1 of the power transmission side capacitor _C1 is charged to a constant voltage while ringing. Although not shown in FIG. 15, the terminal voltage _VC2 of the power receiving side capacitor C is a negative value at this timing. After an initial voltage is applied to the transmission-side capacitor _C1 to store electric charge, the primary-side drive switch SW1 is turned off as shown in FIG. As mentioned above, it is assumed that the charging voltage _VCS is high at this point.

As shown in FIG. 15, after the primary drive switch SW1 is turned off, the power transmission side switch SW2 and the power reception side switch SW3 are simultaneously turned on in step S32 after a certain period of time. When the power transmission side switch SW2 is turned on, the electric energy stored in the power transmission side capacitor _C1 is accumulated in the power transmission side coil _L1 via the power transmission side coil current, and further, the characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 occurs. When the electric energy stored in the power transmission side capacitor _C1 moves to the power transmission side coil _L1 , the inter-terminal voltage _VC1 indicated by the thin broken line in FIG. 15 takes a negative maximum value and then becomes 0V. The electrical energy transmitted to the power receiving side coil _L2 by characteristic harmonic transmission from the primary side circuit 2 to the secondary side circuit 3 charges the power receiving side capacitor _C2 with the power receiving side coil current because the power receiving side switch SW3 is in the ON state. When charging of the power receiving side capacitor _C2 starts, the voltage _VC2 across the terminals of the power receiving side capacitor C starts to increase from a negative maximum value as indicated by the thick dashed line in FIG. While the terminal voltage _VC2 takes a negative value, the charging current _ICS does not flow, but when the terminal voltage _VC2 becomes a positive value, the charging current _ICS starts to rise as indicated by the dashed-dotted line in the center of FIG.

At the timing when the charging current _ICs starts to rise, the power transmitting side switch SW2 and the power receiving side switch SW3 are turned off in step S33. The OFF state of the power transmission side switch SW2 and the power reception side switch SW3 is the time when _the terminal voltage _VC2 indicated by the thick broken line in FIG. The charging current _IC indicated by the dashed line in the center of FIG. 15 increases even after the power transmission side switch SW2 and the power reception side switch SW3 are turned off, reaches a peak value, then decreases, and becomes zero in step S34.

As shown in the right side of the center of FIG. 15, the maximum value of the terminal voltage _VC2 indicated by the thick dashed line becomes a slightly lower constant value when the charging current _IC starts to decrease, forming a stepped (shouldered) waveform. Even after the charging current _IC becomes zero, the terminal voltage _VC2 indicated by the thick broken line in FIG. 15 maintains a value lower than the maximum value when the power transmission side switch SW2 is turned off. After the power transmission side switch SW2 is turned off, the maximum value of the terminal voltage _VC2 decreases after a certain period of time has elapsed. However, if the charging voltage _VCS is high at the time of step S31, the amount of decrease in the maximum value of the terminal voltage _VC2 due to the charging current _IC is small, and the effect on the characteristic harmonic transmission between the primary circuit 2 and the secondary circuit 3 is small.

In step S35 after the charging current _IC reaches 0 A, the power transmitting side switch SW2 and the power receiving side switch SW3 are turned on again at the same time. Due to the ON state of the power transmission side switch SW2 and the power reception side switch SW3 in step S35, the inter-terminal voltage _VC2 indicated by the thick dashed line on the right side of FIG. At this time, the inter-terminal voltage _VC1 indicated by a thin dashed line on the right side of FIG. 15 also starts to decrease, reaches a maximum negative value, and then increases to a positive value.

Next, in step S36, when the inter-terminal voltage _VC1 becomes maximum and the inter-terminal voltage _VC2 becomes 0V, the power transmitting side switch SW2 and the power receiving side switch SW3 are turned off. As shown in FIG. 15, the inter-terminal voltage _VC1 indicated by the thin dashed line at step S36 has the same value as the inter-terminal voltage VC2 indicated by the thick dashed line at step S34, and is maintained at a constant value after the power transmitting side switch _SW2 and the power receiving side switch SW3 are turned off. be. Although not shown, the terminal voltage _VC1 and the terminal voltage of the load element 6 have the same value at step S36. For this reason, by measuring the terminal voltage _VC1 , it is possible to monitor the transmission efficiency of the characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 and the charging status of the rechargeable battery as the load element 6, in the same manner as described with reference to FIG. 9(d). Comparing the timing chart of the power transmission device according to the second embodiment shown in FIG. 11 with the timing chart of the power transmission device according to the third embodiment shown in FIG. 15, it can be seen that even if one switch SW3 on the power receiving side is added, the waveforms showing temporal changes (transient response) such as the inter-terminal voltage VC ₁ and the inter-terminal voltage VC ₂ in the characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 are almost the same.

"Resonance" is a concept used in an AC circuit that freely oscillates, but the power transmission device according to the third embodiment includes a primary side drive switch SW1, a power transmission side switch SW2, and a power reception side switch SW3 that limit the free vibration of the primary side circuit 2 and the secondary side circuit 3 and realize transient current-voltage changes in the primary side circuit 2 and the secondary side circuit 3. Therefore, in the power transmission device according to the third embodiment, the transient response characteristics of the primary side circuit 2 can be transmitted to the secondary side circuit 3 by the new concept of "characteristic harmonic transmission." Since electrical energy can be transmitted using a non-sinusoidal transient response characteristic based on a DC power supply 5 which has a simple configuration and is inexpensive, an expensive AC power supply circuit for generating sinusoidal vibration higher than the commercial frequency is not required for the primary side circuit 2. - 特許庁

Therefore, according to the power transmission device according to the third embodiment of the present invention, similarly to the power transmission devices according to the first and second embodiments, the DC power supply 5, which has a simple control circuit and peripheral circuits and is inexpensive, can be used. Therefore, an expensive switching power supply is unnecessary, the circuit configuration is simplified, and the power loss on the control circuit side is minimized. As a result, according to the power transmission device according to the third embodiment, the overall configuration of the power transmission device is simplified, making it possible to reduce the weight, size, and efficiency of the power transmission device, and it is possible to inexpensively manufacture a wireless power transmission device with improved overall power transmission efficiency including the loss of the power supply circuit (zero-order circuit). As in the power transmission devices according to the first and second embodiments, the power transmission device according to the third embodiment simplifies the circuit configuration, making it difficult to break and easy to design the circuit. In addition, it is possible to raise the limit power of power transmission to infinity in principle, extend the limit distance of power transmission to infinity in principle, and increase the power transmission efficiency to a value close to 100% in principle.

(Fourth embodiment)
As shown in FIG. 16A, the power transmission device according to the fourth embodiment of the present invention has a configuration in which a load control switch SW4 is added to the power transmission device according to the third embodiment. The “load control switch SW4” is a circuit element that limits the free oscillation of the secondary circuit 3 and realizes a transient current-voltage change in the secondary circuit 3, like the power receiving switch SW3.

As the primary side drive switch SW1, the power transmission side switch SW2, the power reception side switch SW3, and the load control switch SW4 shown in FIG. 16A, power semiconductor switching elements including thyristors such as FETs, SITs, and BJTs, such as GTO thyristors and SI thyristors, can be used as in the power transmission devices according to the first to third embodiments. Considering the requirement for low internal resistance, it is preferable to employ MOSFETs as the primary side drive switch SW1, the power transmission side switch SW2, the power reception side switch SW3, and the load control switch SW4 of the mounting circuit shown in FIG.

In the power transmission device according to the fourth embodiment, only four power semiconductor switching elements are used as the primary side drive switch SW1, the power transmission side switch SW2, the power reception side switch SW3, and the load control switch SW4. Therefore, a cooling structure that prevents the generation of Joule heat can be easily designed, and the generation of stray resistance, stray capacitance, and stray inductance can be minimized. In addition, since simple ON/OFF control of the primary side drive switch SW1 and the power transmission side switch SW2 is sufficient, it is possible to increase the voltage of the primary side circuit 2 to suppress the generation of Joule heat.

In the implementation circuit shown in FIG. 16(b), the first freewheeling diode FWD ₁ is connected between the source and drain of the MOSFET as the first semiconductor switching element Q1, the second freewheeling diode FWD ₂ is connected between the source and drain of the MOSFET as the second semiconductor switching element Q2, the third freewheeling diode FWD ₃ is connected between the source and drain of the MOSFET as the third semiconductor switching element Q3, and the fourth freewheeling diode MOSFET MOSFET ₄ is connected between the source and drain of the MOSFET as the fourth semiconductor switching element Q4. They are connected in parallel as protective elements between the source and drain of the FET. As shown in FIG. 16(b), the third freewheeling diode FWD ₃ is provided in the direction in which the freewheeling current from the receiving side coil L ₂ flows, so that the second freewheeling diode FWD ₂ is provided in the opposite direction as in FIG. 13(b). 4(a), 10(b) and 13(b), in order to prevent the circulating current from the power transmission side coil _L1 from circulating to the DC power supply 5, the power supply side diode D1 is connected in series between the DC power supply 5 and the first semiconductor switching element Q1. In the mounting circuit shown in FIG. 16(b), the equivalent impedance X _Leq of the load element 6 is also expressed by approximating the charge capacity C _s .

As already described, when comparing the timing diagram of the power transmission device according to the second embodiment shown in FIG. 11 and the timing diagram of the power transmission device according to the third embodiment shown in FIG. 15, even if one switch SW3 on the power receiving side is added, the waveforms showing temporal changes (transient response) such as the inter-terminal voltage VC ₁ and the inter-terminal voltage VC ₂ in the characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 are almost the same. As shown in FIG. 16(a), even if the load control switch SW4 is added to the power transmission device according to the third embodiment, the essence of the characteristic harmonic transmission remains unchanged, and the basic operation and the waveforms showing temporal changes (transient response) of the inter-terminal voltage _VC1 , the inter-terminal voltage _VC2 , etc. in the characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 are almost the same.

However, as shown in FIG. 16A, in the configuration having four switches, ie, the primary side drive switch SW1, the power transmission side switch SW2, the power reception side switch SW3, and the load control switch SW4, at the timing when the primary side drive switch SW1 and the load control switch SW4 are cut off and the power transmission side switch SW2 and the power reception side switch SW3 are made conductive, the circuit on the DC power supply 5 side and the circuit on the load element 6 side are separated from the primary side circuit 2 and the secondary side circuit 3, respectively. It becomes possible for the secondary side circuit 3 to vibrate freely. That is, since the LC resonant circuit of the primary side circuit 2 and the LC resonant circuit of the secondary side circuit 3 can be treated as a circuit coupled by mutual inductance M, the idea of multiple resonance in AC theory can be adopted. That is, at the timing when the primary-side drive switch SW1 and the load control switch SW4 are in the cut-off state, and the power-transmitting-side switch SW2 and the power-receiving-side switch SW3 are in the conductive state, the efficiency of characteristic harmonic transmission can be examined by the coupling equations (2) and (3) already described.

However, in the implementation circuit, it is necessary to consider the ON resistance of the power semiconductor switching elements used in the four switches of the primary side drive switch SW1, the power transmission side switch SW2, the power reception side switch SW3, and the load control switch SW4. Therefore, in the operation at the timing when the primary side drive switch SW1 and the load control switch SW4 are cut off and the power transmission side switch SW2 and the power receiving side switch SW3 are made conductive, it is necessary to consider a circuit in which the LCR resonance circuit of the primary side circuit 2 and the LCR resonance circuit of the secondary side circuit 3 are coupled by mutual inductance M.

In addition, since it is necessary to consider energy transmission in a transient response such as a step response when the primary side drive switch SW1 and the load control switch SW4 are turned on, not all the power transmission devices according to the fourth embodiment of the present invention can be interpreted by the conventional AC theory. That is, in the region of free vibration shown by hatching in FIG. 3B, the conventional sine wave AC theory can be used, but in the operating environment of the power transmission device according to the fourth embodiment in which the boundary conditions of the circuit are constantly changed using the four switches of the primary side drive switch SW1, the power transmission side switch SW2, the power reception side switch SW3, and the load control switch SW4, it is necessary to analyze including the transient response such as the sawtooth wave-like rise characteristic as illustrated in FIG. 2B.

(Simulation of characteristic harmonic transmission waveform)
The waveforms of the terminal voltage VC ₁ and the terminal voltage VC ₂ of the power transmission side capacitor C ₁ and the power reception side capacitor C ₂ in the configuration illustrated in FIG. The coupling coefficient K _AC between the primary side circuit 2 and the secondary side circuit 3 is 0.6, the capacitances of the power transmitting side capacitor _C1 and the power receiving side capacitor _C2 are both 65 μF, and the inductances of the power transmitting side coil _L1 and the power receiving side coil _L2 are both 60 μH.

First, the power transmission side switch SW2, the power reception side switch SW3, and the load control switch SW4 are turned off, the primary side drive switch SW1 is turned on, and an initial voltage of 20 V is applied to the power transmission side capacitor _C1 to store electric charge in the power transmission side capacitor _C1 . Next, when the primary side drive switch SW1 is turned off and the power transmission side switch SW2 and the power reception side switch SW3 are turned on, it can be expected that characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 will occur. FIG. 17A shows the waveforms of the terminal voltage _VC1 and the terminal voltage _VC2 obtained by simulation. Both the waveform of the terminal voltage _VC1 and the waveform of the terminal voltage _VC2 are waveforms obtained by synthesizing a large amplitude sine wave and a small amplitude sine wave, which are different from the sine waves in ordinary AC theory.

In FIG. 17A, the power transmission side switch SW2 and the power reception side switch SW3 are turned on at 0.2 ms. At 0.45ms, the voltage across terminal _VC1 becomes 0V and the voltage across terminal _VC2 becomes 20V. This indicates that all the energy on the power transmission side is transmitted to the power reception side, and when the power transmission side switch SW2 and the power reception side switch SW3 are turned off at 0.45 ms, power can be efficiently transmitted by characteristic harmonic transmission.

(Measurement of characteristic harmonic transmission waveform by mounting circuit)
Subsequently, the waveform of the characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 is measured using the mounted circuit having the configuration illustrated in FIG. 16(a). The equivalent coupling coefficient K of the primary side circuit 1 and the secondary side circuit 2 is 0.6, the capacitance of the power transmission side capacitor C ₁ and the power reception side capacitor C ₂ are both 65 μF, the inductance of the power transmission side coil L ₁ and the power reception side coil L ₂ are both 60 μH, and the initial voltage applied to the power transmission side capacitor C ₁ is 20 V. These are the same values as in the simulation. _Waveforms of the terminal voltage _VC1 and the terminal voltage _VC2 of the power transmission side capacitor _C1 and the power reception side capacitor _C2 obtained by measurement are shown in FIG _.

Since there is a parasitic resistance in the mounted circuit, the waveform attenuates with time unlike the result of the simulation based on the normal AC theory shown in FIG. 17(a). In FIG. 17(b), transmission starts at 0.2 ms, and the inter-terminal voltage _VC2 reaches the maximum of 15 V at 0.45 ms. At this time, the terminal voltage VC ₁ is −3 V. Not all of the energy on the power transmission side is transmitted to the power reception side, and some energy remains on the power transmission side, but it can be confirmed that the energy on the power transmission side is transmitted to the power reception side. As already described, even if the load control switch SW4 is added to the power transmission device according to the third embodiment, the waveforms showing temporal changes (transient response) such as the terminal voltage _VC1 and the terminal voltage _VC2 are almost the same. That is, the waveforms of the terminal voltage _VC1 and the terminal voltage _VC2 shown in FIG. 17(b) are diagrams showing macroscopic changes. Macroscopically, the waveforms look like a combination of a large-amplitude sine wave and a small-amplitude sine wave.

(Change in equivalent coupling coefficient and change in characteristic harmonic transmission)
In the configuration exemplified in FIG. 16(a), when the primary side drive switch SW1 and the load control switch SW4 are turned off, and the power transmission side switch SW2 and the power receiving side switch SW3 are turned on after storing electric charge in the power transmission side capacitor _C1 , characteristic harmonic transmission occurs between the primary side circuit 2 and the secondary side circuit 3. According to conventional AC theory, this operation is represented by the coupling equations of equations (2) and (3) already mentioned.

In the conventional AC theory, solving the coupled equations of Equations (2) and (3) to find the voltage _VC2 across the terminals of the receiving side capacitor C2 generated by the characteristic harmonic transmission between the primary circuit ₂ and the secondary circuit 3, using the mutual induction function φ(k) defined by Equation (10),

_VC2 = _VC0 /2*( _L2 / _L1 ) ^(1/2) *φ(k) (11)

becomes. Here, VC ₀ is the initial voltage across the terminals of the power transmission side capacitor C ₁ , ω ₀ is the resonance angular frequency, and ω ₀ =L ₁ ×C ₁ =L ₂ ×C ₂ . When the voltage _VC1 between the terminals of the capacitor on the transmitting side is 0, the maximum value _VC0 ×( _L2 / _L1 ) ^(1/2) is obtained in equation (11), and according to the normal AC theory, all the energy on the transmitting side is transmitted to the receiving side at this time.

In the configuration illustrated in FIG. 16(a), the change in the waveform of the characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 when the coupling coefficient K _AC according to the normal AC theory is changed is obtained by simulation based on the normal AC theory. The capacitance of the power transmission side capacitor _C1 and the power reception side capacitor _C2 are both 500 μF, the inductance of the power transmission side coil _L1 and the power reception side coil _L2 are both 10 μH, and the initial voltage applied to the power transmission side capacitor _C1 is 25V. In the following explanation, it is approximated that the coupling coefficient K _AC according to the AC theory is equal to the equivalent coupling coefficient, and simulations are performed according to the usual AC theory with the equivalent coupling coefficient K set to 0.00, 0.1, 0.6, 0.8, and 0.88. Waveforms of the terminal voltage _VC1 and the terminal voltage _VC2 obtained as a result of simulation based on the ordinary AC theory are shown in FIGS. 18(a) to 19(c). As shown in FIG. 18(a), when the coupling coefficient K=0.00 according to the normal AC theory, the primary circuit 1 and the secondary circuit do not interact with each other, and the characteristic harmonic transmission between the primary circuit 2 and the secondary circuit 3 does not occur.

When the equivalent coupling coefficient K=0.1, 0.6, 0.8 and 0.88, characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 occurs. 2.2 ms when the equivalent coupling coefficient K=0.1 as shown in FIG. 18(b), 0.28 ms when the equivalent coupling coefficient K= _0.6 as shown in FIG. has the same value as the initial _voltage VC ₀ between , and all the energy on the transmitting side is transmitted to the receiving side _. As shown in FIG. 19B, when the equivalent coupling coefficient K=0.8, the maximum value of the voltage _VC2 between the terminals of the power receiving side capacitor is smaller than the initial voltage _VC0 between the terminals of the power transmitting side capacitor _C1 . Compared to when the equivalent coupling coefficients K=0.1, 0.6, and 0.88, when the equivalent coupling coefficient K=0.8, power transmission cannot be performed efficiently.

Further, when the equivalent coupling coefficient K=0.1, the time required for the terminal-to-terminal voltage _VC2 of the receiving side capacitor to reach its maximum value is longer than when the equivalent coupling coefficient K=0.6 and 0.88. Equation (11) is expressed as the sum of two modes,

(1+k) ^(1/2) /(1-k) ^(1/2) =2 ……(12)

, that is, when the equivalent coupling coefficient K=0.6, the time until the voltage _VC2 across the terminals of the power receiving side capacitor _C2 takes the maximum value is the shortest, and the next shortest is

(1+k) ^(1/2) /(1-k) ^(1/2) = 4 ……(13)

, that is, when the equivalent coupling coefficient K=0.88. In the mounted circuit, the waveform attenuates with time due to the parasitic resistance r _L = R _str (L ₁ ) = R _str (L ₂ ) of the coil and the parasitic resistance r _C of the capacitor, so the equivalent coupling coefficient K = 0.6, 0.88. Also, when the parasitic resistances r _L and r _C are low, power can be efficiently transmitted even when the equivalent coupling coefficient K=0.1. As the distance d increases, the equivalent coupling coefficient K decreases. Therefore, if the parasitic resistances r _L and r _C are sufficiently low, it can be said that they can be sent over a long distance.

FIG. 20(a) shows enlarged waveforms of the terminal voltage _VC1 and the terminal voltage _VC2 when the equivalent coupling coefficient K=0.6 shown in FIG. 19(a). FIG. 20(b) shows the waveforms of the currents _I1 and _I2 of the power transmission side coil _L1 and the power reception side coil _L2 at this time. At 0.28 ms, the voltage _VC1 across the terminals becomes 0V, and the voltage across the terminals _VC2 becomes the same value as the initial voltage _VC0 across the terminals of the power transmitting capacitor _C1 , and at the same time, the currents _I1 and _I2 become 0A. When the power transmission side switch SW2 and the power reception side switch SW3 are turned off at 0.28 ms, power transmission can be performed with maximum efficiency. Further, at this time, the currents _I1 and _I2 are 0 A, and the back electromotive force generated in the power transmission side coil _L1 and the power reception side coil _L2 becomes 0, so that the power transmission side switch SW2 and the power reception side switch SW3 can be prevented from being destroyed.

(Optimal combination of inductance L and capacitance C)
In the configuration illustrated in FIG. 16( a ), the combination of the inductance L of the coil and the capacitance C of the capacitor with the highest transmission efficiency when transmitting the energy stored on the power transmitting side to the power receiving side by characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 is obtained by the following procedure. The transmission efficiency P is expressed by equation (14), where P _on C _e-tr is the energy to be transmitted at one time, P _on C _e-loss is the energy lost at one time due to the parasitic resistance r _L of the coil and the parasitic resistance r _C of the capacitor, and τ _on C _e is the time required for one transmission.

P= _{( PonCe -tr} - _{PonCe -loss} )/ _τonCe (14)

The energy P _on C _e-tr to be transmitted in one transmission is 1/2×CV ₀ ² , the energy P _on C _e-loss lost in one transmission is (r _L +r _C )/2×C/L×V ₀ ² , and the time τ _{on C} e required for one transmission is 2(1.6) ^1/2 π(LC) ^1/2 .

P=(1/2×CV ₀ ² −(r _L +r _C )/2×C/L×V ₀ ² )/(2(1.6) ^1/2 π(LC) ^1/2 ) (15)

Here, V ₀ is the initial voltage applied to the power transmission side capacitor C ₁ .

Let K _L =r _L /L and K _C =r _C × _C for _the parasitic resistance r _L of the coil and the parasitic resistance r C of the capacitor, and let I _max be the ^maximum current flowing through the coil.

P=I _max V {(1.6) ^−1/2 π ⁻¹ −(K _L (LC) ^1/2 +K _C (LC) ^−1/2 )}
…………(16)

becomes,

K _L (LC) ^1/2 + K _C (LC) ^−1/2 >= 2(K _L K _C ) ^1/2 (17)

is.

When the transmission efficiency P is maximized

K _L (LC) ^1/2 + K _C (LC) ^−1/2 = 2(K _L K _C ) ^1/2 (18)

and at this time

LC= _KC / _KL (19)

becomes. The transmission efficiency is maximized when the inductance L of the coil and the capacitance C of the capacitor satisfy the expression (19).

In the configuration exemplified in FIG. 16A, the transmission efficiency when changing the inductance of the coil and the capacitance of the capacitor is obtained by a simulation based on ordinary alternating current theory. It is assumed that V=36 V, coupling coefficient K _L =600Ω/H, and coupling coefficient K _C =3.00×10 ⁻⁶ ΩF. FIG. 21 shows changes in transmission efficiency with respect to capacitor capacitance when coil inductances are 1, 2, 5, 10, 20 and 50 μH, obtained as a result of simulation based on ordinary alternating current theory. As shown in FIG. 21, when the coil inductance is 1, 2, 5, 10, 20, and 50 μH, the capacitor capacitance C is 5000, 2500, 1000, 500, 250, and 100 μF, respectively, and the combination of the coil inductance L and the capacitor capacitance C that maximizes the transmission efficiency satisfies the equation (19).

(Power transmission when the voltage between the terminals of the load element is low)
The first wireless power transmission method according to the fourth embodiment when the voltage across the terminals of the load element 6 as a rechargeable battery is low will be described with reference to the flowchart shown in FIG. 23 and the timing chart shown in FIG. However, when characteristic harmonic _transmission between the primary circuit 2 and the secondary circuit 3 occurs, the equivalent coupling coefficient K is adjusted such that the maximum value of the terminal voltage VC ₂ becomes the same as the initial voltage VC ₀ between the terminals of the power transmission side capacitor C ₁ , and the voltage VC ₁ between the terminals becomes 0 V at that time.

First, in step S11, the power transmission side switch SW2, the power reception side switch SW3, and the load control switch SW4 are turned off, and the primary side drive switch SW1 is turned on. After an initial voltage is applied to the transmission-side capacitor _C1 to store electric charge, the primary-side drive switch SW1 is turned off. It is assumed that the voltage across the terminals of the load element 6 is sufficiently low at this point. Next, in step S12, when the power transmitting side switch SW2 and the power receiving side switch SW3 are turned on, characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 occurs. Next, in step S13, the absolute value of the terminal voltage _VC2 becomes maximum due to characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3, and when the terminal voltage _VC1 becomes 0 V, the power transmitting side switch SW2 and the power receiving side switch SW3 are turned off.

Next, in step S14, when the load control switch SW4 is turned on, a charging current _ICS is generated and the inter-terminal voltage _VC2 is reduced. Next, in step S15, the load control switch SW4 is turned off when the charging current _ICS becomes zero. At this time, the terminal voltage _VC2 and the terminal voltage of the load element 6 have the same value. If the voltage between the terminals of the load element 6 is sufficiently low at the time of step S11, the voltage between the terminals of the power receiving side capacitor _VC2 at the time of step S15 is 0 V, or sufficiently low to the extent that it can be regarded as 0 V, and the process can return to step S11 without discharging the power receiving side capacitor _C2 .

(Power transmission method when the voltage between the terminals of the load element is not low)
The second wireless power transmission method according to the fourth embodiment when the voltage across the terminals of the load element 6 is not low will be described with reference to the flowchart shown in FIG. 24(b) and the timing chart shown in FIG. However, it is assumed that the equivalent coupling coefficient K is adjusted in the same manner as when the voltage across the terminals of the load element 6 is low.

Steps S21 to S24 are the same as steps S11 to S14. In step S25, the load control switch SW4 is turned off when the charging current _ICS becomes zero. The inter-terminal voltage _VC2 at this time becomes the same value as the charging voltage _VCS . In the next step S26, the power receiving side capacitor is discharged.

In step S26, when the power transmission side switch SW2 and the power reception side switch SW3 are turned on, characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3 occurs again. In step S27, the absolute value of the terminal voltage _VC1 becomes maximum due to characteristic harmonic transmission between the primary side circuit 2 and the secondary side circuit 3, and when the terminal voltage _VC2 becomes 0 V, the power transmitting side switch SW2 and the power receiving side switch SW3 are turned off. Since the inter-terminal voltage VC _-1 at the time of step S27 is the same value as the inter-terminal voltage VC- _{2 at the time of step S25, the inter-terminal voltage VC-1 and} the charging voltage VC- _S are the same value at the time of step S27. Therefore, in this case, the charging voltage VC _S can be monitored with the voltage VC ₁ between the terminals.

As described above, according to the power transmission device according to the fourth embodiment of the present invention, similarly to the power transmission devices according to the first to third embodiments, the DC power supply 5, which has a simple control circuit and peripheral circuits and is inexpensive, can be used. This eliminates the need for an expensive switching power supply, simplifies the circuit configuration, minimizes the power loss on the control circuit side, minimizes breakage, and facilitates circuit design. As a result, according to the power transmission device according to the fourth embodiment, the overall configuration of the power transmission device can be simplified, and the overall power transmission efficiency including the loss of the power supply circuit (zero-order circuit) can be simplified to a value close to 100% in principle, the limit power of power transmission can be pushed up to infinity in principle, and the limit distance of power transmission can be extended to infinity in principle.

(Fifth embodiment)
In the power transmission device according to the fifth embodiment of the present invention, as shown in FIG. 26, the primary side circuit 2C and the secondary side circuit 3C are electrostatically coupled by a first mutual coupling capacitor _C23 and a second mutual coupling capacitor _C24 . According to the law of electromagnetic induction and Maxwell's equations, such replacement of coils and capacitors is possible. That is, as shown in FIG. 26 , the power transmission device according to the fifth embodiment includes a power transmission side capacitor C ₂₁ for accumulating electrostatic energy, a power transmission side coil L ₂₁ connected in parallel to the power transmission side capacitor C 21 _{and sent from the power transmission side capacitor C 21 as} magnetic energy, and having a power transmission side coil L ₂₁ for circulating this magnetic energy to the power transmission side capacitor C 21 .

26, the power transmission apparatus according to the fifth embodiment further includes a first mutual coupling capacitor _C23 having one electrode connected to one node connecting the power transmission side capacitor _C21 and the power transmission side coil _L21 in parallel, and a second mutual coupling capacitor _C24 having one electrode connected to the other node connecting the power transmission side capacitor _C21 and the power transmission side coil _L21 in parallel. It is different from power transmission equipment. In the power transmission device according to the fifth embodiment, one electrode is connected to the other electrode of the first mutual coupling capacitor _C23 , the other electrode is connected to the other electrode of the second mutual coupling capacitor _C24 , the secondary side circuit has a power receiving side capacitor _C22 that receives electrostatic energy from the primary side circuit 2, and a power receiving side coil _L22 that is connected in parallel to the power receiving side capacitor _C22 and stores the electrostatic energy accumulated in the power receiving side capacitor _C22 as magnetic energy. 3.

Furthermore, as shown in FIG. 26, the power transmission device according to the fifth embodiment includes a DC power supply 5 that forms a circuit that connects between one terminal and the other terminal of the power transmission side coil _L21 , and a primary side drive switch SW1 that is connected in series between one terminal of the power transmission side coil _L21 and the DC power supply 5, and that inputs an intermittent DC voltage stepwise to the power transmission side coil _L21 . Also, a circuit connecting between one terminal and the other terminal of the power receiving side coil _L22 is configured, and a load element 6 for receiving magnetic energy from the power receiving side coil _L22 and a load side diode D2 having an anode connected to one terminal of the power receiving side coil _L22 and a cathode connected to the load element 6 are provided. Even with electrostatic coupling as shown in FIG. 26, the power transmission device according to the fifth embodiment can transmit electrical energy from the primary circuit 2 to the secondary circuit 3 without contact. The waveforms of the currents flowing through the power transmitting side coil _L21 and the power receiving side coil _L22 are obtained by simulation based on ordinary alternating current theory, and the characteristic harmonic transmission waveforms between the primary side circuit 2 and the secondary side circuit 3 are confirmed.

The equivalent coupling coefficient of the primary side circuit 21 and the secondary side circuit 22 is 0, the inductance of both the power transmission side coil L ₂₁ and the power reception side coil L ₂₂ is 0.1 μH, the capacitance of both the power transmission side capacitor C ₂₁ and the power reception side capacitor C ₂₂ is 400 pF, and the capacitance of the first mutual coupling capacitor C ₂₃ and the second mutual coupling capacitor C ₂₄ is both 500 pF. The DC power supply 5 is a constant voltage source as in the first embodiment. FIG. 27(a) shows waveforms of currents flowing through the power transmission side coil _L21 and the power reception side coil _L22 . FIG. 27(b) shows waveforms of voltages V21 and _V22 between the terminals of the power transmission side capacitor _C21 and the power reception side capacitor C22, respectively. At 0 ns, the currents flowing through the power transmitting side coil _L21 and the power receiving side coil _L22 are 30 A and 0 A, respectively, and at 60 ns, the currents flowing through the power transmitting side coil _L21 and the power receiving side coil _L22 are 30 A and 0 A, respectively, resulting in characteristic harmonic transmission between the primary circuit 2 and the secondary circuit 3.

As described above, according to the power transmission device according to the fifth embodiment of the present invention, even in the case of electrostatic coupling, as in the case of magnetic coupling in the power transmission devices according to the first to fourth embodiments, an inexpensive DC power supply 5 with simple control circuits and peripheral circuits can be used, so an expensive switching power supply is unnecessary. In addition, even with electrostatic coupling, the circuit configuration is simplified as in the power transmission devices according to the first to fourth embodiments, the circuit design is simplified, and the power loss on the control circuit side is minimized. As a result, according to the power transmission device according to the fifth embodiment of the present invention, the overall configuration of the power transmission device is simplified, making it possible to reduce the weight, size, and improve efficiency. It is possible to inexpensively manufacture a wireless power transmission device in which the overall power transmission efficiency including the loss of the power supply circuit (zero-order circuit) is increased to a value close to 100%, the limit power of power transmission is increased to infinity in principle, and the limit distance of power transmission is extended to infinity in principle.

(Other embodiments)
As described above, the present invention has been described by the first to fifth embodiments, but the statements and drawings forming part of this disclosure should not be understood to limit the present invention. Various alternative embodiments, implementations and operational techniques will become apparent to those skilled in the art from this disclosure. For example, in the power transmission device according to the fifth embodiment of the present invention, the circuit configuration including only one primary side drive switch SW1 has been described as the electrostatic coupling method. It is only an example. As described in the power transmission device according to the second embodiment of the present invention, even in the case of the circuit configuration of the electrostatic coupling method, the configuration including the primary side drive switch SW1 and the power transmission side switch SW2 is possible.

Similarly, even in the case of the circuit configuration of the electrostatic coupling method as described in the power transmission device according to the third embodiment of the present invention, it is possible to adopt a configuration including the primary side drive switch SW1, the power transmission side switch SW2, and the power reception side switch SW3. Further, the configuration may be the same as that of the power transmission device according to the fourth embodiment of the present invention, including the primary side drive switch SW1, the power transmission side switch SW2, the power reception side switch SW3, and the load control switch SW4.

1(a), 10(a), 13(a), 16 (24 and 26) can be combined with each other to configure the power transmission device according to the present invention. In addition, the magnetic coupling degree control mechanism described with reference to FIGS. The invention includes various embodiments not described in the specification and drawings, and the technical scope of the invention is defined only by the matters specifying the invention according to the scope of claims that are valid from the above description.

1 power supply side circuit 2, 2C primary side circuit 3, 3C secondary side circuit 5 DC power supply 6 load element 32 spacer 33 wheel stop 41 ranging unit 411 light emitting unit 412 light receiving unit 42, 47 logic operation control unit 421 distance operation unit 422, 472 coupling coefficient calculation unit 43 coupling coefficient adjustment drive device 45 data storage device 46 transmission efficiency measurement Unit 461 Ammeter 462 Voltmeter 471 Transmission efficiency calculator 71 Substrate 72 Source region 73 Drain region 81 Gate oxide film 82 Source electrode 83 Drain electrode 84 Gate electrode

Claims (8)

a primary side circuit that has a power transmission side capacitor, a power transmission side coil that is connected to the power transmission side capacitor and that accumulates electrostatic energy sent from the power transmission side capacitor as magnetic energy and circulates the magnetic energy to the power transmission side capacitor, and that exhibits a rise and fall transient response in which a knobby sawtooth wave is dulled by the accumulation and the circulation;
a secondary side circuit having a power receiving side coil facing the power transmitting side coil and receiving the magnetic energy from the power transmitting side coil, a power receiving side capacitor connected to the power receiving side coil and storing the magnetic energy accumulated in the power receiving side coil as electrostatic energy, and exhibiting an oscillating waveform having no symmetry with the rising and falling transient responses;
, wherein the time constant of the rising/falling transient response and the time constant inherent in the circuit characteristics of the secondary side circuit are set so as to harmonize, and electric energy is transmitted from the primary side circuit to the secondary side circuit without contact. 2. The power transmission device according to claim 1, further comprising a primary-side drive switch having an output terminal connected to a connection node between one terminal of the power transmission side capacitor and one terminal of the power transmission side coil, and stepwise inputting a DC voltage to the power transmission side capacitor to limit free oscillation of the rising/falling transient response. 3. The power transmission device according to claim 2, further comprising a DC power supply forming a circuit connecting between the input terminal of the primary side drive switch and the other terminal of the power transmission side capacitor. A power transmission method for contactlessly transmitting electrical energy from a primary side circuit including a power transmission side capacitor and a power transmission side coil connected in parallel to the power transmission side capacitor to a secondary side circuit including a power receiving side coil and a power receiving side capacitor connected in parallel to the power reception side coil,
a step of sending the electrostatic energy accumulated in the power transmission side capacitor to the power transmission side coil, storing it as magnetic energy in the power transmission side coil, and allowing the accumulated magnetic energy to flow back to the power transmission side capacitor;
a step of receiving a portion of the magnetic energy accumulated in the power transmitting coil by the power receiving coil and accumulating the received portion of the magnetic energy;
a step in which the magnetic energy accumulated in the power receiving side coil is sent to the power receiving side capacitor to be accumulated as electrostatic energy, and a part of the accumulated electrostatic energy is circulated to the power receiving side coil;
A power transmission method, comprising: 5. The power transmission method according to claim 4, further comprising the step of receiving the electrical energy circulated to the power receiving side coil by the power transmitting side capacitor, and accumulating the received electrical energy together with the electrostatic energy already stored in the power transmitting side capacitor. 5. The power transmission method according to claim 4, wherein the magnetic energy accumulated in the power receiving side coil is sent to the power receiving side capacitor, and combined with the electrostatic energy already accumulated in the power receiving side capacitor, is accumulated as electrostatic energy. further comprising a step of receiving, by the power transmission side capacitor, the electrical energy step-inputted to the power transmission side capacitor, and accumulating the received electrical energy together with the electrostatic energy already stored in the power transmission side capacitor;
At least part of the electrostatic energy accumulated together becomes electrostatic energy accumulated in the power receiving side capacitor,
The power transmission method according to claim 5 or 6, wherein there is a moment when the electrostatic energy accumulated in the power transmission side capacitor, the magnetic energy accumulated in the power transmission side coil, and the magnetic energy accumulated in the power reception side coil become zero or minimal. In the vicinity of the time when at least part of the electrostatic energy accumulated in the power transmission side capacitor becomes the electrostatic energy of the power reception side capacitor,
8. The power transmission method according to claim 7, further comprising supplying power to a load connected as a parallel circuit of said power receiving side capacitor.

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