JP7616607B2 - Power Converter - Google Patents

Description

The present invention relates to a power converter.

Conventionally, RF (Radio Frequency) power amplifiers have been developed that can obtain large power by connecting multiple resonant power sources in parallel. RF power amplifiers are used for various applications such as induction heating, plasma processes, and wireless power supply. In these applications, RF power amplifiers are required to be small and highly efficient in terms of ease of installation and power utilization rate, and high output is required to achieve higher processing capabilities and power supply.

For example, a class E resonant power supply using a semiconductor switch is suitable for making RF power amplifiers smaller and more efficient. In principle, the output power of a class E resonant power supply can be increased by increasing the DC input voltage, but a voltage 3.56 times the input voltage is applied to the semiconductor switch. For this reason, it is limited by the withstand voltage of the semiconductor switch.

By combining the power of multiple class E resonant power supplies using a power combining circuit, it is possible to achieve an output that exceeds the output limit set by the voltage resistance of the semiconductor switch. However, since a separate power combining circuit must be installed, this can lead to a deterioration in power conversion efficiency. There are also research examples in which the output of class E resonant power supplies is directly combined without the need for a separate power combining circuit.

In this research example, to achieve high output power, it is necessary to match the output magnitude and phase of each class E resonant power supply. Even if each class E resonant power supply has the same configuration, it is difficult to match the output magnitude and phase of each resonant power supply because there is variation in the elements within each class E resonant power supply. Output imbalance increases the loss of the semiconductor switching elements within a particular class E resonant power supply, preventing it from achieving high output.

In light of these circumstances, the technology described in Patent Document 1 proposes a configuration in which a phase-controlled matching circuit equipped with a Π-connected variable capacitor or variable inductor is added and a resonant power supply is connected in parallel. In this configuration, the input current, output current, and output voltage to the resonant power supply are detected, and an automatic matching control circuit controls the phase-controlled matching circuit based on the results, thereby matching the magnitude and phase of the output of the resonant power supply.

However, the phase-control matching circuit described in Patent Document 1 uses variable capacitors and variable inductors, which leads to an increase in the size of the circuit and the resonant power supply. In addition, the specific configuration of the automatic matching control circuit and the specific method of controlling the detection value are not described.

In the configuration shown in Figure 3 of Patent Document 2, a phase difference control circuit is added. This is used to control the high frequency pulse drive circuit, and by controlling the phase difference between the switching elements of the parallel-connected resonant power supply, it is possible to match the magnitude and phase of the output of the power supply circuit. Since this configuration controls the phase difference between the switching elements, no large additional parts are required. However, even in the technology described in Patent Document 2, the specific configuration of the phase difference control circuit and the specific method of phase control are not described.

Japanese Patent No. 5832702 (Fig. 1) Japanese Patent No. 6305438 (Fig. 3)

The present invention was made in consideration of the above, and its purpose is to provide a power converter that can match the magnitude and phase of the output of each resonant power source as closely as possible using a feedback circuit that is as simple as possible.

According to the invention described in claim 1, two or more systems of class E resonant power supplies are provided, each of which outputs AC using a switching element and has outputs connected in parallel. The current detection unit detects the current of each resonant power supply. The judgment unit judges whether the current detection value of the resonant power supply by the current detection unit is larger or smaller than the reference current value. The phase adjustment circuit adjusts the switching phase of the switching element of each resonant power supply according to the judgment result of the judgment unit, and controls the input currents to match by leading or lagging the switching phase.

The phase adjustment circuit delays the switching phase of the resonant power supply if the current detection value by the current detection unit is greater than the reference current value, and advances the switching phase of the resonant power supply if the current detection value is smaller, and controls the phase to be fixed when the reference current value and the current detection value match. This allows the circuit to be configured using as simple a feedback circuit as possible, and makes it possible to match the magnitude and phase of the output of each resonant power supply as closely as possible. Here, it is desirable that the reference current value is set to a predetermined value in advance, or is set based on the current detection value of the resonant power supply connected in parallel.

Electrical configuration diagram of a power converter in the first embodiment Electrical diagram of the phase adjustment circuit Simulation result diagram 1 Simulation result diagram 2 Experimental results diagram 1 Experimental results diagram 2 Electrical configuration diagram of a resonant power supply in the second embodiment Electrical configuration diagram of a resonant type power supply in the third embodiment A diagram showing the optimum operating waveform of a class E amplifier. Electrical configuration diagram of a power converter according to a fourth embodiment Electrical configuration example used for PI control in the fifth embodiment Simulation results comparing I control and PI control Electrical configuration diagram when two resonant power supplies are combined in the sixth embodiment. Illustrative diagram of limit conditions for suppressing imbalance through phase control FIG. 13 is an electrical configuration diagram of a multi-stage phase adjustment circuit in the seventh embodiment.

Below, several embodiments of the power converter will be described with reference to the drawings. Note that parts that are substantially common to each embodiment will be described with the same or similar reference numerals, for example, the same reference numerals in the tens and ones digits.

First Embodiment
A first embodiment of a power converter 10 will be described below with reference to Fig. 1 to Fig. 6. The power converter 10 is configured to have two or more systems of resonant class E resonant power supplies 21, 22, ..., 2n. The resonant power supplies 21, 22, ..., 2n are connected in parallel and have their outputs commonly connected. The resonant power supplies 21, 22, ..., 2n are configured to output AC in a class E system using switching elements _M1 , _M2 , ..., _Mn, respectively. In the following description, components of the kth system (where k = 1 ... n) of the resonant power supply 2 will be described with a number corresponding to the suffix k.

Each resonant power supply 21, 22, ..., 2n is divided into a master resonant power supply 21 that outputs a reference current value, and a slave resonant power supply 22, ..., 2n that operates while inputting and correcting the reference current value. The resonant power supply 21 includes a switching element _M1 . The resonant power supply 21 includes a choke inductor _Lf1 , a resonant capacitor _Cr1 , and a resonant inductor _Lr1 that are connected in series, and a drain-source connection of the switching element _M1 and a shunt capacitor _C1 between a common connection point of the choke inductor _Lf1 and the resonant inductor _Lr1 and the ground. A square wave is input to the gate of the switching element _M1 from a square wave oscillator 30 through a driver 111. A current sensor 41 is provided at the input of the resonant power supply 21.

Power is supplied to the input of each of the resonant power supplies 21, 22, ..., 2n from a DC power supply 100. The outputs of each of the resonant power supplies 21, 22, ..., 2n are commonly connected, and a load resistor R _L is connected to the common connection point. Each of the resonant power supplies 21, 22, ..., 2n is provided with current sensors 41, 42, ..., 4n as input current detection units that detect the input current from the DC power supply 100. The current sensor 41 that detects the input current of the resonant power supply 21 uses the input current detection value as a reference current value, and outputs the reference current value to the resonant power supplies 22, ..., 2n.

The resonant power supplies 22, ..., 2n on the slave side each include a switching element _M2 , ..., _Mn . The resonant power supplies 22, ..., 2n are also configured by connecting choke inductors _Lf2 , ..., _Lfn , resonant capacitors _Cr2 , ..., _Crn , and resonant inductors _Lr2 , ..., _Lrn in series, and connecting the drain-source of the switching elements _M2 , ..., _Mn and shunt capacitors _C2 , ..., _Cn between the common connection point of the choke inductors _Lf2 , ..., _Lfn and the resonant inductors _Lr2 , ..., _Lrn and ground. A square wave is input to the gates of the switching elements _M2 , ..., _Mn from the square wave oscillator 30 through phase adjustment circuits 62, ..., 6n and drivers 112, ..., 11n.

The resonant power supplies 22, ..., 2n on the slave side are each equipped with a current sensor 42, ..., 4n at their input. The resonant power supplies 22, ..., 2n are each equipped with an error amplifier 52, ..., 5n and a phase adjustment circuit 62, ..., 6n. The error amplifiers 52, ..., 5n are circuits that function as a judgment unit that, when a reference current value is input, judges whether the input current detection value by the current sensor 42, ..., 4n of the resonant power supply 22, ..., 2n itself is larger than the reference current value. The error amplifiers 52, ..., 5n output the judgment result to the phase adjustment circuits 62, ..., 6n.

The internal components of the resonant power supplies 22, ..., 2n on the slave side are all the same. Therefore, the internal configuration of the resonant power supply 22 will be described, and the description of the internal components of the other resonant power supplies 23, ..., 2n on the slave side will be omitted. The resonant power supply 22 is configured with an error amplifier 52. As illustrated in FIG. 2, the error amplifier 52 is configured with an integrator including an operational amplifier OP having resistors _Rm and _Rp connected to an inverting input terminal and a non-inverting input terminal, and a capacitor _Cf connected between the output of the operational amplifier OP and the inverting input terminal. The error amplifier 52 inputs the input current detection value by the current sensor 42 of its own resonant power supply 22 to the inverting input terminal through the resistor _Rm , and inputs a reference current value to the non-inverting input terminal through the resistor _Rp , and amplifies the difference between them and outputs it to the phase adjustment circuit 62.

The current sensor 42 is constructed using a magnetic field detection type current sensor. For example, the current sensor 42 is constructed by combining a magnetic core with a slit in one part, which is arranged to surround the current path, and a Hall element arranged in the slit, and detects the current by detecting the magnetic field using the Hall element. By using this current sensor 42, it is possible to prevent power loss. The current sensor 42 may also be constructed by installing a Hall IC in the current path flowing through the board. By using such a current sensor 42, the implementation is simplified and it becomes easier to handle.

The phase adjustment circuit 62 of the resonant power supply 22 adjusts the phase of the drive voltage for driving the switching element _M2 (hereinafter referred to as the switching phase) in accordance with the determination result by the error amplifier 52. If the input current detection value is greater than the reference current value, the phase adjustment circuit 62 delays the switching phase for driving the switching element _M2 of the resonant power supply 22. If the input current detection value is smaller than the reference current value, the phase adjustment circuit 62 advances the switching phase for driving the switching element _M2 of the resonant power supply 22. The phase adjustment circuit 62 controls so as to fix the switching phase when the reference current value and the input current detection value match.

2, the phase adjustment circuit 62 is configured using an integrator 72, a resistive voltage divider circuit 82 configured by connecting resistors Ra and Rb in series, a D flip-flop 92, and other peripheral circuits. The integrator 72 receives a square wave output from a square wave oscillator 30, delays the phase of the voltage according to a predetermined time constant to dull the voltage waveform, and inputs the resulting voltage to the clock terminal of the D flip-flop 92 through a bypass capacitor _Cb . The D flip-flop 92 is configured by pulling up the D terminal and connecting the resistive voltage divider circuit 82 to the clock terminal. A peripheral circuit 102 is connected to the /Q terminal and /CLR terminal of the D flip-flop 92.

The output of the error amplifier 52 is applied to the input of the resistive voltage divider circuit 82. In other words, the voltage according to the output of the error amplifier 52 is superimposed on the output voltage of the integrator 72 and input to the clock terminal. As a result, a voltage obtained by adding or subtracting a voltage that changes according to the output voltage of the error amplifier 52 is input to the clock terminal of the D flip-flop 92.

The clock terminal of the D flip-flop 92 accepts a rising pulse when the voltage exceeds a predetermined voltage level and accepts a falling pulse when the voltage falls below the predetermined voltage level. Therefore, a square wave whose phase has been shifted by an amount corresponding to the output voltage of the error amplifier 52 is input to the clock terminal of the D flip-flop 92. The D flip-flop 92 outputs a square wave voltage from the Q terminal in synchronization with the input voltage of the clock terminal, and can output a shaped voltage whose phase has been shifted according to the output of the error amplifier 52. As a result, the phase adjustment circuit 62 outputs a phase-adjusted result based on the integration result by the integrator 72.

This shaping voltage is input to the gate terminal of the switching element _M2 . Therefore, it is possible to change the phase of the square wave applied to the gate terminal of the switching element _M2 of the resonant power supply 22 on the slave side relative to the phase of the square wave applied to the gate terminal of the switching element _M1 that drives the resonant power supply 21 on the master side.

<Simulation results>
The inventors performed a simulation by giving a normal distribution variation equivalent to 3σ, where σ is the standard deviation of the element value variation of the resonant inductors _Lr1 , ..., _Lrn used when generating an AC signal by each of the resonant power sources 21, ..., 2n. The number n of parallel connections of the resonant power sources 22, ..., 2n was set to 4, and the number of trials was set to 300. The results are shown in Figs. 3 to 6.

3 and 4 show the drain loss difference on the vertical axis and the worst drain loss on the horizontal axis, both plotted on the same scale. In the case of no phase adjustment, when the element values vary, as shown in Fig. 3, it was confirmed that the drain loss difference of the switching elements _M1 , ..., _Mn and the worst drain loss value also vary. By using the phase adjustment circuits 62, ..., 6n of this embodiment, it was confirmed that the drain loss, i.e., the switching element loss difference, can be reduced, as shown in Fig. 4, and the worst loss value of the switching elements _M1 , ..., _Mn can also be reduced.

Fig. 5 shows the experimental results of the drain loss of the switching elements _M1 , ..., _Mn versus the output power when there is no phase adjustment, and Fig. 6 shows the experimental results of the drain loss of the switching elements _M1 , ..., _Mn versus the output power when there is phase adjustment. Figs. 5 and 6 compare and illustrate the drain loss on the vertical axis and the output power on the horizontal axis on the same scale. As shown in Fig. 6, the experimental results with phase adjustment confirmed that the increase in loss of the switching elements _M1 , ..., _Mn can be suppressed even when the output power is increased. Furthermore, the output of each resonant power source 21, ..., 2n can be leveled, and the deterioration associated with the loss of each resonant power source 21, ..., 2n can be leveled.

The effects of the present embodiment will be described below. If the input current detection value is greater than the reference current value, the phase adjustment circuits 62, ..., 6n delay the switching phase of the resonant power supplies 22, ..., 2n that become slaves. If the input current detection value is smaller than the reference current value, the phase adjustment circuits 62, ..., 6n advance the switching phase of the resonant power supplies 22, ..., 2n. The phase adjustment circuits 62, ..., 6n control the phase to be fixed when the reference current value and the input current detection value match. Even if the inductance of the resonant inductors _Lr1 , ..., _Lrn varies greatly, the magnitude and phase of the output of each resonant power supply 21, ..., 2n can be matched as much as possible by using a feedback circuit as simple as possible.

The phase adjustment circuits 62, ..., 6n can be constructed using small components, and the magnitude and phase of the outputs of the resonant power supplies 21, ..., 2n can be matched as much as possible. The error amplifiers 52, ..., 5n do not require rectification because their detection signals are direct current, making processing easy. This allows the error amplifiers 52, ..., 5n and phase adjustment circuits 62, ..., 6n to be constructed simply. This can be achieved by simple control in which the switching phase is feedback-controlled according to the magnitude of the input current detection value of the current sensors 41, ..., 4n and the reference current value. Moreover, there is no need to limit the composite number of the resonant power supplies 21, ..., 2n.

In addition, the effect can be achieved simply by adding small components such as current sensors 41, ..., 4n, error amplifiers 52, ..., 5n, and phase adjustment circuits 62, ..., 6n. Furthermore, even if there is variation in the passive elements in the individual resonant power supplies 21, ..., 2n, the power can be combined with the output magnitude and phase of each resonant power supply 21, ..., 2n being matched, and an increase in loss in the switching elements _M1 , ..., _Mn can be suppressed.

Second Embodiment
The second embodiment will be described with reference to Fig. 7. A push-pull configuration may be used for each of the resonant power supplies 21, ..., 2n. Fig. 7 shows the electrical configuration of one system of the resonant power supply 221, but the configuration of this resonant power supply 221 may be applied to the configurations of all the resonant power supplies 21, ..., 2n, and these configurations may be connected in parallel.

The resonant power supply 221 is configured by connecting in parallel a series circuit of choke inductors _{Lf1a, Lf1b, resonant inductors Lr1a, Lr1b, and resonant capacitors Cr1a, Cr1b. Switching elements M1a , M1b and capacitors C1a, C1b are connected} between the intermediate nodes of the choke inductors _Lf1a , _Lf1b , and the resonant inductors _{Lr1a , Lr1b} of each series-connected circuit and the ground. At this time, a square wave oscillator 30 applies square wave pulses in complementary opposite phases to the gates of the switching elements M1a _{, M1b} through the drivers 111a, 111b, thereby forming a push-pull configuration. By adopting such a push-pull configuration, it is possible to reduce the imbalance _that is likely to occur in the resonant power supply 221.

Third Embodiment
The third embodiment will be described with reference to Fig. 8. As in the second embodiment, the resonant power supplies 21, ..., 2n may be configured in parallel as shown in Fig. 8. As in the second embodiment, Fig. 8 illustrates the configuration of one resonant power supply 321, but the configuration of this resonant power supply 321 may be applied to the configurations of all of the resonant power supplies 21, ..., 2n.

As shown in Fig. 8, a resonant power supply 321 may be configured by connecting choke inductors _Lf1a and _Lf1b , switching elements _M1a and _M1b , and capacitors _C1a and _C1b , and directly connecting a common resonant inductor _Lr1 and a resonant capacitor _Cr1 to this circuit. In this case, it is preferable that the square wave oscillator 30 applies the same pulse of a square wave to the gates of the pair of switching elements _M1a and _M1b through the drivers 111a and 111b. By adopting such a parallel configuration, the imbalance can be reduced.

Fourth Embodiment
The fourth embodiment will be described with reference to Fig. 9 and Fig. 10. For example, since the input current Idc input to the resonant power supply 21 of the first embodiment is passed through the switching element _M1 , there is a certain relationship between the input current Idc and the current is flowing through the switching element _M1 .

For example, in a power converter operating in class E, when the on-duty ratio of switching elements _M1 ... _Mn is D as shown in Figure 9, the current is of switching element _M1 becomes Idc- _I0 sin(ωt+φ) when 0<ωt≦ _πD , and becomes 0 when πD≦ωt≦2πD.

For this reason, as shown in the power converter 410 of Fig. 10, the drain and gate terminals of the current sense elements _Ms1 ... _Msn may be connected to the drain and gate of the switching elements _M1 ... _Mn , respectively, and the sources of the current sense elements _Ms1 ... _Msn may be connected to the sources of the switching elements _M1 ... _Mn via detection resistors, so that the current is flowing through each switching element _M1 ... _Mn may be obtained by the detection resistor. Then, the detection values of the current sense elements _Ms1 ... _Msn and the detection resistor may be input to the error amplifiers 52 ... 5n, so that the imbalance suppression control may be performed based on the detection values of the current sense elements _Ms1 ... _Msn without using the current sensors 42 ... 4n. This allows the current sensors 42 ... 4n to be reduced, resulting in a smaller size.

Fifth Embodiment
The fifth embodiment will be described with reference to Figs. 11 and 12. As shown in Fig. 11, an error amplifier 552 replacing the error amplifier 52 is provided with a feedback resistor _Rf connected in series to a capacitor _Cf. This allows PI control to be performed. Fig. 12 shows a comparison of simulation results of transient phenomenon characteristics plotting output voltage versus time when I control is performed and when PI control is performed. It has been confirmed that the time required to control the output voltage to a constant value within a certain range can be improved by a time difference Δt by performing PI control compared to when I control is performed.

Sixth Embodiment
The sixth embodiment will be described with reference to Fig. 13 and Fig. 14. In the above-mentioned embodiments, a configuration in which n resonant power sources 21, ..., 2n are combined is described, but in this embodiment, a configuration in which two systems of resonant power sources 21, 22 are combined, as shown in a power converter 610 in Fig. 13, is described.

Figure 14 illustrates the results of a simulation of the drain losses that occur in the switching elements M1 and M2 of the two resonant power supplies 21 and 22 when, for example, the inductance of the resonant inductor _Lr1 of one resonant power supply ₂₁ is changed and the element values of the other resonant power supply 22 (for example, the inductance of the resonant inductor _Lr2 and the resonant capacitor _{Cr2 )} are fixed.

14, it can be seen that when the drain loss characteristics of both resonant power supplies 21, 22 deviate from a predetermined range from the design value of the inductance, the drain loss rises sharply and the drain loss in the resonant power supplies 21, 22 becomes easily unbalanced. In other words, this shows that if the drain-source voltage amplitudes of the switching elements _M1 , _M2 of the two resonant power supplies 21, 22 are easily matched, the unbalance can be dramatically suppressed, and that when the unbalance in the voltage amplitudes increases, the drain loss rises sharply, causing the unbalance.

The theoretical boundary condition for the limit of imbalance that can be suppressed by phase control can be calculated as sin(θ2-θ1)=( _Xr2 - _Xr1 )/(2·R _L ). Here, _Xr1 = _ωLr1-1 / _ωCr1 , _Xr2 = _ωLr2-1 / _ωCr2 , and θ1 and θ2 indicate the phase difference from the reference value of the square wave voltage applied to the switching elements _M1 and _M2 . If this condition is changed into a formula for the condition between element values, it becomes 2R _L > | _Xr2 - _Xr1 |. By satisfying this condition, imbalance can be suppressed.

Seventh Embodiment
The seventh embodiment will be described with reference to Fig. 15. As shown in Fig. 15, a phase adjustment circuit 62 that receives the output of an error amplifier 52 and adjusts the phase may be configured in a 2-stage cascade configuration, for example. The cascade configuration is not limited to a 2-stage configuration, and may be configured in a 3-stage or more cascade configuration. By configuring in multiple stages, it is possible to increase the adjustment range of the circuit.

Other Embodiments
The present invention is not limited to the above-described embodiment, and can be implemented in various modifications and variations without departing from the spirit and scope of the present invention. For example, the following modifications or extensions are possible.

In the above embodiment, the reference current value is set based on the input current detection value of one of the resonant power supplies 21 connected in parallel, but this is not limited to this. The reference current value may be set to a predetermined value in advance, or the reference current value may be set to the average value of the input current detection values of all the resonant power supplies 21, 22, ..., 2n connected in parallel. It is desirable that the reference current value be set to a predetermined value in advance, or based on the current detection values of the resonant power supplies connected in parallel. The type of flip-flop is not limited to the D flip-flop.
Although the resonant power supplies 21, 22, . . . , 2n are configured to operate in class E, a current source type resonant power supply such as class E, class Φ, class F, etc. may also be used.

Although the present invention has been described in accordance with the above-described embodiment, it is understood that the present invention is not limited to the embodiment or structure. The present invention also includes various modifications and modifications within the scope of equivalents. In addition, various combinations and forms, as well as other combinations and forms that include one, more, or less than one element, are also within the scope and concept of the present invention.

In the drawing, 10, 410, 610 indicate power converters, 21, ..., 2n indicate resonant power supplies, 41, ..., 4n indicate current sensors (current detection units, input current detection units), 52, ..., 5n indicate error amplifiers (determination units), 62, ..., 6n indicate phase adjustment circuits, 72 indicates an integrator, 82 indicates a resistive voltage divider circuit, and 92 indicates a D flip-flop (flip-flop).

Claims (12)

Two or more systems of resonant power supplies (21, ..., 2n) are provided, each of which outputs AC by using a switching element ( _M1 , ..., _Mn ) and is connected in parallel;
A current detection unit (41, ..., 4n) for detecting a current flowing through each resonant power supply;
a determination unit (52, . . . , 5n) for determining whether a current detection value detected by the current detection unit is larger than a reference current value;
a phase adjustment circuit (62, ..., 6n) that adjusts a switching phase of the switching element of each resonant power supply in accordance with a determination result of the determination unit, and controls the switching phase to be advanced or delayed so as to match the current detection value with the reference current value,
the phase adjustment circuit is configured to control so that if the current detection value by the current detection unit is greater than the reference current value, the phase adjustment circuit delays the switching phase of the resonant power supply, and conversely, if the current detection value is smaller than the reference current value, the phase adjustment circuit advances the switching phase of the resonant power supply, and fixes the phase when the reference current value and the current detection value match;
A power converter in which the reference current value is determined in advance to a predetermined value, or is determined based on the current detection value of the resonant power supply connected in parallel. The power converter according to claim 1, wherein the reference current value is set based on the current detection value of one of the predetermined resonant power sources (21) connected in parallel. The power converter according to claim 1, wherein the reference current value is set to a predetermined value. The power converter according to claim 1, wherein the reference current value is set to the average value of the current detection values of all the resonant power sources (21, ..., 2n) connected in parallel. The resonant type power supply is provided for two systems, each of which includes a circuit in which resonant capacitors (C _r1 , C _r2 ) and resonant inductors (L _r1 , L _r2 ) are connected in series, and is configured to combine the outputs of the two systems and output the combined output to a load resistor (R _L );
The power converter according to any one of claims 1 to 4, wherein the variations associated with the resonant capacitor and the resonant inductor constituting the resonant power supply and the load resistance are configured to satisfy the relationship 2R _L > |X _r2 -X _r1 |, where X _r1 = ωL _r1 - 1/ωC _r1 and X _r2 = ωL _r2 - 1/ωC _r2 . The power converter according to any one of claims 1 to 5, wherein the current detection unit detects the input current input to each resonant power supply as a direct current. The power converter according to any one of claims 1 to 5, wherein the current detection unit detects a current passing through the switching element configured in the resonant power supply. The resonant power supplies are each provided with the current detection unit,
an integrator (72) that integrates the current detection value by the current detection unit;
The power converter according to claim 1 , wherein the phase adjustment circuit adjusts the phase based on a result of integration by the integrator. The present invention is configured using an error amplifier (52, ..., 5n) that performs PI control based on a difference between the current detection value and the reference current value,
9. The power converter according to claim 8, wherein the phase adjustment circuit performs phase adjustment using the error amplifier. the determination unit is configured using an error amplifier (52, ..., 5n) that outputs a difference between the current detection value and the reference current value,
10. The power converter according to claim 1, wherein the phase adjustment circuit comprises: an integrator (72) that receives a rectangular wave voltage of a rectangular wave oscillator and delays the phase of the voltage according to a predetermined time constant to dull the voltage waveform; a resistive voltage divider circuit (82) that receives an output of the error amplifier; and a flip-flop (92) that superimposes and inputs an output of the resistive voltage divider circuit and an output of the integrator to a clock terminal, and the flip-flop outputs the rectangular wave voltage in synchronization with an input voltage to the clock terminal. The power converter according to any one of claims 1 to 10, wherein the phase adjustment circuit has a multi-stage configuration. 12. The power converter according to any one of claims 1 to 11, wherein the resonant power supply is a class E resonant power supply.

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