JP7658286B2 - Motor control device - Google Patents

Description

The present invention relates to a control device that controls the operation of an electric motor.

There is a motor control device that uses high-frequency current contained in the current flowing through the motor to estimate the rotor position of the motor with 180 degrees of ambiguity, and then determines the polarity of the rotor based on the magnitude relationship of the current that changes when a positive pulse voltage and a negative pulse voltage are alternately superimposed on the voltage command value, and then determines the rotor position using the determined polarity. Related technology is described in Patent Document 1.

In this way, the above control device is configured to tentatively estimate the rotor position, determine the rotor polarity, and then use the determined polarity to determine the rotor position, so there is a concern that it may take a relatively long time to estimate the rotor position.

JP 2002-171798 A

The object of one aspect of the present invention is to shorten the time it takes to estimate the rotor position in a motor control device that estimates the rotor position of a motor using high-frequency current contained in the current flowing through the motor.

The motor control device according to one embodiment of the present invention includes an inverter circuit that drives the motor based on the comparison result between the voltage value of the carrier wave and a voltage command value, an estimation unit that determines the polarity of the rotor of the motor and estimates the position of the rotor using the polarity, a current conversion unit that converts the current flowing through the motor into a d-axis current and a q-axis current using the position, a current command value output unit that outputs a d-axis current command value and a q-axis current command value based on the difference in rotation speed between the rotor's rotation speed and the rotation speed command value, a voltage command value calculation unit that calculates a d-axis voltage command value so that the difference between the d-axis current and the d-axis current command value is small, and calculates a q-axis voltage command value so that the difference between the q-axis current and the q-axis current command value is small, a superposition unit that superimposes a superposition voltage having a higher frequency than the d-axis voltage command value on the d-axis voltage command value, and a voltage command value conversion unit that converts the d-axis voltage command value output from the superposition unit and the q-axis voltage command value output from the voltage command value calculation unit into the voltage command value using the position.

The estimation unit uses the high-frequency current contained in the d-axis current to determine an evaluation value including the change due to magnetic saturation in the d-axis inductance of the motor, and determines the polarity of the rotor according to the sign of the evaluation value.

For example, when the evaluation value is negative, that is, when the d-axis inductance is relatively small, it can be determined that magnetic saturation occurs in the motor, so that the direction of the magnetic pole generated by the d-axis current and the direction of the magnetic pole generated by the permanent magnet are the same direction, and it can be determined that no error occurs between the estimated position and the actual position. Also, when the evaluation value is positive, that is, when the d-axis inductance is relatively large, it can be determined that magnetic saturation does not occur in the motor, so that the direction of the magnetic pole generated by the d-axis current and the direction of the magnetic pole generated by the permanent magnet are different directions, and it can be determined that an error of 180 degrees occurs between the estimated position and the actual position. In this way, the high-frequency current contained in the d-axis current is used to obtain an evaluation value including the change due to magnetic saturation of the d-axis inductance of the motor, and the presence or absence of an error between the estimated position and the actual position, that is, the polarity of the rotor, can be determined according to the sign of the evaluation value.

In addition, because the rotor position is estimated while the rotor polarity is being determined, the time required to estimate the rotor position can be reduced compared to a configuration in which the rotor position is tentatively estimated, the rotor polarity is determined, and the determined polarity is used to determine the rotor position.

Furthermore, when a high-frequency current included in the d-axis current is _iγh , the amplitude of the superimposed voltage is _Vh , the angular frequency of the superimposed voltage is _ωh , the d-axis inductance of the motor is _Ld , the q-axis inductance of the motor is _Lq , _Ld0 = ( _Ld + _Lq ) / 2, _Ld1 = ( _Ld - _Lq ) / 2, the d-axis current is _id , _id = _{ih sinωht} , and the high-frequency current is _iγh , the estimation unit multiplies the high-frequency current by _sinωht which is a frequency component of the superimposed voltage as a demodulation signal to obtain Equation 1 below, removes (1- _cos2ωht ) which is an AC component from Equation 1 below using a band-elimination filter to obtain Equation 2 below, takes the reciprocal of Equation 2 below to obtain Equation 3 below, and then adds the _sinωh The following formula 4 may be obtained by multiplying by t, and the following formula 5 may be obtained as the evaluation value by extracting a DC component from the following formula 4 using a moving average filter.

The estimation unit may also be configured to include a medium-high speed range position estimation unit that estimates the position of the rotor using an extended induced voltage excited by superimposing the superimposed voltage on the d-axis voltage command value, a low-speed range position estimation unit that estimates the position of the rotor using a high-frequency current included in the d-axis current, a polarity determination unit that sets the position error between the position estimated by the medium-high speed range position estimation unit and the low-speed range position estimation unit and the actual position to 0 degrees when the evaluation value is negative, and sets the position error to 180 degrees when the evaluation value is positive, and a position estimation unit that sets the result of adding the position error determined by the polarity determination unit to the weighted average of the position estimated by the medium-high speed range position estimation unit and the position estimated by the low-speed range position estimation unit to the position used in the current conversion unit and the voltage command value conversion unit.

The position estimation unit may be configured to increase the weighting corresponding to the position estimated by the low-speed region position estimation unit as the rotation speed of the rotor decreases, and the superposition unit may be configured to increase the amplitude of the superposition voltage as the rotation speed of the rotor decreases.

As a result, the rotor rotation speed becomes relatively small, so that the extended induced voltage used in the medium-high speed range position estimation unit becomes relatively small. Even if the accuracy of the position estimated by the medium-high speed range position estimation unit decreases, the accuracy of the position estimated by the low speed range position estimation unit can be improved, so that the accuracy of the position estimated by the position estimation unit can be prevented from decreasing.

According to the present invention, in a motor control device that estimates the rotor position of a motor using high-frequency current contained in the current flowing through the motor, it is possible to shorten the time it takes to estimate the rotor position.

FIG. 2 is a diagram illustrating an example of a control device for an electric motor in the first embodiment. FIG. 4 is a diagram illustrating an example of an estimation unit in the first embodiment. FIG. 11 is a diagram illustrating an example of a control device for an electric motor according to a second embodiment. FIG. 11 illustrates an example of an estimation unit in the second embodiment. 10A and 10B are diagrams illustrating another example of the overlapping portion in the first embodiment.

The details of the embodiment are explained below with reference to the drawings.

First Embodiment
FIG. 1 is a diagram illustrating an example of a control device for an electric motor according to a first embodiment.

The control device 1 shown in FIG. 1 controls the operation of an electric motor M mounted on a vehicle such as an electric forklift or a plug-in hybrid vehicle, and includes an inverter circuit 2 and a control circuit 3. The electric motor M is, for example, an embedded magnet motor or a surface magnet motor.

The inverter circuit 2 drives the electric motor M with power supplied from the power source P, and includes a capacitor C, switching elements SW1 to SW6 (e.g., IGBTs (Insulated Gate Bipolar Transistors)), and current sensors Se1 and Se2. That is, one terminal of the capacitor C is connected to the positive terminal of the power source P and the collector terminals of the switching elements SW1, SW3, and SW5, and the other terminal of the capacitor C is connected to the negative terminal of the power source P and the emitter terminals of the switching elements SW2, SW4, and SW6. The connection point between the emitter terminal of the switching element SW1 and the collector terminal of the switching element SW2 is connected to the U-phase input terminal of the electric motor M via the current sensor Se1. The connection point between the emitter terminal of the switching element SW3 and the collector terminal of the switching element SW4 is connected to the V-phase input terminal of the electric motor M via the current sensor Se2. The connection point between the emitter terminal of switching element SW5 and the collector terminal of switching element SW6 is connected to the W-phase input terminal of motor M.

Capacitor C smoothes the voltage output from power supply P and input to inverter circuit 2.

The switching element SW1 is turned on or off based on the drive signal S1 output from the control circuit 3. The switching element SW2 is turned on or off based on the drive signal S2 output from the control circuit 3. The switching element SW3 is turned on or off based on the drive signal S3 output from the control circuit 3. The switching element SW4 is turned on or off based on the drive signal S4 output from the control circuit 3. The switching element SW5 is turned on or off based on the drive signal S5 output from the control circuit 3. The switching element SW6 is turned on or off based on the drive signal S6 output from the control circuit 3. By turning on or off the switching elements SW1 to SW6, respectively, the DC voltage output from the power source P is converted into three AC voltages whose phases differ from each other by 120 degrees, and these AC voltages are applied to the input terminals of the U phase, V phase, and W phase of the electric motor M, causing the rotor of the electric motor M to rotate.

Current sensor Se1 is composed of a Hall element, a shunt resistor, etc., and detects U-phase current Iu flowing through the U-phase of motor M and outputs it to control circuit 3. Current sensor Se2 is composed of a Hall element, a shunt resistor, etc., and detects V-phase current Iv flowing through the V-phase of motor M and outputs it to control circuit 3.

The control circuit 3 includes a memory unit 4, a drive circuit 5, and a calculation unit 6.

The memory unit 4 is composed of RAM (Random Access Memory) or ROM (Read Only Memory), etc.

The drive circuit 5 is composed of an IC (Integrated Circuit) and compares the voltage value of the carrier wave (triangular wave, sawtooth wave, inverse sawtooth wave, etc.) with the U-phase voltage command value Vu*, V-phase voltage command value Vv*, and W-phase voltage command value Vw* output from the calculation unit 6, and outputs drive signals S1 to S6 according to the comparison results to the gate terminals of the switching elements SW1 to SW6.

For example, when the U-phase voltage command value Vu* is equal to or greater than the voltage value of the carrier wave, the drive circuit 5 outputs a high-level drive signal S1 and a low-level drive signal S2. When the U-phase voltage command value Vu* is smaller than the voltage value of the carrier wave, the drive circuit 5 outputs a low-level drive signal S1 and a high-level drive signal S2. When the V-phase voltage command value Vv* is equal to or greater than the voltage value of the carrier wave, the drive circuit 5 outputs a high-level drive signal S3 and a low-level drive signal S4. When the V-phase voltage command value Vv* is smaller than the voltage value of the carrier wave, the drive circuit 5 outputs a low-level drive signal S3 and a high-level drive signal S4. When the W-phase voltage command value Vw* is equal to or greater than the voltage value of the carrier wave, the drive circuit 5 outputs a high-level drive signal S5 and a low-level drive signal S6. When the W-phase voltage command value Vw* is smaller than the voltage value of the carrier wave, the drive circuit 5 outputs a low-level drive signal S5 and a high-level drive signal S6.

The calculation unit 6 is configured by a microcomputer or the like, and includes a current conversion unit 7, an estimation unit 8, a subtraction unit 9, a torque command value calculation unit 10, a current command value output unit 11, a subtraction unit 12, a subtraction unit 13, a voltage command value calculation unit 14, a superposition unit 15, and a voltage command value conversion unit 16. For example, the current conversion unit 7, the estimation unit 8, the subtraction unit 9, the torque command value calculation unit 10, the current command value output unit 11, the subtraction unit 12, the subtraction unit 13, the voltage command value calculation unit 14, the superposition unit 15, and the voltage command value conversion unit 16 are configured by the microcomputer executing a program stored in the storage unit 4.

The current conversion unit 7 uses the U-phase current Iu detected by the current sensor Se1 and the V-phase current Iv detected by the current sensor Se2 to determine the W-phase current Iw flowing through the W-phase of the motor M.

In addition, the current conversion unit 7 converts the U-phase current Iu, the V-phase current Iv, and the W-phase current Iw into a d-axis current Id (a current component for generating a weakened field in the motor M) and a q-axis current Iq (a current component for generating torque in the motor M) using the position θ^ estimated by the estimation unit 8.

For example, the current converter 7 converts the U-phase current Iu, the V-phase current Iv, and the W-phase current Iw into the d-axis current Id and the q-axis current Iq using the conversion matrix C1 shown in the following equation 6.

The currents detected by the current sensors Se1 and Se2 are not limited to the combination of U-phase current Iu and V-phase current Iv, but may be the combination of V-phase current Iv and W-phase current Iw, or the combination of U-phase current Iu and W-phase current Iw. When the current sensors Se1 and Se2 detect the V-phase current Iv and W-phase current Iw, the current conversion unit 7 uses the V-phase current Iv and W-phase current Iw to determine the U-phase current Iu. When the current sensors Se1 and Se2 detect the U-phase current Iu and W-phase current Iw, the current conversion unit 7 uses the U-phase current Iu and W-phase current Iw to determine the V-phase current Iv.

If the inverter circuit 2 further includes a current sensor Se3 that detects the W-phase current Iw flowing through the W-phase of the motor M in addition to the current sensors Se1 and Se2, the current conversion unit 7 may be configured to convert the U-phase current Iu, V-phase current Iv, and W-phase current Iw detected by the current sensors Se1 to Se3 into the d-axis current Id and the q-axis current Iq using the position θ^ estimated by the estimation unit 8.

The estimation unit 8 uses the d-axis current Id and q-axis current Iq output from the current conversion unit 7, the d-axis voltage command value Vd*' output from the superposition unit 15, and the q-axis voltage command value Vq* output from the voltage command value calculation unit 14 to determine the polarity of the rotor of the electric motor M (the error between the estimated position and the actual position), and estimates the rotor position θ^ using the polarity.

In addition, the estimation unit 8 estimates the rotation speed ω^ by dividing the position θ^ by a fixed time (such as the clock period of the calculation unit 6).

The subtraction unit 9 calculates the rotation speed difference Δω between the rotation speed command value ω* input from the outside and the rotation speed ω^ estimated by the estimation unit 8.

The torque command value calculation unit 10 calculates the torque command value T* using the rotation speed difference Δω output from the subtraction unit 9. For example, the torque command value calculation unit 10 refers to information (not shown) stored in the memory unit 4 that associates the rotation speed (angular frequency) of the rotor of the electric motor M with the torque of the electric motor M, and determines the torque corresponding to the rotation speed equivalent to the rotation speed difference Δω as the torque command value T*.

The current command value output unit 11 uses the torque command value T* to output the d-axis current command value Id* and the q-axis current command value Iq*. For example, the current command value output unit 11 refers to information (not shown) stored in the memory unit 4 in which the torque of the motor M is associated with the d-axis current command value Id* and the q-axis current command value Iq*, and determines the d-axis current command value Id* and the q-axis current command value Iq* that correspond to the torque command value T*.

That is, the current command value output unit 11 outputs the d-axis current command value Id* and the q-axis current command value Iq* based on the rotation speed difference Δω between the rotor rotation speed ω^ and the rotation speed command value ω*.

The subtraction unit 12 calculates the difference ΔId between the d-axis current command value Id* output from the current command value output unit 11 and the d-axis current Id output from the current conversion unit 7.

The subtraction unit 13 calculates the difference ΔIq between the q-axis current command value Iq* output from the current command value output unit 11 and the q-axis current Iq output from the current conversion unit 7.

The voltage command value calculation unit 14 calculates the d-axis voltage command value Vd* and the q-axis voltage command value Vq* by PI control using the difference ΔId output from the subtraction unit 12 and the difference ΔIq output from the subtraction unit 13. For example, the voltage command value calculation unit 14 calculates the d-axis voltage command value Vd* by calculating the following formula 7, and calculates the q-axis voltage command value Vq* by calculating the following formula 8. Note that Kp is a constant for the proportional term of the PI control, Ki is a constant for the integral term of the PI control, Lq is the q-axis inductance of the motor M, Ld is the d-axis inductance of the motor M, ω^ is the rotation speed estimated by the estimation unit 8, and Ke is an induced voltage constant.

d-axis voltage command value Vd* = Kp x difference ΔId + ∫ (Ki x difference ΔId) - ω^LqIq ... Equation 7

q-axis voltage command value Vq* = Kp x difference ΔIq + ∫ (Ki x difference ΔIq) + ω^LdId + ω^Ke ... Equation 8

That is, the voltage command value calculation unit 14 calculates the d-axis voltage command value Vd* so that the difference ΔId between the d-axis current Id and the d-axis current command value Id* is small, and calculates the q-axis voltage command value Vq* so that the difference ΔIq between the q-axis current Iq and the q-axis current command value Iq* is small.

The superimposing unit 15 superimposes a superimposed voltage _Vs , which has a higher frequency than the d-axis voltage command value Vd* output from the voltage command value calculating unit 14, on the d-axis voltage command value Vd* and outputs it as a d-axis voltage command value Vd*'. Note that the amplitude of the superimposed voltage _Vs is _Vh , and the frequency component of the superimposed voltage _Vs is sin _ωht .

For example, as shown in FIG. 1, the superimposing unit 15 adds the d-axis voltage command value Vd* and the superimposed voltage Vs by the adder 151, and sets the result of the addition as the d-axis voltage command value Vd*'.

The voltage command value converter 16 converts the d-axis voltage command value Vd* and the q-axis voltage command value Vq* into a U-phase voltage command value Vu*, a V-phase voltage command value Vv*, and a W-phase voltage command value Vw* using the position θ^ estimated by the estimator 8. For example, the voltage command value converter 16 converts the d-axis voltage command value Vd* and the q-axis voltage command value Vq* into a U-phase voltage command value Vu*, a V-phase voltage command value Vv*, and a W-phase voltage command value Vw* using a conversion matrix C2 shown in the following formula 9.

Figure 2 shows an example of the estimation unit 8 in the first embodiment.

The estimation unit 8 shown in FIG. 2 includes a medium-to-high speed range position estimation unit 81, a low speed range position estimation unit 82, a polarity determination unit 83, a position estimation unit 84, and a rotation speed estimation unit 85.

The medium to high speed range position estimator 81 estimates the rotor position θ HIGH, using the d-axis current Id and q-axis current Iq output from the current converter 7, the d-axis voltage command value Vd*' output from the superimposing unit 15, and the q-axis voltage command value Vq* output from the voltage command value calculation unit 14 _.

For example, the medium to high speed range position estimator 81 estimates the position θ _HIGH using an extended induced voltage e excited by superimposing a superimposed voltage _Vs on a d-axis voltage command value Vd*. A model M _LdLq of the electric motor M including a d-axis inductance Ld and a q-axis inductance Lq as parameters that can be adjusted to any value is shown in the following formula 10. The d-axis inductance Ld in the following formula 10 is a value estimated in advance by an experiment, a simulation, or the like. Also, R is a resistance component of the electric motor M, p is a differential operator, ω is an angular frequency of the rotor (for example, the rotation speed ω^ calculated by the rotation speed estimator 85 at the previous control timing), I is a unit matrix shown by the following formula 11, and J is a unit matrix shown by the following formula 12.
{(R+pLd)I−ω(Ld−Lq)J} ...Formula 10

First, the medium-high speed range position estimator 81 obtains extended induced voltages e _α and e _β corresponding to the difference between the voltages v _α and v _β and the voltage output from the model M _LdLq . The voltages v _α and v _β are input to the motor M, and the currents i _α and i _β output from the motor M are input to the model M _LdLq . The voltage v _α is a voltage obtained by converting the d-axis voltage command value Vd*' output from the superimposing unit 15 into the α-axis of a fixed coordinate system corresponding to the U-phase of the U-phase, V-phase, and W-phase of the motor M. The voltage v _β is a voltage obtained by converting the q-axis voltage command value Vq* output from the voltage command value calculator 14 into the β-axis of a fixed coordinate system advanced 90 degrees from the α-axis. The current i _α is a current obtained by converting the d-axis current Id output from the current converter 7 into the α-axis. The current _iβ is a current obtained by converting the q-axis current Iq output from the current converter 7 into a β-axis current. The extended induced voltages _eα and _eβ are expressed by the following equation 13.

Next, the medium to high speed range position estimating unit 81 estimates the extended induced voltages e _αω and e _βω by performing filtering processing or the like on the extended induced voltages e _α and e _β .

Then, the medium to high speed range position estimating unit 81 calculates the following equation (14) to obtain the position θ _HIGH .

The low speed range position estimator 82 estimates the rotor position θ _LOW using the q-axis current Iq output from the current converter 7 .

For example, the low speed range position estimator 82 estimates the position θ _Low using a high frequency current i _δh included in the q-axis current Iq. The high frequency current i _δh is represented by the following formula 15. Let L ₀ be the value shown in the following formula 16, L ₁ be the value shown in the following formula 17, and Δθ be the error between the position estimated by the mid/high speed range position estimator 81 or the low speed range position estimator 82 and the actual position.

First, the low speed range position estimator 82 performs synchronous detection on the high frequency current i _δh to obtain a value correlated with the position error Δθ as shown in the following formula 18. Note that in the above synchronous detection, the high frequency current i _δh is multiplied by the frequency component sin ω _ht of the superimposed voltage V _s as a demodulated signal, and then filtered by a low pass filter.

Next, the low speed range position estimation unit 82 performs PI control so that the position error Δθ in the above formula 18 becomes zero, thereby acquiring the rotation speed ω _LOW as shown in the following formula 19. Note that Kp is a constant of the proportional term of the PI control, Ki is a constant of the integral term of the PI control, and s is a differential operator.

Then, the low speed range position estimating unit 82 obtains the position θ _LOW by integrating the rotation speed ω _LOW ^ as shown in the following equation 20.

The polarity determination unit 83 includes a multiplication unit 831, a band elimination filter (BEF) 832, an inverse calculation unit 833, a multiplication unit 834, a moving average filter (MAF) 835, and a position error output unit 836, and uses the high-frequency current _iγh contained in the d-axis current Id to determine an evaluation value _eh including the change due to magnetic saturation (magnetic saturation component _Ld1 ) of the d-axis inductance Ld of the motor M, and determines the position error Δθ based on the sign (negative or positive) of the evaluation value _eh .

The high frequency current _iγh is represented by the following formula 21. If the position error Δθ is assumed to converge to 0 degrees or 180 degrees and cos2Δθ≈1, formula 21 is modified to formula 22. If it is assumed that formula 23 holds true for the d-axis inductance Ld of formula 22, formula 22 is modified to formula 24. Let L _d0 be the steady-state component of the d-axis inductance Ld, L _d1 be the change in the d-axis inductance Ld due to magnetic saturation (magnetic saturation component), and id be the d-axis current Id.

The polarity determination unit 83 obtains an equation (equation 5 described later) that positively includes the magnetic saturation component _Ld1 from the above equation 24 as an evaluation value _eh , and determines whether the sign of the evaluation value _eh , i.e., the sign of the magnetic saturation component _Ld1 , is negative or positive.

For example, when the sign of the evaluation value _eh (magnetic saturation component _Ld1 ) is negative, the d-axis inductance Ld on the left side of the above formula 23 becomes relatively small, so it can be determined that magnetic saturation occurs in the motor M and that the direction of the magnetic pole generated by the d-axis current and the direction of the magnetic pole generated by the permanent magnet are consistent with each other. Therefore, since it is assumed that the position error Δθ converges to 0 degrees or 180 degrees, it can be determined that the error between the position estimated by the medium/high speed range position estimator 81 or the low speed range position estimator 82 and the actual position is 0 degrees.

On the other hand, when the sign of the evaluation value _eh (magnetic saturation component _Ld1 ) is positive, the d-axis inductance Ld on the left side of the above equation 23 becomes relatively large, so it can be determined that no magnetic saturation has occurred in the motor M and that the direction of the magnetic pole generated by the d-axis current and the direction of the magnetic pole generated by the permanent magnet do not match each other. Therefore, since it is assumed that the position error Δθ has converged to 0 degrees or 180 degrees, it can be determined that the error between the position estimated by the medium/high speed range position estimator 81 or the low speed range position estimator 82 and the actual position is 180 degrees.

The multiplier 831 multiplies the high-frequency current i _γh shown in the above formula 24 by the frequency component sin ω _ht of the superimposed voltage V _s as a demodulation signal, thereby obtaining the following formula 1.

The band rejection filter 832 obtains the following equation 2 by removing the AC component (1-cos2ω _h t) from the above equation 1. Here, id=i _h sin ω _h t.

The reciprocal calculation unit 833 obtains the following equation 3 by taking the reciprocal of the above equation 2.

The multiplier 834 multiplies the above-mentioned equation 3 by the frequency component sin ω _h t of the superimposed voltage V _s as a demodulated signal to obtain the following equation 4.

The moving average filter 835 extracts the DC component from the above formula 4 to obtain the following formula 5 as the evaluation value _eh .

If the evaluation value _eh is negative, the position error output unit 836 outputs 0 degrees as the position error Δθ to the position estimation unit 84, and if the evaluation value _eh is positive, the position error output unit 836 outputs 180 degrees as the position error Δθ to the position estimation unit 84.

The position estimation unit 84 estimates the position θ̂ using the position θ _HIGH estimated by the medium/high speed range position estimation unit 81, the position θ _LOW estimated by the low speed range position estimation unit 82, and the position error Δθ output from the position error output unit 836.

For example, the position estimator 84 determines the position θ^ by adding the position error Δθ to the weighted average of the positions θ _HIGH and θ _LOW . Note that the position estimator 84 increases the weight corresponding to the position θ _HIGH and decreases the weight corresponding to the position θ _LOW as the rotation speed ω^ calculated by the rotation speed estimator 85 increases. Also, the position estimator 84 decreases the weight corresponding to the position θ _HIGH and increases the weight corresponding to the position θ _LOW as the rotation speed ω^ calculated by the rotation speed estimator 85 decreases.

The rotation speed estimation unit 85 estimates the rotation speed ω^ by dividing the position θ^ calculated by the position estimation unit 84 by a fixed time (such as the clock period of the calculation unit 6).

Thus, according to the control device 1 of the first embodiment, when the evaluation value _eh is negative, i.e., when the d-axis inductance Ld is relatively small, it can be determined that the direction of the magnetic pole generated by the d-axis current Id and the direction of the magnetic pole generated by the permanent magnet are the same direction because magnetic saturation occurs in the motor M, and it can be determined that no error occurs between the position estimated by the medium to high speed region position estimator 81 and the low speed region position estimator 82 and the actual position. Also, when the evaluation value _eh is positive, i.e., when the d-axis inductance Ld is relatively large, it can be determined that the direction of the magnetic pole generated by the d-axis current Id and the direction of the magnetic pole generated by the permanent magnet are different directions because magnetic saturation does not occur in the motor M, and it can be determined that an error of 180 degrees occurs between the position estimated by the medium to high speed region position estimator 81 and the low speed region position estimator 82 and the actual position. In this way, the high-frequency current _iγh contained in the d-axis current Id is used to obtain an evaluation value _eh including the change due to magnetic saturation (magnetic saturation component _Ld1 ) of the d-axis inductance Ld of the motor M, and the presence or absence of an error between the estimated position and the actual position, i.e., the polarity of the rotor, can be determined depending on the sign of the evaluation value _eh .

In addition, in the control device 1 of the first embodiment, as shown in FIG. 2, the medium-to-high speed range position estimation unit 81, the low speed range position estimation unit 82, and the polarity determination unit 83 are connected in parallel to each other, so that the medium-to-high speed range position estimation unit 81, the low speed range position estimation unit 82, and the polarity determination unit 83 can each perform processing simultaneously. Therefore, according to the control device 1 of the first embodiment, it is possible to estimate the rotor position θ^ while determining the polarity of the rotor, and it is possible to shorten the time required for estimating the rotor position compared to a configuration in which the rotor position is tentatively estimated, the rotor polarity is determined, and the rotor position is confirmed using the determined polarity.

In addition, according to the control device 1 of the first embodiment, compared to the case where the polarity of the rotor is determined using the extended induced voltage excited by superimposing a high-frequency current on the d-axis current command value Id*, no processing is required to determine the extended induced voltage, and therefore the load on the calculation unit 6 can be reduced accordingly.

Second Embodiment
Fig. 3 is a diagram showing an example of a motor control device in the second embodiment. Note that the same components as those shown in Fig. 1 are given the same reference numerals, and the description thereof will be omitted.

The control device 1' shown in FIG. 3 differs from the control device 1 shown in FIG. 1 in that it has an estimation unit 8' instead of the estimation unit 8.

The estimation unit 8' uses the d-axis current Id and the q-axis current Iq output from the current conversion unit 7 to determine the polarity of the rotor of the electric motor M and estimate the rotor position θ^.

In addition, the estimation unit 8' estimates the rotation speed ω^ by dividing the position θ^ by a fixed time (such as the clock period of the calculation unit 6).

Figure 4 is a diagram showing an example of an estimation unit 8' in the second embodiment. Note that the same components as those shown in Figure 2 are given the same reference numerals, and their explanations are omitted.

The estimation unit 8' shown in FIG. 4 includes a low-speed region position estimation unit 82, a polarity determination unit 83, a position estimation unit 84', and a rotation speed estimation unit 85'. Note that the control device 1' does not include a medium-to-high speed region position estimation unit 81 in the estimation unit 8', and is therefore suitable for an electric motor M whose rotation speed ω^ is constantly relatively low.

The position estimation unit 84 ′ estimates the position θ̂ using the position θ _LOW estimated by the low speed range position estimation unit 82 and the position error Δθ determined by the polarity determination unit 83 .

For example, the position estimator 84' adds the position error Δθ to the position θ _LOW estimated by the low speed range position estimator 82, and sets the result as the position θ^.

The rotation speed estimation unit 85' estimates the rotation speed ω^ by dividing the position θ^ calculated by the position estimation unit 84' by a fixed time (such as the clock period of the calculation unit 6).

According to the control device 1' of the second embodiment, as in the control device 1 of the first embodiment, the rotor position θ^ is estimated while determining the polarity of the rotor. This reduces the time required to estimate the rotor position compared to a configuration in which the rotor position is tentatively estimated, the rotor polarity is determined, and the determined polarity is used to determine the rotor position.

In addition, in the control device 1' of the second embodiment, the medium-high speed range position estimation unit 81 is omitted in the estimation unit 8', so the load on the calculation unit 6 can be reduced compared to the control device 1 of the first embodiment.

The present invention is not limited to the above embodiments, and various improvements and modifications are possible without departing from the spirit of the present invention.

<Modification>
FIG. 5 is a diagram showing another example of the overlapping unit 15 in the first embodiment.

The superimposition unit 15 shown in FIG. 5 includes an addition unit 151, a multiplication unit 152, and an amplitude output unit 153.

The amplitude output section 153 outputs a larger amplitude _Vh as the rotation speed ω^ decreases, and outputs an amplitude _Vh closer to zero as the rotation speed ω^ increases.

The multiplier 152 multiplies the frequency component sin ω _h t by the amplitude V _h output from the amplitude output unit 153 to output a superimposed voltage V _s .

The adder 151 adds the superimposed voltage _Vs to the d-axis voltage command value Vd* output from the voltage command value calculation unit 14, and outputs the result as a d-axis voltage command value Vd*'.

As a result, the rotor rotation speed ω^ becomes relatively small, and the extended induced voltage e used in the medium to high speed range position estimator 81 becomes relatively small. Even if the estimation accuracy of the position θ _HIGH estimated by the medium to high speed range position estimator 81 decreases, the estimation accuracy of the position θ _LOW estimated by the low speed range position estimator 82 can be improved, and therefore, it is possible to suppress a decrease in the estimation accuracy of the position θ^ estimated by the position estimator 84.

Reference Signs List 1, 1' Control device 2 Inverter circuit 3 Control circuit 4 Memory unit 5 Drive circuit 6 Calculation unit 7 Current conversion unit 8 Estimation unit 9 Subtraction unit 10 Torque command value calculation unit 11 Current command value output unit 12 Subtraction unit 13 Subtraction unit 14 Voltage command value calculation unit 15 Superimposition unit 16 Voltage command value conversion unit 81 Medium to high speed range position estimation unit 82 Low speed range position estimation unit 83 Polarity determination unit 84, 84' Position estimation unit 85, 85' Rotational speed estimation unit P Power supply C Capacitors Se1, Se2 Current sensor M Electric motor

Claims (3)

an inverter circuit that drives an electric motor based on a comparison result between a voltage value of the carrier wave and a voltage command value;
an estimation unit that estimates a rotor position using an evaluation value including a change due to magnetic saturation of a d-axis inductance of the electric motor;
a current converter for converting a current flowing through the electric motor into a d-axis current and a q-axis current using the position;
a current command value output unit that outputs a d-axis current command value and a q-axis current command value according to a difference between a rotation speed of the rotor and a rotation speed command value;
a voltage command value calculation unit that calculates a d-axis voltage command value so as to reduce a difference between the d-axis current and the d-axis current command value, and calculates a q-axis voltage command value so as to reduce a difference between the q-axis current and the q-axis current command value;
a superimposing unit that superimposes a superimposed voltage having a higher frequency than the d-axis voltage command value on the d-axis voltage command value;
a voltage command value conversion unit that converts the d-axis voltage command value output from the superimposition unit and the q-axis voltage command value output from the voltage command value calculation unit into the voltage command value by using the position;
Equipped with
The estimation unit is
a medium to high speed range position estimating unit that estimates a position of the rotor using an extended induced voltage excited by superimposing the superimposed voltage on the d-axis voltage command value;
a low speed range position estimator that estimates a position of the rotor using a high frequency current included in the d-axis current;
a polarity determination unit that sets a position error between the position estimated by the medium/high speed range position estimation unit and the low speed range position estimation unit and an actual position to 0 degrees when the evaluation value is negative, and sets the position error to 180 degrees when the evaluation value is positive;
a position estimator that adds a weighted average of a position estimated by the medium/high speed range position estimator and a position estimated by the low speed range position estimator to a position error determined by the polarity determination unit, and sets the result as a position used in the current conversion unit and the voltage command value conversion unit;
A control device for an electric motor comprising: The motor control device according to claim 1,
The estimation unit is
The high frequency current included in the d-axis current is iγh, the amplitude of the superimposed voltage is Vh, the angular frequency of the superimposed voltage is ωh, the d-axis inductance of the motor is Ld, the q-axis inductance of the motor is Lq, Ld0=(Ld+Lq)/2, Ld1=(Ld-Lq)/2, the d-axis current is id, id=ihsinωht, the high frequency current is iγh, and
In the case where the symbol BEF indicates a value obtained by removing AC components from a value, and the symbol MAF indicates a value obtained by extracting DC components from a value,
When the high frequency current is multiplied by sinωht, which is the frequency component of the superimposed voltage as a demodulation signal, to obtain the following formula 1, the AC component (1-cos2ωht) is removed from the following formula 1 by a band rejection filter to obtain the following formula 2, the reciprocal of the following formula 2 is obtained as the following formula 3, the following formula 3 is multiplied by the sinωht to obtain the following formula 4, and the DC component is extracted from the following formula 4 by a moving average filter to obtain the following formula 5,
The value on the left side of the following formula 5 is the evaluation value.

A control device for an electric motor. The motor control device according to claim 1,
the position estimation unit increases a weight corresponding to the position estimated by the low speed range position estimation unit as the rotation speed of the rotor decreases,
The motor control device, wherein the superimposing unit increases the amplitude of the superimposed voltage as the rotation speed of the rotor decreases.

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