JPS6030196B2 - Power supply circuit for magnetic phase shifter - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to a power supply circuit for an air phase shifter, and more particularly to a power supply circuit for a magnetic phase shifter suitable for obtaining linear magnetic phase shift characteristics with a small input.

When the output of a magnetic amplifier is used as a control signal for a semiconductor with a control pole such as a thyristor, the output of the semiconductor with a control pole such as a thyristor can be controlled by the control input of the magnetic amplifier. It is simple and robust; by providing control inputs in the form of control currents or controlled magnetomotive forces, many control inputs can be added and subtracted in isolation from each other, and the results can be amplified; and they are not susceptible to noise signals. In particular, the characteristics of semiconductors, such as the ability to efficiently control large amounts of power with a relatively small device, are combined to provide a power control device suitable for industrial use.

A magnetic amplifier used in this manner is called a magnetic transfer device in the sense that the firing angle of a thyristor or the like is determined by its output conduction angle. Of course, it is desirable for this energy phase shifter to be one that can be controlled by 4 inputs as much as possible and has excellent phase shifting characteristics. In a transfer device, the linearity of the phase shift characteristic tends to be poor due to the poor squareness ratio of the iron core.

Therefore, a method has been proposed in which the linearity of the phase shift characteristic is improved by using a pulsed voltage at the rising edge as a power supply for the magnetic phase shifter, but the design is complicated and it is difficult to obtain stable characteristics. It has drawbacks such as being difficult and making the circuit complicated. Furthermore, in the conventional method, when the iron core of the magnetic phase shifter is sufficiently saturated, the above-mentioned pulse voltage may be applied to the load, which has the disadvantage that the load voltage becomes excessive.

SUMMARY OF THE INVENTION An object of the present invention is to provide a power supply circuit which is simple in design, has stable characteristics, is simple in circuit, and is suitable for improving the characteristics of an air phase shifter.

The present invention connects a saturation reactor to an AC power supply for a magnetic phase shifter, and combines the pulse voltage induced in the winding of the saturation reactor with the AC power supply voltage to excite the magnetic phase shifter. After the iron core is saturated, this pulse voltage is reduced to zero. FIG. 1 shows an embodiment of the present invention, in which an AC power source Ea is connected through a resistor Ra.
The base emitter of the transistor Tr, and the base of the transistor Tr2.

A current flows through the circuit of the diode Dd2 connected in anti-parallel to the emitter, or the circuit of the diode Dd connected in anti-parallel to the base emitter of the transistor Tr2 and the base emitter of the transistor Tr. Further, from the DC power source Ed, primary windings N, . , the collector of transistor Tr,
A current flows through the emitter circuit, the primary winding N2 wound around the iron core C, and the collector-emitter circuit of the transistor Tr2. Secondary winding N2 wound around iron core C,
, and the iron core C of the saturation reactor SL connected in series with this
From the secondary winding n2, wound around the magnetic phase shifter M
Current flows through the circuit of the output winding NL, diode Dd3, and load RL of the PS, and the current flows through the secondary winding N22 wound around the iron core C of the PS and the iron core C2 of the saturation reactor SL, which is connected in series with the secondary winding N22. A magnetic phase shifter MP is connected to the wound secondary winding.
A current flows through the circuit including the output winding NL2 of S, the diode Dd4, and the load RL2. Furthermore, 3 wound around the iron core C,
From the next winding N3, a saturation reactor SL is passed through a resistor Rs.
Current flows through the primary winding n, wound around the iron core C2.
In addition, a diode Dd is connected to both ends of the secondary winding n2 and handle 2.
5, Dd6 are butt-connected, and the collectors and emitters of transistors T and others are connected between the butt point, the secondary winding mountain, and the connection point of n, and the base and emitters of transistor Tr3 are connected to the load. R.L. , RL2, a current flows through the diodes Dd7, ○&'' and the resistor RB. In addition, a control input is applied to the phase shifter M gate by the control winding Nc. Note that all the windings have . represents that the polarity is the same, and the windings N,. and N,2;N
2, N22; and crest 2 have the same number of turns. In such a configuration, for example, if the AC power source Ea has the polarity as shown in the figure, a base current flows to the transistor Tr through the resistor Ra, the base emitter of the transistor Tr, and the diode D.
The collector and emitter of the transistor Tr are short-circuited. As a result, the DC power source Ed is 1 of the power transformer T.
Next winding N, . EO power is applied to Here, the primary windings N, . ,N. Secondary winding N2 for 2,,
If the winding ratio of N22 is Q, then a voltage QEd with the * mark in the positive direction will be induced in the secondary winding N2. Also, the primary windings N, . If the turns ratio of the tertiary winding N3 to , N, 2 is 6, a voltage BEd with the * mark in the positive direction is induced in the tertiary winding N3, but this voltage passes through the resistor Rs to the saturation reactor SL is applied to the primary winding n, of Therefore, "the turns ratio of the secondary winding n small n22 to the primary winding n of the saturation reactor SL is set,
If the voltage drop across the resistor Rs during the non-saturation period of the saturated reactor SL is ignored, a voltage BYEd occurs in the secondary winding w, with the mark . And the secondary winding N2,
The induced voltage QEd of the secondary windings and the induced voltage ByEd of the secondary windings are in the same direction and in the forward direction with respect to the diode Dd3. , to the output winding NL of the magnetic phase shifter MPS, (QEd+8yEd)=(Q+8y)Ed
A voltage is applied. However, diode Dd3, load R
The voltage drop at L, is ignored. Therefore, AC power supply E
Magnetic phase shifter M during the period when a is of opposite polarity to that shown
If the PS magnetic flux reset amount is △, the output winding NL,
The time △t from when a voltage is applied to until the iron core (not shown) of the magnetic phase shifter M is saturated is △t:N and △? /(
Q+8y)Ed...(1}).

However, it is assumed that the saturated reactor SL is in a non-saturated period.
When the iron core of the magnetic phase shifter MPS is saturated, the total value of the induced voltages of the secondary windings N2 and n2 is almost applied to the load R]. However, voltage drops in the output windings NL, diode Dd3, etc. are ignored. Then,
Since the base current flows from both ends of the load RL to the transistor Tr3 through the diode Dd7 and the resistor RB, the collector and emitter of the transistor Tr3 are short-circuited, and the secondary winding peak, the diode D, and the transistor T
3, and if the voltage drop across the diode Dd5 and transistor Tr3 is ignored, the voltage becomes almost zero. Therefore, when the iron core of the magnetic phase shifter M-circle S is saturated, the voltage applied to the load RL is approximately equal to the induced voltage of the secondary winding N2, and its value becomes QEd as described above. To summarize what has been said above, according to the embodiment shown in FIG. When the core of the phase converter MPS is saturated, the load RL has a Q
A voltage Ed will be applied.

Therefore, the relationship between the voltage waveforms of each part is as shown in FIG.
When the polarity of the AC signal Ea is reversed from that shown in Fig. 1, a voltage (Q+8y)Ed is applied to the output winding NL2 until the iron core of the magnetic phase shifter M is saturated. It can be easily inferred from the above explanation that when the iron core of M circle S is saturated, a voltage QEd will be applied to the load RL2.

On the other hand, in Fig. 11, if the saturation reactor SL is not used, the time △t' until the iron core of the magnetic phase shifter MPS is saturated for the same magnetic flux reset amount △ is △t' = N
L. Since △○/QEd...{21, it is clear that △t'>△t when compared with formula m.

By the way, the time tc of the voltage applied to the load RL is given by t - △, where t is the half-cycle time of the AC power supply Ea, and △t is the time until the magnetic phase shifter MPS is saturated.
It becomes t.

If this △t is expressed by the formula {1}, wide: t-N
] △○/(Q18y) Ed ...{1)', and if expressed by the formula ■, then = t-NL, △○/QEd
...{21'.

In an actual iron core, △ is a component △ that changes with the controlled magnetomotive force of the magnetic phase shifter? c, and the squareness ratio of the iron core is 1 as described later.
It is the sum of the difference Δ from the residual magnetic flux to the saturation magnetic flux, which is caused by the smaller r.

Due to this Δr, the maximum value during the output time is limited regardless of the control magnetomotive force. In other words,
The phase shift range due to the controlled magnetomotive force of the magnetic phase shifter is tc=0~
{t-NLΔ0r/(power supply voltage)}. Than this,
According to this embodiment, for the same magnetic flux reset amount of the magnetic phase shifter M-circle S, the time width of the voltage applied to the load RL becomes wider.

In other words, it is clear that this has the effect of expanding the phase shift control range. In this embodiment, the constants of the saturation reactor circuit can be designed as follows, for example. That is, if the iron core of the air phase shifter MPS has an ideal squareness ratio of 1, the first
In the figure, if the input from the control winding Nc is zero, the output winding NL, is excited in only one direction through the diode Dd3, and the output winding NL2 is excited in one direction through the diode Dd4, respectively, for every half cycle of the AC signal Ea. However, since the squareness ratio is 1, the residual magnetic flux and the saturation magnetic flux are equal. In other words, magnetic flux reset amount △? is zero, and even if there is no saturation reactor SL, Δt should be 0. but,
In actual iron cores such as supermalloy, which are particularly suitable for small input applications, the squareness ratio is less than 1, so even if the core is energized in one direction by the output winding every half cycle as described above and saturated, The magnetic flux returns to the residual magnetic flux during the half cycle when it is not excited.

So, the difference between saturation magnetic flux and residual magnetic flux is △? In order to change the residual magnetic flux from the residual magnetic flux to the saturation magnetic flux during the time △t, using the formula {1}, 8y=△hime d/Nu△? r-Q
...(3'), so the turns ratio 8, y should be designed to satisfy this relationship.

Also, since the non-saturation period of the saturation reactor SL needs to be longer than △t, if the saturation magnetic flux of the iron core C2 is 0s, then n, 4s and BEd△t
”・[Must be 4'.

In the end, '31, n. OS/82Ed△t
…[Number of turns n, saturation magnetic flux so that it becomes 51? Determine the relationship between s and turns ratio 6, and y=(△field d/N
L, △? y may be determined from r-Q)/P...■.

In this way, according to this embodiment, by providing the necessary magnetic flux change amount and change time, it is possible to easily and accurately design the constants of the saturation reactor circuit.

In addition, according to this embodiment, D is also ~Dd8, Tr3, and after the iron core of the magnetic phase shifter S is saturated as shown in FIG. Since the voltage is short-circuited to zero, there is no risk of excessive voltage being applied to the loads RL, RL2.

In addition, in the above description, when the magnetic phase shifter MPS is saturated,
For example, when a voltage is applied to the load RL, the collector and emitter of the transistor Tr3 become short-circuited, and the
Next winding n2, diode Dd5, transistor Tr
3, but if the value of load RL, is relatively small, the induced voltage in secondary winding N2 increases the current flowing through load RL, and therefore secondary winding -, and this causes When the saturation reactor SL is excited from the secondary side, the polarity of the electromotive force of the secondary windings is reversed, and the side without the mark becomes positive.

In this case, a voltage is also induced in the secondary winding ~2 wound around the same iron core C2, with the side without the mark having positive polarity, so the diode D is also
The secondary winding n22 is short-circuited and therefore the secondary winding n2. will also be equivalently shorted. In this way, according to this embodiment, the magnetic phase shifter MPS
After the iron core of saturating reactor SL is saturated, the secondary windings N2, , n22 of the saturation reactor SL will definitely be short-circuited, regardless of the characteristics of the induced voltage in the secondary windings n2, , n22. This has the effect of ensuring that a predetermined voltage is applied. Although the above description has been made for the case where the AC power source $a has the polarity shown in FIG. 1, it is clear from the symmetry of the circuit that the same operation is performed in the case of reverse sacrifice.

Of course, the original purpose of D can be achieved even without Dd8, Tr3, and R8. FIG. 3 shows another embodiment of the present invention, and the difference from FIG. 1 is that the power transformer T has a tertiary winding N3.
, the primary winding n, of the saturation reactor SL is the same as the primary winding N, . , N,2 is excited by the voltage across both ends.

In this way, the primary windings N, . ,N
, 2 and the turns ratio 8 of the tertiary winding N3 is equivalently 8=2, and the '5} and '61 formulas are n, 0s/22Ed, respectively.
△t...'5ry=(△hime d/NL,
△Gr-Q)/2 ...(6)', and this 2
The winding n, the saturation magnetic flux ◇s, and the turns ratio y can be determined from the formula.

According to this embodiment, the structure is simple because the winding N3 is not required, and since the turns ratio 3 is constant, there is an advantage that the circuit constants of the saturation reactor SL can be designed even more easily.

FIG. 4 shows another embodiment of the present invention, and the difference from FIG. 3 is that a parallel circuit of a capacitor Cs and a resistor Rs' is connected in series with the resistor Ks.

In such a configuration, if the value of the resistor Rs' is infinite, the primary windings n,
A current flows through the circuit until capacitor C3 finishes charging. For example, the primary windings N, . The one with the mark is 1
In the half cycle in which the polarity is positive with respect to the side of the secondary winding N, 3 that is not marked, the primary winding n, has i in Fig. 4.
A current flows as shown in FIG. 4, but in the previous half cycle, the capacitor Cs was charged with the polarity shown in FIG. , N, 2 is equal to d. Therefore, in the half cycle in which the current i flows as shown in Fig. 4, the time constant RsCs of this circuit is the saturation reactor S
If the length of the non-saturation period of the core C2 of L is selected to be sufficiently large compared to the length of the non-saturation period of the core C2, the primary windings N, . ,N,2
The sum of the induced voltage unevenness d across both ends of the capacitor Cs and the voltage d charged in the capacitor Cs is added. This corresponds to the primary windings N, . , N, 2 and the turns ratio of tertiary winding N3 is 8.
=4. And '51, '6} formulas are n, 0s/42Ed△t, respectively.
A5) ″y=(△old d/NL, △Jr-o)/4
...{6r, and it is sufficient to determine the winding n, the saturation magnetic flux s, and the turns ratio y from these two equations, as in the case of the embodiment shown in FIG.

Note that if the value of the resistor Ks is set to a finite value, the voltage charged to the capacitor Cs can be adjusted to be less than d. According to this embodiment, as in FIG. 3, not only the configuration is simple and the design is easy, but also the current stops flowing through the primary winding n when the capacitor Cs is charged. In the embodiment shown, after the core C2 of the saturation reactor SL is saturated, the primary winding n is passed through the resistive snake s during its half cycle.
, the power loss can be significantly reduced compared to when the current continues to flow through the transistors Tr,
, Tr2, and the capacity of the DC power supply Ed can also be reduced.

FIG. 5 shows another embodiment of the present invention, and the difference from FIGS. 1, 3, and 4 is that the primary winding n of the saturation reactor SL
, are points directly excited by the AC power source Ea.

This example can be applied when the capacity of the AC power source Ea is sufficient to excite the saturation reactor SL.

According to this embodiment, compared to the other embodiments described above, it is not necessary to supply power for exciting the saturation reactor SL from the power transformer T, so that the power transformer T, the transistor Tr,
This has the effect that the capacity of the DC power source Ed can be minimized. FIG. 6 shows still another embodiment of the present invention, in which an AC power source Ea is connected through a resistor Ra to a diode D connected in inverse parallel to the base-emitter of the transistor Tr and the base-emitter of the transistor Tr2. A current flows through the circuit, or the circuit of the diode Dd, which is connected in antiparallel to the base emitter of the transistor Tr2 and the base emitter of the transistor Tr, and the transistor Tr changes depending on the polarity of the AC power source Ea.
．． The first reason is that Tr2 is alternately shorted and opened.
This is the same as in the embodiment shown in the figure.

However, the output winding N of the magnetic phase shifter S is connected to the transistor Tr through a diode D, and the output winding NL of the magnetic phase shifter S is connected to the transistor Tr2 through a diode Dq. When Tr, is in a short-circuited state, the sum of the secondary winding of the saturation reactor SL and the DC power source Ed is applied to the output winding NL, through the load RL and the diode Dd5, and when the transistor Tr2 is in a short-circuited state, the output winding NL, The sum of the secondary winding 2 of the saturation reactor SL and the DC power source Ea is added to NL2 through the load RL2 and the diode DQ. On the other hand, the transistors Tr, Tr2 also have diodes D connected to saturation reactors SL via Dd and o, respectively.
The primary winding n. ,n,' (the number of turns is equal to n) are connected, and the midpoint of the primary winding n,,n,' is a resistor Rs.
Since it is connected to the DC power source d through the transistor Tr. , Tr2 are short-circuited, a voltage Ed is applied to the primary windings n, , n,' through the resistor Rs.

Since the present embodiment has the above-described configuration, the voltage applied to the output windings NL, NL2 is equal to the turns ratio Q and 3 of the power transformer T in the embodiment shown in FIG.
This is equivalent to expanding the phase shift control range of the magnetic phase shifter M, and it is economical because the power transformer T is not required. Moreover, 2 of the saturation reactor SL
Diodes Dd5 and D4 are connected to the next winding peaks and peaks 2 and 2, respectively.
The transistor Tr3 is connected through the loads RL, .
Diodes Dd7 and Dd8 are connected by the output generated in RL2.
, the transistor Tr3 is short-circuited via the resistor RB, as in FIG. 1, and no excessive voltage is applied to the loads RL, RL2. FIG. 7 shows still another embodiment of the present invention, in which the transistor Tr3 and the resistor R are shown in FIG.
8. Except for diodes Dd7 and Dd8, transistor T
Diodes Dd, . is the saturation reactor S
The difference is that it is connected in antiparallel to the output of L, but the other points are the same as in FIG.

In this embodiment, until the magnetic phase shifter MPS generates an output, that is, the secondary winding ratio, the diode Dd5,
DC power supply Ed, transistor Tr, diode Dd3
, output winding NL, , load L. , or secondary winding Q2, diode Dd6, DC power supply Ed, transistor Tr2,
While the excitation current of the magnetic phase shifter MPS is flowing in the circuit consisting of the diode D4, the output winding NL2, and the load R2, the diodes Dd, . Since the voltage of the secondary winding & or peak 2 is applied in the opposite direction, it is in an open state, which is equivalent to the transistor Tr3 being in an open state in FIG.

The iron core of the air phase shifter MPS is saturated and the loads RL,,RL
The current flowing through the diodes Dd, . If it is in the conductive state, it passes through it and returns to the power source d. Diode D
d,. is conductive, so if you ignore the forward voltage drop, the voltage across it is zero. Therefore diode D
The cathode of Dd6 and the connection point of the secondary winding, side 2, will be short-circuited. This is equivalent to the transistor Tr3 being in a short-circuited state in FIG.
Then, the short circuit current is flowing through the secondary winding is . IE
d...[7}IS=completed/horse.

On the other hand, the magnetic phase shifter MPS is saturated and the loads RL,,RL2
The current iL flowing through the diodes Dd, . Since it is assumed that is the optimal state, . Ed
...(8}1L: Decline and decline.

However, RL is the value of the loads RL, , RL2. Therefore, the diodes Dd, . In order for the above-mentioned assumption that is in the optimal state to hold true, it is sufficient that iL is larger than is and the following condition is satisfied. Agriculture > Temple head name'/R3 > Ward 2
--・[It should be set to 91y.

In this embodiment, Rs, RL
and shall be set. According to this embodiment, not only the same effects as in FIG. 6 can be obtained, but also the circuit can be simplified because the transistor Tr3 and the like are not required.

In addition, as in the embodiment shown in FIG. 4, a capacitor C is connected to the resistor Rs.
If the diodes Dd, . It is easier to keep it in optimal condition.

FIG. 8 shows still another embodiment of the present invention, in which the secondary windings of the saturation reactor SL in FIG.
d6, Dd, . Instead, a bridge-connected diode Dd, 2 is used, and the rest is the same as in Fig. 7.As can be seen from Fig. 2, it is not clear that the diodes Dd, 2 are connected in a bridge. Since the voltage is generally relatively high,
Although the number of turns of the secondary windings is quite large, in this embodiment, only one set of secondary windings is required, so the number of windings is 1'2, which is economical.

As described above, according to the present invention, as a power supply circuit suitable for improving the characteristics of a magnetic phase shifter, it is possible to obtain a simple design, stable characteristics, and a simple circuit.

Also, taking FIG. 1 as an example, Dd5~D&, Tr3, RB
Providing this has great practical effects, such as preventing the voltage applied to the load from becoming excessive.

[Brief explanation of the drawing]

FIG. 1 is a diagram showing an embodiment of the present invention, FIG. 2 is a diagram showing its operating waveforms, FIGS. 3, 4, 5, 6, and 7.
8 are diagrams showing other embodiments of the present invention. Explanation of symbols, Ea... AC power supply, Tr,, Tr2.
...Transistor, Tr3...Transistor, Ed...DC power supply, T...Power transformer, M
PS...Magnetic phase shifter, SL...Saturation reactor, RL., RL2. ．．・．． ．．・Load, Dd, . ,Dd,
2...Diode. Kaya'inhke series 2 figures Tsutomu 3 figures 4 figures 5 figures poly' figures 7 figures 8 figures

Claims (1)

[Scope of Claims] 1. A device comprising a magnetic phase shifter and a power source for exciting the magnetic phase shifter, which includes a saturation reactor excited by the power source, and a winding of the saturation reactor is connected to the power supply voltage. A power supply circuit for a magnetic phase shifter, characterized in that the magnetic phase shifter is excited by superimposing a line voltage. 2. The power supply circuit for a magnetic phase shifter according to item 1, further comprising a semiconductor switch driven by the output of the magnetic phase shifter, and the semiconductor switch is connected in parallel to the winding of the saturation reactor. A power supply circuit for a magnetic phase shifter. 3. In the power supply circuit for a magnetic phase shifter according to the first aspect, a second diode is connected antiparallel to the series body of the winding of the saturation reactor and the first diode, and the output circuit of the magnetic phase shifter is a winding of the saturation reactor, when the second diode is connected in the forward direction and the output current of the magnetic phase shifter flows through the second diode;
A power supply circuit for a magnetic phase shifter, characterized in that a short circuit current of a saturation reactor flowing through the first diode and the second diode is smaller than an output current of the magnetic transfer device.