JPS6035914B2 - Control method of induction motor - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION (Field of Industrial Application) The present invention relates to an improvement in transformer vector control that allows an induction motor (hereinafter simply referred to as a motor) to have performance similar to that of a DC motor.

(Prior art and its problems) In the conventional transformer vector control of an induction motor as disclosed in Japanese Patent Laid-Open No. 51-11125, for example,
Command current (excitation current component command) io that provides secondary magnetic flux, command current proportional to torque (torque current component command) i,
q (hereinafter referred to as q-axis component primary current command value).

) is applied in an open loop, but if the secondary resistance fluctuates due to temperature changes, the generated torque may fluctuate,
There was a drawback that the linearity between the generated torque and the torque command value was lost. As a method to improve this drawback, attempts have been made to make the slip angular frequency s proportional to the torque, but the goal of transformer vector control is torque linearity and no time delay between the generated torque and current. In other words, since ws has no direct relation to quick response, it was not very effective. (Means for solving problems)
Therefore, as a result of various experimental studies, the present inventors determined that the slip frequency command (T2') obtained from the excitation current component command io and the torque current component commands i, t, and the rotor angular velocity of the induction motor. A two-phase sine wave signal having a secondary magnetic flux angular frequency o of the induction motor obtained by adding and excitation current component detection value i
In a control method for an induction motor in which the primary current of the induction motor is commanded from signals i and d corresponding to the deviation from o',
The performance of transformer vector control is improved by determining the deviation between the torque current component commands i, q and the detected torque current component value iM', and adding a signal corresponding to this deviation to S of the slip frequency command. It was extremely successful.

(Embodiment) The embodiment shown in the drawings will be described below.

FIG. 1 is a block diagram showing the overall configuration of an embodiment of the present invention, in which 1 is a controlled three-phase induction motor (hereinafter simply referred to as the motor).
It is.

The motor 1 has each primary current command value i side for the q-axis and d-axis components.
It is operated via a current intensifier 3 by a current calculation circuit 2 which creates control currents ia to ic based on i and d'.

As the q-axis component primary current command values i, q, a signal corresponding to the speed deviation passed through the arithmetic intensifier 4 is used, and as the d-axis component primary current command values i, d, the excitation current component command io is used. and the excitation current component detection value io', and the current calculation circuit 2 performs calculations based on the equation (2) below from the primary current command values i, q and i, d to calculate the control currents ia to ic. Sent out.

Next, for convenience of explanation, each relational expression will be explained.

Now, the electrical angular velocity of the secondary magnetic flux of motor 1 is ■.

Then, of. The voltage balance equation of the secondary circuit of the motor seen from the d-q axis coordinate system rotating at is expressed by the following equations {1- and ■. However, it is assumed that no voltage is applied to the secondary circuit from the outside and that it is short-circuited, and that the q-axis rotates at a rad lead than the d-axis. PMi from the equilibrium equation about the d-axis.

10R2i2d=0 ・・・・・・・・・
・[1} In the above formula, P is a differential operator and indicates poison. M is the mutual inductance between the secondary and the secondary, R2 is the secondary resistance, io is the current that provides the secondary magnetic flux that coincides with the d-axis, and i2d is the secondary q-axis current. Also, S●M,i from the voltage balance equation for the q-axis.

10R2, ilqni. ......■ In the above equation, i2q is the secondary q-axis current, s is the slip angular frequency, and the electrical angular velocity of the motor rotor is in the following relationship. s = o - h ``''''''''{3} Also, the relationship between jo and the primary and secondary currents is expressed by the following equation.

Regarding the d-axis, i.

=i. d+band i2d ‐‐‐‐‐‐(4, regarding the q axis.=i.q Tokii square ‐‐‐‐‐‐'5, however, L
is the second-order self-inductance from the equations ① and ``5};

......■ However, t'maagi time constant good ochi T2 = army Further, the generated torque 7 is expressed by the following equation.

3 M”……{7}7=back n
・FI.

1・q However, n is the number of pole pairs and the motor has three phases.

Therefore, when jo, i, and q shown in equation (2) are given, the torque d is linear with i, q.

Further, if io is constant, torque 7 will be proportional to i and q. Also, from equations [6} and [7}, the slip angular frequency s is proportional to the torque 7. The d-axis component primary current is {1}, [4
From equation 1, it is expressed by the following equation. IM=i. 10T2. P・1
0 """{8}The actual primary current ia, ·b, iC for motor control is expressed by the following equation.
As mentioned above, the current calculation circuit 2 performs the calculation based on the above formula [9}. In order to perform the above calculation, the current calculation circuit 2 uses a circuit that converts the input signals COSwot and sm ot into three phases. , a multiplier that multiplies DC signals by i, d, i, and q, and an adder.

Then, the output currents ia to ic from the current calculation circuit 2 are increased by the current multiplier 3 to a current sufficient to drive the electric motor 1.

Conventionally, the q-axis component primary current commands i, q, which are proportional to the speed deviation, are obtained by an arithmetic intensifier 4, and then the slip angular frequency S is generated by an arithmetic circuit based on equation (6), and the slip angle is Frequency s and d-axis and q-axis component primary command current i,
Since the motor 1 was controlled by giving the actual current command obtained from calculations based on formulas '8] and [9} from d, i, and q,
When the secondary resistance R2 changes according to the temperature change of the electric motor,
Since the slip angular frequency s is forcibly fixed, (
Since io, i, and q in Equation 61 have to change, the linearity of the torque with respect to i and q shown in Equation [7} may collapse, or even if a constant torque command is given, the linearity may change depending on the temperature of the motor. The disadvantage is that different torques are generated. 3 * combinations were occurring.

For example, tension control that requires torque control, such as in a rolling mill, was not very good. In this embodiment, the line current of the motor 1 is detected by the current detector 10, and the detection circuit 11 calculates the angular velocity from this line current. The d-axis component rotated by 1 of the d-axis component
The next detected currents i, d', i, q' are calculated and determined, the deviation between i, q and i, q' is calculated in the calculation intensifier 8, and this deviation is used as the slip frequency command in the calculation intensifier 9. s = Obimono calculation and desk / i. I have a hunch about the autumn deviation. The disadvantages of the conventional method are improved by passing the calculated values through a differentiator 13 and an arithmetic intensifier to determine i and d.

Since i, d', i, q' and ia, ib are expressed by the following equations, using detection currents ia, ib, cos ot, and sin ot signals, i is calculated by a circuit consisting of a multiplier and an adder. ，
d', i, q' can be obtained.

FIG. 2 is a detailed diagram of the detection circuit 11, in which 21 to 23 are arithmetic circuits, 24 to 27 are multipliers, 28 is a polarity inverter, and R is a resistor.

Now, in the device of the present invention, first, the arithmetic intensifier 7
The deviation between the command value and the detected value jo' obtained through the first-order delay element 12 based on formula (8) is increased,
The arithmetic intensifier 5 adds it to the lo command value, and the differentiator 13 and the arithmetic circuit 6 add it to the d-axis component 1 that satisfies the formula '8}.
It is automatically controlled so that the actual excitation current component io is always equal to the excitation current component io command by giving it to the current calculation circuit 2 as the next current command value itd.

On the other hand, the output i, q of the arithmetic intensifier 4 and the io command are used to calculate the equation (2) using the divider 14, and the arithmetic intensifier 8 increases the deviation between the command current itq and the detected current itq'. Then, the arithmetic multiplier 9 adds the ws to the oscillator 15.

Therefore, if the secondary resistance R2 is now increased, io
is constant, so if S from equation [6} is constant, i will decrease and the torque will decrease, but in this case, on the contrary, the secondary resistance R
Since S changes with respect to the fluctuation of 2, it works to keep the detected value itq' constant. In other words, the secondary resistance increases,
When the detected current iM' becomes smaller, the output of the arithmetic intensifier 8 becomes larger, so S becomes larger, and i, q' become larger, so that the detected current i, q' becomes equal to the command current i, q. automatically controlled so that Therefore, in this device, {7
}A torque proportional to the command currents i and q shown in the formula is generated.

The conventional device disclosed in Tokukan Sho 51-11125 lacks 5, 7, 8, 9, 11 and 12 in Fig. 1, and does not have open loop control for inu, itq or io. Therefore, in addition to the torque fluctuation caused by the change in the secondary resistance R2 mentioned above, there were the following problems.

That is, the characteristics of the current multiplier 3 are as follows.

The higher the value, the worse the torque becomes, and there is a drawback that the torque linearity is similarly lost when a current lower than the command value flows.
Further, when the secondary resistance R2 fluctuates or the frequency characteristics of the current multiplier 3 are poor, io fluctuates, making it impossible to obtain a high gain in the overall speed loop. Furthermore, if the linearity of the ws oscillator 15 is poor, the torque linearity may be distorted. However, with the configuration of the present invention, it is possible to extremely minimize disturbance factors, and the loop gain of the speed feedback loop can be set high, thereby improving the performance, which makes it possible to control induction motors for tension control that require torque control. This is a suitable method.

Note that if the detection circuit is configured as shown in Figure 2,
Detection of io primary current and cos.

Above, sin. Since the calculation can be performed using the signal of t, there is a practical advantage that it can be applied even to squirrel cage induction motors in which only the primary current can be detected. FIG. 3 shows a different embodiment of the present invention in which i, d negative feedback control is used instead of the io negative feedback control shown in FIG. 1, but its operation is essentially the same as that shown in FIG. 1. This is the same as the embodiment shown in .

Furthermore, in the embodiment shown in FIG. 1, if the operational multiplier 4 is omitted and no speed feedback circuit is provided,
Torque control is controlled by the io command and the i and q commands.

BRIEF DESCRIPTION OF THE DRAWINGS FIG. 1 is a block diagram of an embodiment of the present invention;
FIG. 2 is a detailed circuit diagram of the detection circuit, and FIG. 3 is a block diagram of a different embodiment.

DESCRIPTION OF SYMBOLS 1...Three-phase induction motor, 2...Current calculation circuit, 3...Current multiplier, 4-9...Calculation multiplier, 10... ...Current detector, 11...
Detection circuit, 12...1st-order delay element, 13...
...Differentiator, 14...Divider, 15...
S oscillator, 16... Frequency adder, 17...
...Single-phase to two-phase converter, 18...Tachogenerator.

Figure 2 Figure 3 figure Gap

Claims (1)

[Claims] 1 Excitation current component command i_0 and torque current component command i_
Slip frequency command ω_s obtained from 1_q = (i_1_
q)/(T_2i_0) (T_2 is a secondary time constant) and a two-phase sinusoidal signal cosω_0t, sinω_0t, which has a secondary magnetic flux angular frequency ω_0 of the induction motor obtained by adding the rotor angular velocity θ of the induction motor; a signal i_ corresponding to the deviation between the torque current component command i_1_q and the excitation current component command i_0 and the excitation current component detected value i_0';
In the induction motor control method in which the primary current of the induction motor is commanded from 1_d, the torque current component command i_
1_q and the detected torque current component value i_1_q', and a signal corresponding to this deviation is used as the slip frequency command ω_
A control method for an induction motor, characterized in that the control method adds the value to s.