JPS6220796B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION [Field of Industrial Application] The present invention relates to a torque control method when operating an induction motor using a variable frequency power source such as an inverter.

[Prior art and its problems]

Conventionally, in induction motors powered by a variable frequency power supply, a slip frequency control method has been generally used to achieve torque control and response similar to that of separately excited DC motors, but this method is based on the secondary current i ₂ and 2
This is based on the relationship that the voltage drop due to the next resistance R ₂ is equal to the induced voltage at which the total magnetic flux Φ ₀ (hereinafter simply referred to as magnetic flux) slips at the slip angular frequency ω _s .

That is, ω _s Φ ₀ = R ₂ i ₂ ... (1), ω _s = R ₂ /Φ ₀ i ₂ = R ₂ /M・i ₂ /i ₀ ...
...(2) However, M: Excitation inductance i ₀ : Since the excitation current is established, the magnetic flux can be controlled by setting the slip angular frequency ω _s for the command values I ₀ and I ₃ of i ₀ and i ₂ , respectively. This is to keep it constant.

ω _s = R ₂ /M・I ₃ /I ₀ ...(3) Therefore, one of the drawbacks of this method is that, in principle, the slip angular frequency ω _s cannot be set with the motor constant R ₂ /M being a constant value. Because of this setting, if the constant changes, the correspondence between the command current and the actual current of the motor will collapse, which will affect the torque characteristics. For example, the value of the secondary resistance R ₂ changes by about 20% due to temperature changes under normal usage conditions, and accordingly, the torque relative to the torque command value I ₂ also changes by about 20%.

That is, the conventional slip frequency control method essentially has the drawback that the torque coefficient changes depending on the operating temperature of the electric motor.

Next, the difficulty in the configuration is that the slip value of the induction motor's slip frequency is small, generally several percent, and less than 1% for large capacity motors. This requires speed detection means.

For this reason, the speed is generally detected as a frequency signal using a position detector or the like, and the calculation with the slip frequency signal is performed using a special frequency adder.

Therefore, in order to control the torque, a special position detector must be installed on the rotating shaft of the motor, and a special circuit for calculating the slip frequency is required, resulting in a significantly complicated configuration.

SUMMARY OF THE INVENTION Therefore, the present invention aims to provide a torque control method using a completely new control principle, which is not affected by changes in the constants of the electric motor in principle, and which requires a simple configuration.

[Means for solving problems]

That is, the present invention fundamentally reconsiders the theory of torque control of induction motors, and if the magnetic speed Φ ₀ interlinked with the secondary circuit of the induction motor is kept at a constant value, the generated torque T of the motor will be equal to the secondary current. The characteristic is that it is proportional to i ₂ , that is, T=KtΦ ₀ i ₂ ...(4) However, if we note that Kt is a constant, detect the induced voltage of the motor, and operate it at a frequency proportional to this voltage, Based on the principle that it is inevitably a torque control system, this system achieves torque control without the drawbacks of conventional methods, that is, torque control and torque response equivalent to the characteristics of a separately excited DC motor.

〔Example〕

Examples of the present invention will be described below.

Figure 1 shows an example in which the device is applied to a two-phase induction motor.
and a voltage sensing winding equivalent to the stator winding of a motor.
Vector operator VEC consisting of voltage detector VD with DW, multipliers 4 to 7, and adders 8 and 10
, current transformers 12 and 13, feedback calculators 9 and 11, and voltage converters AMPα and AMPβ.

The power converters AMPα and AMPβ are current-controlled and receive command inputs 1 and 2, that is, a torque command current I ₃ which is an equivalent secondary current command (meaning a command for a secondary current converted to the primary side) and excitation. Command current
It is designed to flow current to the primary winding of the motor IM according to I ₀ , and the two-phase sine wave generator OSC
generates a two-phase sine wave with a frequency proportional to the input voltage. The motor is configured such that the induced voltage of the motor detected by the voltage detector VD becomes the input voltage of the two-phase sine wave generator OSC.

Now, the primary current commands I ₁ α and I ₁ β of the electric motor are generated from the two-phase signals of the excitation command current I ₀ and the torque command current I ₃ by the vector calculator VEC based on the following equation.

I ₁ α＝I ₀ cosω _0t −I ₃ sinω _0t ……(5) I ₁ β＝I ₀ sinω _0t +I ₃ cosω _0t ……(6) where ω _s : Angular frequency of exciting current And these primary currents Currents iα and iβ as shown by the following equations corresponding to the commands are applied to the power converter.
It is supplied to the primary winding of the motor IM via AMPα and AMPβ.

iα=i ₀ cosω _0t −i ₃ sinω _0t ……(7) _iβ =i ₀ sinω _0t +i ₃ cosω _0t ……(8) where i ₀ : Motor current for I 0 i ₃ : Motor current i ₀ for I ₃ , i ₃ is the spatial current vector, and the primary current iα,
iβ is a current that rotates the spatial current vectors i ₀ and i ₃ at an angular velocity ω _s when i ₀ and i ₃ are each given a constant amplitude.

That is, the method of applying the primary current according to the present invention consists in creating primary currents i ₀ and _{i 3} that respectively create rotating magnetic fields in response to command currents I ₀ and I ₃ that can be set independently.
Moreover, consideration is given so that i ₃ has a spatial phase difference of π/2 with respect to i ₀ .

Next, constant magnetic flux control will be described.

Strictly speaking, the magnetic flux referred to here is the total magnetic flux interlinked with the secondary circuit of the motor, and the induced voltage is the induced voltage corresponding to this magnetic flux viewed from the stator side.

Figure 2 shows the equivalent circuit of an induction motor, where Φ ₀ is the magnetic flux and e ₀ is the induced voltage.

The induced voltage e ₀ is the voltage detection winding DW and voltage detector VD
(The detection method will be explained later)
Provided as the input voltage of the two-phase sine wave generator OSC. Since the two-phase sine wave generator OSC is made so that the output angular frequency is proportional to the input voltage, the output angular frequency ω _c is as follows.

ω _s = Ke ₀ (9) where K: proportionality constant On the other hand, the induced voltage e ₀ is expressed by the following equation, assuming that the actual magnetic flux is Φ ₀ ′ and the actual exciting current i ₀ ′.

e ₀ = Φ ₀ ′ω _s = Mi ₀ ′ω _s ……(10) Therefore, ω _s = KMi ₀ ′ω _s ……(11) From KMi ₀ ′=1 ……(12) Here, the electric motor If the predetermined magnetic speed of is Φ ₀ = Mi ₀ and the constant K is set as K = 1/Φ ₀ = 1/Mi ₀ (13), then the actual excitation current i ₀ ' from equation (12) will match the set excitation current i ₀ .

This is because the induced voltage e ₀ is determined only by the magnetic flux and its angular frequency, regardless of the magnitude of the primary voltage or the rotational speed of the motor. The proportionality constant K sets the magnetic flux.

Therefore, if the magnetic flux is controlled to a constant predetermined value, the torque command current and the secondary current will necessarily correspond, and the torque will be proportional to the torque command.

Next, the torque transient response when the torque command current I ₃ suddenly changes will be explained.

Now, let us assume that the motor is under no load and is operating at a synchronous angular velocity ω _n .

I ₀ =i ₀ and I ₃ =0, and the primary current is only the exciting current i ₀ . Also, the induced voltage is expressed by the following equation assuming that the motor has two poles.

e ₀ = ω _n Mi ₀ ... (14) At this time, when the torque command current I ₃ is given in a stepwise manner, the secondary current i ₂ flows correspondingly,
Since the magnetic flux is already created in the direction perpendicular to i ₂ , the torque instantly corresponds to i ₂ . Therefore, it is necessary to keep the magnetic flux constant by flowing i ₂ in response to I ₃ even as time passes.

When _i2 flows due to the input of _I3 , a voltage drop occurs as an induced voltage across the secondary resistor _R2, and the induced voltage is as follows.

e ₀ = ω _n Mi ₀ + R ₂ i ₂ = Mi ₀ (ω _n + R ₂ i ₂ /Mi ₀ ) ... (15) In the case of induced voltage, the secondary resistance voltage drop corresponds to R ₂ i ₂ and is The frequency ω _i2 is changed by ω _i2 =R ₂ i ₂ /Mi ₀ . If the induced voltage generated in the secondary circuit due to ω _i2 is e ₂ , e ₂ becomes e ₂ =ω _i2 Φ ₀ =R ₂ i ₂ (16). In other words, control is performed so that i ₂ is continuously supplied with respect to the command value I ₃ . Further, since the secondary current i ₂ flows in response to the torque command current I ₃ , no change occurs in the magnetic flux, and the value is maintained at a constant value.

In other words, it can be seen that the condition of constant magnetic flux is maintained even when the torque command suddenly changes, and the torque response also responds instantaneously.

FIG. 3 shows the relationship between torque command current _I3 and torque according to the present invention.

Next, we will discuss the means for detecting the induced voltage e ₀ .

This induced voltage e ₀ is an induced voltage based on the total magnetic flux interlinking with the secondary circuit, and the air gap induced voltage e _g generated by the normal air gap magnetic velocity is the voltage due to the leakage inductance of the secondary circuit. Only different.

However, in a steady state, e ₀
There is almost no difference between the values of and e _g . In a transient manner, a large transient voltage is generated in the leakage inductance due to a sudden change in the secondary current.

Therefore, if the air gap induced voltage e _g is detected and the transient voltage of the secondary leakage inductance is removed, it can be used as e ₀ .

FIG. 4 shows an example of an induced voltage detection circuit, in which a three-phase voltage detection winding DW is provided. The voltage detector VD is composed of a three-phase diode bridge circuit, and obtains a DC voltage _g proportional to the air gap induced voltage e _g . And on the other hand, the torque command voltage
I ₃ is differentiated by the differentiating circuit 14, multiplied by a proportional coefficient k _e corresponding to the secondary leakage inductance, and subtracted from _g by the adder 15 to obtain a DC voltage ₀ proportional to the desired induced voltage e ₀ . I can do it.

Figure 5 shows a different embodiment of the induced voltage detection circuit, in which the terminal voltage of the primary winding is detected as a DC voltage _t by the voltage detection VD, and this is added to the voltage drop due to the resistance of the primary winding and the voltage drop due to the resistance of the primary winding. and the transient voltage due to the secondary leakage inductance is compensated by the current command values I ₀ and I ₃ via the voltage compensator VDC,
This is designed to approximately obtain the desired induced voltage.

In FIG. 5, 17 and 18 are constant devices, k ₀ and k ₃ are constants, 19 is a differential circuit, and k is a proportional coefficient corresponding to the primary and secondary leakage inductances.

FIG. 6 shows another embodiment in which the present invention is applied to a three-phase current source inverter-driven induction motor, and this embodiment shows an example of speed control of the three-phase induction motor IM.

As is well known, this motor drive circuit is composed of a current controllable thyristor converter COM, a 120° conduction type thyristor inverter INV, and a three-phase induction motor IM.

The current of the thyristor converter COM is controlled by a phase shifter PS and a current controller ACR, and a current according to the command is given to the inverter INV. Furthermore, DCL
is a DC reactor and CT is a DC detector.

Inverter INV also includes ring counter RC,
The gate circuit GC outputs a square wave current with a conduction width of 120°.

Then, according to the method of the present invention described above, the induced voltage of the motor is detected by the voltage detector VD and provided to the frequency generator FC.

The relationship between this voltage and frequency is set so that the motor has a predetermined voltage/frequency.

In addition, in order to orthogonalize the primary current characteristics of the motor corresponding to the excitation command current I ₀ and the torque command current I ₃ described in the basic principle, a phase compensator is used in this embodiment.
I am using a PC. This is because the change in the torque command current _I3 is taken as a change in frequency, and the output power of the phase compensator PC is also added to the input of the frequency generator FC.

On the other hand, speed control is performed using the setting voltage REO of the speed setting device REO and the speed detection voltage e _sp from the speed generator TG.
The deviation Δe _s from the above creates a torque command voltage I ₃ via the speed controller SCR, and this command value I ₃ is given to the motor predetermined excitation command current I ₀ and the effective value calculator RMS, so that I ₁ = √ ₀ ² + ₃ ² , and is given as the primary current command of the motor IM.

As described above, the present invention elucidates the principle of torque generation in an induction motor powered by a current-controlled converter, and provides a control method for maintaining a constant magnetic flux for torque control. This takes advantage of the fact that it is proportional only to the product of magnetic flux and frequency, regardless of current.

〔Effect of the invention〕

The present invention is not subject to changes in machine constants, unlike torque control using slip frequency control.
In addition, the structure is extremely simple and does not require a special detector unlike the conventional one, and is practically suitable as a torque control method for this type of induction motor.

[Brief explanation of the drawing]

FIG. 1 is an electrical schematic diagram of an embodiment of the present invention, and FIG.
The figure is an equivalent circuit diagram of an induction motor, Figure 3 is a diagram showing the relationship between torque command current and torque in an embodiment of the present invention, Figure 4 is a detailed diagram of an induced voltage detection circuit, and Figure 5 is a diagram showing the relationship between torque command current and torque in an embodiment of the present invention.
The figures are detailed diagrams of different induced voltage detection circuits, and FIG. 6 is a schematic electrical diagram of a different embodiment of the present invention. IM...Induction motor, DW...Voltage detection winding,
VD...Voltage detector, VEC...Vector calculator,
AMPα and AMPβ……power converter, OSC……2
Phase sine wave generator, FC... Frequency generator, 1 and 2... Command input, 4 to 7... Multiplier, 8 and 10
... Adder, 9 and 10 ... Feedback calculator, 12 and 13 ... Current transformer.

Claims (1)

[Scope of Claims] 1. In an induction motor that is supplied with power from a current control type converter, the primary current of the motor is divided into an excitation current command component and a torque current command component that can be set independently. At the same time, only the magnitude of the induced voltage of the motor that has compensated for the transient voltage due to the leakage inductance is detected, and the detected voltage is converted into a two-phase sine wave generator proportional to the magnitude of the induced voltage. A torque control method for an induction motor, comprising converting the primary current into a frequency and controlling the primary current according to the frequency. 2 The compensation of transient voltage due to leakage inductance is proportional to the air gap induced voltage e _g from DC voltage ₀ to k _e
Claim 1, characterized in that d/dtI ₃ (where _ke is a proportional coefficient corresponding to secondary leakage inductance, and I ₃ is torque command voltage) is subtracted.
Torque control method for an induction motor as described in . _3. Compensation for transient _voltage due to _leakage inductance is proportional to _the terminal _{voltage of the} primary winding.
2. The torque control method for an induction motor according to claim 1, wherein k is a proportional coefficient corresponding to primary and secondary leakage inductances.