JPS6131655B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to an equalizer in a receiver for orthogonal multiplexed signals, and in particular, a plurality of baseband PAM signals (pulse amplitude modulation signals) are modulated with carriers having different frequencies or phases, The present invention relates to an equalizer in a receiver for orthogonally multiplexed signals, in which a signal obtained by adding all these modulated signals becomes an orthogonally multiplexed QAM signal (quadrature amplitude modulated signal).

The transmitting and receiving method of orthogonally multiplexed QAM signals is described in detail in Patent Application No. 68523 of 1978. That is, on the transmitting side, as shown in FIG. 1, an orthogonal multiplexed signal is generated in the process described below.

In FIG. 1, input terminals 1 ₁ to 1 _2N have 2N data sets {a(1) _k }, {a(2) _k }, ..., {a ^(2N) _k }
(However, k=
-∝,...,-1,0,1,...+∝} Pulse amplitude modulated signal α ₁
(t), α ₂ (t), ... α _2N (t) are applied. The N pulse amplitude modulated signals α ₁ (t), ... α _N (t) applied to the input terminals 1 ₁ , 1 ₂ , ... 1 _N have mutual frequency response characteristics expressed by 3 ₁ , 3 ₂ , ... 3N, respectively. are equal and two are continuously connected, their transfer characteristics are input to N baseband filters that satisfy the Nyquist condition. One input end 1 _N+
The other N pulse amplitude modulation signals α _{N+1 (} t), ..., α _2N (t) added to 1 , 1 N+2 , ...1 _2N are 2 _N+1 , 2 _{N+2 ,} respectively _. ,...2 After receiving a delay of T/2 seconds through N delay elements represented by _2N , other N baseband filters 3 _N+ having the same frequency response characteristics as the N baseband filters are applied. ₁ ,
3 _N+2 ,...3 Input to _2N . Furthermore, the output β of the N baseband filters 3 ₁ , 3 ₂ ,...3 _N
1 (t), β ₂ (t)..., β _N ( _t ) are _respectively
It is modulated by carriers expressed as cosω ₁ t, sinω ₂ t, cosω ₃ t, ...sinω _N t. (However, ω _l −ω _l-1 = 2
π/T, and here N is assumed to be an even number for convenience of explanation) Further, the outputs β _N+1 (t) of the other N baseband filters 3 _N+1 , 3 _N+2 , ...3 _2N , β _N+2
(t), ..., β _2N (t) are each other N modulators 4
sinω ₁ t, cos at _N+1 , 4 _N+2 ,…, 4 _2N respectively
It is modulated by carriers expressed as ω ₂ t, sin ω ₃ t, ..., cosω _N t. 2N modulators 4 ₁ , 4 ₂ ,
..., 4 _2N are all added together by an adder 5, and then sent from an output end 6 to a transmission line.

As shown in FIG. 2, the receiving side restores the original data by performing inverse transformation to the transmitting side. That is, in FIG. 2, an orthogonal multiplexed signal transmitted via a transmission path is applied to the input end 11,
This orthogonal multiplexed signal is transmitted to 2N demodulators 12 ₁ , 12
₂ ,...12 _2N , cosω ₁ t, sinω ₂ t, cos
ω ₃ t,…sinω _N t, sinω ₁ t, cosω ₃ t,…, cosω _N
It is demodulated by t. 2N demodulators 12 ₁ , 1
The outputs of 2 ₂ ,...12 _2N each have the same frequency response characteristics as the transmitting baseband filter.
2N baseband filters 13 ₁ , 13 ₂ ,
...13 Input to _2N . Out of the outputs of these 2N baseband filters 13 ₁ , 13 ₂ , ... 13 _2N , the outputs of 13 ₁ , 13 ₂ , ... 13 _N are 14
After receiving a delay of T seconds through delay elements denoted by ₁ , 14 ₂ , . . . 14 _N , the output terminals 15 ₁ , .
15 ₂ ,...15 _N , and the outputs of other baseband filters 13 _N+1 , 13 _N+2 ,...13 _2N are delay elements represented by 14 _N+1 , 14 _N+2 ,...14 _2N are applied to the respective outputs 15 _N+1 , 15 _N+2 , . . . 15 _2N after a delay of T/2 seconds. In this way, the output ends 15 ₁ , 15 ₂ ,...1
Signals γ ₁ (t), γ ₂ (t), ..., obtained at 5 _2N
If we sample γ _2N (t) at an appropriate time, we get
A set of 2N original data {a(1) _k }, {a(2) _k }, ... {a
^(2N) _k } is restored.

The above explanation assumes that there is no distortion in the transmission path, but
Generally, since a transmission path always has some kind of distortion, it is necessary to perform equalization corresponding to this distortion on the receiving side to remove the distortion.

As an equalizer used for this purpose, a band-pass equalizer inserted between the input terminal 11 and the signal branch point 16 in FIG. 2 can be considered. however,
Such a band-pass equalizer needs to equalize the entire signal transmission band, so the band-pass filter used as the band-pass equalizer has a large fractional band, making it difficult to improve equalization accuracy. That's funny. Furthermore,
Generally, when a bandpass type transmission line is used, distortion increases at both ends of the band, and distortion is often relatively small near the center of the band. For such a transmission path, a bandpass type equalizer that equalizes the entire band may become a very inefficient equalizer.

SUMMARY OF THE INVENTION In view of the above-mentioned drawbacks of the band type equalizer, an object of the present invention is to provide a new equalizer in a receiver for orthogonal multiplexed signals, which enables a baseband equalizer.

FIG. 3 is an equivalent transmission line model of the above-mentioned orthogonal multiplex signal transmission system. In FIG. 3, for example, if k is an odd number, 101 and 102 are respectively (N
+k-1)th input end 1 _N+k-1 and (k+
1) is the input terminal corresponding to the input terminal 1 _k-1 , and 141 and 142 are the input terminals corresponding to (N+
k-1)th output end 15 _N+k-1 and (k-
1) An output terminal corresponding to the 15th output terminal 15k _-1 , and 201 and 202 are the k-th input terminal 1k and the (N+k)th input terminal 1 _{N+k in} FIG. 1, respectively.
241 and 242 are the input terminals corresponding to the k-th output terminals 15k and (N+
k)th output terminal 15 _N+k , and 301 and 302 are the output terminals corresponding to (N+k) in FIG.
+1)th input terminal 1 _N+k+1 and (k+1)th input terminal 1 _k+1 , which corresponds to 34
1 and 342 are output ends corresponding to the (N+k+1)th output end 15 _N+k+1 and the (k+1)th output end 15 _k+1 in FIG. 2, respectively;
114, 123, 204, 213, 224, 30
3, 314, and 323 are all time delay circuits that provide a time delay of T/2 seconds; 104, 113, and 12;
4,203,214,223,304,313,
324 are time advance circuits all giving a time advance of T/2 seconds, 105-108 and 115-118 and 125-128 and 205-208 and 215-218 and 225-228 and 30
5-308 and 315-318 and 325-
328 is an equivalent baseband transmission line, and 10
9,110,119,120,129,130,
131, 132, 209, 210, 219, 22
0,229,230,231,232,309,
310, 319, 320, 329, 330, 33
1 and 332 are adder circuits that add a plurality of input signals according to the polarity codes + and - in the figure. 20 again
The dashed lines indicated by are the output terminals 241 and 242.
The portions that contribute to the output signal pair obtained in the figure are shown.

Let G(ω) be the transfer function of the baseband filter represented by ₃₁ in Fig. 1, and let Q(ω) be the transfer function of the transmission path through which the orthogonal multiplexed signal is transmitted.
Then, the transfer function of the equivalent baseband transmission path represented by 125 and 128 in FIG _.
The transfer function of the equivalent baseband transmission path represented by (ω), 126 and 127 is S ₂ (ω), and the transfer function of the equivalent baseband transmission path represented by 215 and 218 is P ₁ (ω), 216 and 217. The transfer function of the equivalent baseband transmission path expressed as P ₂
(ω), the transfer function of the equivalent baseband transmission path represented by 315 and 318 is R ₁ (ω), and the transfer function of the equivalent baseband transmission path represented by 316 and 317 is R ₂ (ω), then P ₁ (ω)=1/2・G ² (ω) [Q(ω+ω _k ) e ^-j 〓+Q(ω−ω _k )e ^j 〓] ...(1) P ₂ (ω)=j/2・G ² (ω) [Q (ω + ω _k ) e ^-j 〓-Q (ω - ω _k ) e ^j 〓] ...(2) R ₁ (ω) = 1/2・G (ω) [G (ω −ω ₀ )Q (ω+ω _k )e ^-j 〓+G(ω+ω ₀ ) Q(ω−ω _k )e ^j 〓] ...(3) R ₂ (ω)=j/2・G(ω)[G (ω−ω ₀ )Q (ω+ω _k )e ^-j 〓−G(ω+ω ₀ ) Q(ω−ω _k )e ^j 〓〕 ……(4) S ₁ (ω)=1/2・G(ω ) [G(ω+ω ₀ )Q (ω+ω _k )e ^-j 〓+G(ω-ω ₀ ) Q(ω-ω _k )e ^j 〓] ...(5) S ₂ (ω)=j/2・G (ω) [G(ω+ω ₀ )Q (ω+ω _k )e ^-j 〓-G(ω-ω ₀ ) Q(ω-ω _k )e ^j 〓] ...(6). However, ω ₀ =2π/T, θ is a phase shift between demodulated carriers cosω _k t and sin ω _k t, and j represents an imaginary unit.

The equalizer in the orthogonal multiplex signal receiver according to the present invention performs baseband equalization on the output signals obtained at the output terminals 241 and 242 in FIG.

4a and 4b show an embodiment of an equalizer in an orthogonal multiplex signal receiver according to the present invention. Note that in FIGS. 4a and 4b, these two types are used depending on the channels to be equalized, as will be described later.

First, in FIG. 4a, 251 and 252
are input terminals connected to output terminals 241 and 242 in FIG. 3, respectively, 261 and 262 are output terminals, and 270, 71, and 272 are all T/2
A T/2 delay circuit 273 that provides a second delay is a T delay circuit that provides a T second delay, 274 is a first equalization circuit,
275 is a second equalization circuit, 276 is the second equalization circuit, a third equalization circuit having the same frequency response characteristics as 275, and 277 is the first equalization circuit 27.
A fourth equalization circuit 278 having the same frequency response characteristic as 4 is a subtraction circuit that subtracts the output of the second equalization circuit 275 from the output of the first equalization circuit 274; It represents an adder circuit that adds the output of the equalization circuit 276 and the output of the fourth equalization circuit 277. The signal applied to the input terminal 251 is delayed for a total of T seconds through T/2 delay circuits 270 and 271, and then equalized by the first equalization circuit 274 and output to the first subtracter 278. becomes the input. The signal applied to the other input terminal 252 is delayed for a total of 3T/2 seconds via a T delay circuit 273 and a T/2 delay circuit 272, and then is equalized by a second equalization circuit 275 indicated by 275. This becomes the second input of subtractor 278. This becomes the second input of subtractor 278. The subtractor 278 subtracts the second input from the first input and outputs the output terminal 261.
The first equalized signal x _k (t) is outputted to. Also,
The signal applied to the input terminal 252 is transmitted to the T delay circuit 27.
3 and then branches to the fourth equalization circuit 277.
and becomes the first input of the adder 279. The signal applied to the other 251 is the T/2 delay circuit 27
After passing through 0, it is equalized by a third equalization circuit 276 and becomes the second input of an adder 279. Adder 279
adds the first input and the second input and outputs a second equalized signal x _k+N (t) at the output terminal 262. The transfer function of the first equalization circuit and the fourth equalization circuit is C ₁ (ω), and the transfer function of the second equalization circuit and the third equalization circuit C ₂ (ω) is the transmission line distortion. Corresponding to , it is roughly given by the following equation.

C ₁ (ω)=Q(ω+ω _k )e ^−j 〓+Q(ω−ω _k )e ^j 〓/2Q(ω+ω _k )・Q(ω−ω _k ) ……(7) C ₂ (ω)= −j[Q(ω+ _ωk )e ^−j 〓−Q(ω− _ωk )e ^j 〓]/2Q(ω+ _ωk )・Q(ω− _ωk )…
...(8) Input terminals 251, 252 of the equalizer in the orthogonal multiple signal receiver according to the present invention shown in FIG. 4a
Output terminal 2 of the equivalent transmission model shown in Figure 3.
41, 242, the input terminal 20 in FIG.
Transfer function from 1 to the output end 261 of Fig. 4a
H ₁ (ω) is expressed as follows using equations (1) to (8). (However, consider excluding the linear delay of T seconds) Similarly, from the input terminal 202 in FIG. 3 to the input terminal 202 in FIG.
The transfer function H ₂ (ω) to the output end 262 of becomes. Here, if the time response of the transfer coefficient H ₁ (ω) is h ₁ (t), then as is well known,
t ₁ (t) is given as the inverse Fourier transform of H ₁ (ω). That is, h ₁ (t)=1/2π∫ ^∞ _−∞ H ₁ (ω)e ^j 〓 ^t dω Therefore, the sample value h ₁ of h ₁ (t) at time nT
(nT) is expressed as h ₁ (nT)=1/2π∫ ^∞ _−∞ H ₁ (ω)e ^j 〓 ^nT dω, but if we focus on the periodicity of the function e ^j 〓 ^nT with respect to ω, the above equation can be It is transformed as follows.

However, in the modification of the above equation, k→−k, ω→ω−
A substitution of 2kπ/T is performed. Therefore, if the inside of [ ] in equation (9) is identically T, then Therefore, h ₁ (t) becomes 1 only at a desired sampling time and becomes 0 at other sampling times, satisfying the so-called Nyquist condition.

Here, a root Nyquist filter is considered as G(ω). That is, G ² (ω) obtained by squaring this is The following shall be satisfied. This allows the time response
As mentioned above, h ₁ (t) becomes 1 only at the desired sampling time and becomes 0 at other sampling times, which indicates an ideal time response without intersymbol interference.
Typically, the band of the baseband filter G(ω) is limited to |ω|<2π/T. in this case
The condition of equation (10) is eventually equivalent to satisfying G ² (ω)+G ² (ω−2π/T)=T in the range |ω|<2π/T. Also, the transfer function H ₃ (ω) for the interference signal from the input end 201 in FIG. 3 to the output end 262 in FIG. 4a, and the interference signal from the input end 202 in FIG. 3 to the output end 261 in FIG. 4a. The transfer function H ₄ (ω) for the equation becomes zero as shown in the following equation.

Furthermore, if the transfer functions for the interference signal from the input terminal 101 in Fig. 3 to the output terminals 261 and 262 in Fig. 4 a are respectively H ₅ (ω) and H ₆ (ω), then It can be seen that by appropriate sampling operations at intervals of T seconds, both H ₅ (ω) and H ₆ (ω) give zero-time responses. In other words, the time responses of H ₅ (ω) and H ₆ (ω) are
If h ₅ (t), h ₆ (t), then the same as in equation (9) However, in these formulas, the above H ₅
(ω) and H ₆ (ω), we get Now the root Nyquist filter G(ω) is |ω
| Considering the imperial limit to <2π/T, k of the above series at 0<ω2π/T
Only the terms =0, 1 will contribute, So, in the end, the sample value h ₅ of h ₅ (t), h ₆ (t)
(nT) and h ₆ (nT) are all zero. That is, inter-channel interference becomes zero at any sampling time. Similarly, the transfer function for other interference signals is calculated from the input terminal 102 in FIG. 3 to a in FIG. 4.
The transfer functions up to the output terminals 261 and 262 of
H ₇ (ω), H ₈ (ω), input terminal 301 in Fig. 3
Let the transfer functions from the input terminal 302 in FIG. 3 to the output terminals 261, 262 in FIG. 4a be H ₉ (ω) and H ₁₀ (ω), respectively, and
If the transfer functions up to 2 are H ₁₁ (ω) and H ₁₂ (ω), respectively, It is expressed as Therefore, by using the equalizer in the orthogonal multiplex signal receiver according to the present invention as shown in FIG. By appropriately sampling the output signals obtained at the output terminals 261 and 262, the k-th and (k+N)-th theta sequences can be accurately restored, respectively.

In the above explanation, the case where k is an odd number has been described, but when k is an even number, the input terminals 101 and 102 in FIG.
1) correspond to the (N+k-1)th input terminal 1 _k-1 and the ( _{N+k-1)th input terminal 1 N+k-1} , and the output terminals 141 and 142 in FIG.
1) th output end 15 _K-1 and (N+k-1)
The input terminals 201 and 202 in FIG. 3 correspond to the _(N+k-1) th output terminal 15 in FIG.
+k))-th input terminal 1 _N+k and k-th input terminal 1 _k , and the output terminals 241 and 242 in FIG. 3 respectively correspond to the (N+k)-th output terminal 15 _N+k and Corresponding to the k-th output end 15k,
Input terminals 301 and 302 in FIG. 3 correspond to the (k+1)th input terminal 1 _k+1 and (N+k+1)th input terminal 1 _N+k+1 in FIG. 1, respectively, and the output terminals in FIG. 341 and 342 correspond to the (k+1)th output terminal 15 _k+1 and the (N+k+1)th output terminal 15 _N+k+1 in FIG. 2, respectively;
104, 113, 124, 203, 21 in Figure 3
4,223,304,313,324 are all T/
time delay circuit providing a 2 second time delay, 103;
114, 123, 204, 213, 224, 30
3, 314, and 323 are all time advance circuits that provide a time advance of T/2 seconds. Correspondingly, FIG. 4b shows a modification of the equalizer configuration in the orthogonal multiplex signal receiver according to the present invention shown in FIG. 4a. However, in FIG. 4, the input terminals 451 and 452 are the output terminals 241 and 2 in FIG. 3, respectively.
42, 470, 471, and 472 are all T/2 delay circuits, 473 is a T delay circuit, and 474, 475, 476, and 477 are each a fourth delay circuit.
The equalization circuits 274, 275, 276, and 277 in Figure A are the same, and 478 and 479 are the same as 278 and 277, respectively.
These are a subtraction circuit and an addition circuit corresponding to 79.

Further, the equalizer in the orthogonal multiplex signal receiver according to the present invention shown in FIGS. 4a and 4b is generally expressed as shown in FIG. 5, excluding the linear delay of T seconds. That is, in the general conceptual diagram of the equalizer in the orthogonal multiplex signal receiver according to the present invention shown in FIG.
1,552 is the input end 561,562 is the output end, 5
74 is a first equalization circuit, 576 is a third equalization circuit, 577 is a fourth equalization circuit, 578 is a subtraction circuit, 579 is an addition circuit, 571 and 57
2 is a T/2 time delay circuit and a T/2 time lead circuit, or a T/2 time lead circuit and a T/2 time delay circuit, depending on the channel to be equalized.

As described above, by using the equalizer in the orthogonal multiplex signal receiver according to the present invention, baseband equalization is possible for each channel, and with a simple configuration, the desired signal can be Equalization and equalization for interchannel interference signals are achieved simultaneously.

[Brief explanation of the drawing]

FIG. 1 is a block diagram showing a transmitting device for orthogonal multiplexed signals, FIG. 2 is a block diagram showing a receiving device corresponding to the transmitting device in FIG. 1, and FIG. 3 is a block diagram showing a transmission path from the transmitting device in FIG. A block diagram showing an equivalent transmission line model of a transmission system in which the orthogonal multiplexed signal sent to the receiver is received by the receiving device shown in FIG. FIG. 5 is a block diagram showing an embodiment of an equalizer in a signal receiver. FIG. 5 is a general conceptual diagram of an equalizer in an orthogonal multiplex signal receiver according to the present invention. In the figure, 1 ₁ , 1 ₂ ... 1 _2N are 2N input terminals, 2 _N+1 , 2 _N+2 , ... 2 _2N are N T/2 second delay elements, 3 ₁ , 3 ₂ , ... 3 _2N is 2N baseband filters, 4 ₁ , 4 ₂ ,...4 _2N is 2N modulators, 5 is adder, 6 is output end, 11 is input end, 1
2 _N+1 , 12 _N+2 ,...12 _2N is 2N demodulators 13
₁ , 13 ₂ , ... 13 _2N are 2N baseband filters, 14 ₁ , 14 ₂ , ... 14 _N are N T-second delay elements, 14 _N+1 , 14 _N+2 , ... 14 _2N are N T/2 second delay elements, 15 ₁ , 15 ₂ ,...15 _2N are
2N output terminals, 103, 104, 113, 11
4,123,124,203,204,213,
214, 223, 224, 303, 304, 31
3, 314, 323, 324, etc. are T/2 second time delay circuits or T/2 second time advance circuits,
105-108, 125-128, 205-20
8,215~218,225~228,305~
308, 315 to 318, 325 to 328, etc. are all equivalent baseband transmission lines, 270, 271, 2
72, 470, 471, 472 are all T/2 delay circuits, 273, 473 are T delay circuits, 274, 4
74 is a first equalization circuit, 275, 475 is a second equalization circuit, 276, 476 is a third equalization circuit, 2
77,477 is the fourth equalization circuit, 571,572
are respectively a T/2 time delay circuit and a T/2 time lead circuit or a T/2 time lead circuit and a T/2 time delay circuit, 574 is a first equalization circuit, 575 is a second equalization circuit, and 576 is a The third equalization circuit 577 is a fourth equalization circuit.

Claims (1)

[Claims] 1 2N synchronized with each other with a clock period of T seconds
pulse amplitude modulated signals α ₁ (t), α ₂ (t)
...Among α _2N (t), N pulse amplitude modulation signals α ₁ (t), α ₂ (t)…α _N (t) are passed through N baseband filters each having the same frequency response characteristic. output β ₁ (t), β ₂ (t)...β _N
(t) and other N pulse amplitude modulated signals α _N+1
(t), α _N+2 (t)...α _2N (t) are each T/2
After a delay of seconds, the output β is passed through another N baseband filters, each having the same frequency response characteristics as the N baseband filters.
Using _N+1 (t), β _N+2 (t), ...β _2N (t), the k-th output β _k (t) (where k=1, 2, ...
N) and (k+N)th output β _k+N (t), when k is an odd number, the in-phase carrier, e.g.
cos ω _k t and orthogonal carrier e.g. sin ω _k t
When k is an even number, by multiplying by sinω _k t and cos ω _k t and then adding the square calculation outputs (however, ω _k −ω _k-1 = 2
π/T) to obtain the k-th orthogonal amplitude modulation signal J _k (t), and thus obtain the N orthogonal amplitude modulation signals J ₁
(t), J ₂ (t), ...J _N (t) are all added up, and then the received signal received via the transmission path is inversely transformed with the transmitting side. Transmission pulse amplitude modulation signals α ₁ (t), α ₂ (t),...
In an orthogonal multiplex signal receiver that obtains received pulse amplitude modulated signals _{γ 1 (} t), γ ₂ (t), ... γ _2N (t) corresponding to α 2N (t), k=(k=1, 2 ，
. . . for each k of N) is an odd number, from the output obtained by passing the k-th received pulse amplitude modulation signal γ _k (t) through the first equalization circuit _C1, (k+
N)th received pulse amplitude modulation signal γ _k+N (t)
is delayed by T/2 seconds relative to γ _k (t), and then the k-th equalized pulse amplitude modulated signal x _k is obtained by subtracting the output obtained through the second equalization circuit C ₂ (t), and the (k+N)th received pulse amplitude modulated signal γ _k+N (t) is converted to a fourth equalizer circuit C ₄ having the same frequency response characteristic as the first equalizer circuit C ₁ . and the k-th received pulse amplitude modulation signal γ _k (t) as γ _k+N (t)
By adding the output passed through the second equalization circuit C ₂ and the third equalization circuit C ₃ having the same frequency response characteristics after advancing T/2 seconds relative to (k+N) Equalized pulse amplitude modulation signal x _k+
N ( _t ), and when _k is an even number, k γ _{k +N}
By advancing T/2 seconds relative to (t) and subtracting the output obtained through the second equalization circuit _C2 , the (k+N)th equalized pulse amplitude modulation signal x _k+N (t), and the k-th received pulse amplitude modulated signal γ _k (t) is input to the fourth equalization circuit C ₄
After delaying the (k+N)th received pulse amplitude modulated signal γ _k+N (t) by T/2 seconds relative to γ _k (t), By adding the output obtained through the equalization circuit _C3 , the k-th equalized pulse amplitude modulation signal x _k
An equalizer for an orthogonal multiplex signal receiver, characterized in that it obtains (t) and equalizes all transmission path distortions.