JPS6243624B2 - - Google Patents
Info
- Publication number
- JPS6243624B2 JPS6243624B2 JP56093615A JP9361581A JPS6243624B2 JP S6243624 B2 JPS6243624 B2 JP S6243624B2 JP 56093615 A JP56093615 A JP 56093615A JP 9361581 A JP9361581 A JP 9361581A JP S6243624 B2 JPS6243624 B2 JP S6243624B2
- Authority
- JP
- Japan
- Prior art keywords
- signal
- demodulation circuit
- level
- subcarrier
- composite
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Expired
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Classifications
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04H—BROADCAST COMMUNICATION
- H04H40/00—Arrangements specially adapted for receiving broadcast information
- H04H40/18—Arrangements characterised by circuits or components specially adapted for receiving
- H04H40/27—Arrangements characterised by circuits or components specially adapted for receiving specially adapted for broadcast systems covered by groups H04H20/53 - H04H20/95
- H04H40/36—Arrangements characterised by circuits or components specially adapted for receiving specially adapted for broadcast systems covered by groups H04H20/53 - H04H20/95 specially adapted for stereophonic broadcast receiving
- H04H40/45—Arrangements characterised by circuits or components specially adapted for receiving specially adapted for broadcast systems covered by groups H04H20/53 - H04H20/95 specially adapted for stereophonic broadcast receiving for FM stereophonic broadcast systems receiving
- H04H40/72—Arrangements characterised by circuits or components specially adapted for receiving specially adapted for broadcast systems covered by groups H04H20/53 - H04H20/95 specially adapted for stereophonic broadcast receiving for FM stereophonic broadcast systems receiving for noise suppression
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
- H04B1/06—Receivers
- H04B1/16—Circuits
- H04B1/1646—Circuits adapted for the reception of stereophonic signals
Landscapes
- Engineering & Computer Science (AREA)
- Signal Processing (AREA)
- Computer Networks & Wireless Communication (AREA)
- Stereo-Broadcasting Methods (AREA)
Description
【発明の詳細な説明】
本発明はステレオ復調回路に関し、特に左右チ
ヤンネル信号の分離度であるいわゆるセパレーシ
ヨン調整機能を有するステレオ復調回路に関する
ものである。DETAILED DESCRIPTION OF THE INVENTION The present invention relates to a stereo demodulation circuit, and more particularly to a stereo demodulation circuit having a so-called separation adjustment function, which is the degree of separation between left and right channel signals.
FMステレオ信号におけるコンポジツト信号Ｖ
（ｔ）は、
Ｖ（ｔ）＝Ｌ（ｔ）＋Ｒ（ｔ）＋
｛Ｌ（ｔ）−Ｒ（ｔ）｝cosωsｔ …(1)
で示される。ここに、Ｌ（ｔ），Ｒ（ｔ）は左右
チヤンネル信号、ωsはサブキヤリヤ信号の角周
波数である。当該コンポジツト信号Ｖ（ｔ）のう
ちメイン信号Ｌ＋Ｒは、50Hz〜15KHzの帯域で
あり、サブ信号Ｌ−ＲはDSB（ダブルサイドバン
ド）方式により23〜53KHzの帯域にある。従つ
て、受信機のIF（中間周波）段におけるIFフイ
ルタの特性により周波数成分の高いサブ信号のエ
ネルギが減少する。一般に、電界強度に応じて
IFフイルタの特性を広狭に切換制御して受信感
度を制御することが行われるが、そのためにサブ
信号エネルギがIFフイルタの特性制御に応じて
大きく変化することになり、
Ｖ（ｔ）＝Ｌ（ｔ）＋Ｒ（ｔ）＋
ｋ｛Ｌ（ｔ）−Ｒ（ｔ）｝cosωsｔ …(2)
となる。尚、ｋは０＜ｋ≦１なる定数である。よ
つて、メイン信号とサブ信号との検波出力レベル
比が変化してセパレーシヨンの悪化を招来してい
る。 Composite signal V in FM stereo signal
(t) is expressed as V(t)=L(t)+R(t)+ {L(t)−R(t)} cosω s t (1). Here, L(t) and R(t) are the left and right channel signals, and ω s is the angular frequency of the subcarrier signal. Of the composite signal V(t), the main signal L+R has a band of 50 Hz to 15 KHz, and the sub signal L-R has a band of 23 to 53 KHz due to the DSB (double side band) system. Therefore, the energy of sub-signals with high frequency components is reduced due to the characteristics of the IF filter in the IF (intermediate frequency) stage of the receiver. Generally, depending on the electric field strength
The receiving sensitivity is controlled by switching the characteristics of the IF filter widely and narrowly, but this causes the sub-signal energy to change greatly depending on the control of the characteristics of the IF filter, and V(t)=L( t)+R(t)+k{L(t)-R(t)} cosωst ...(2). Note that k is a constant satisfying 0<k≦1. Therefore, the detection output level ratio between the main signal and the sub signal changes, resulting in deterioration of separation.
従つて、セパレーシヨンを最良とするには、サ
ブ信号Ｌ−Ｒを１／ｋ倍する必要があるが、実際
には第１図に示す如き方法が採られている。図に
おいては、(2)式で示されるFM検波出力であるコ
ンポジツト信号と38KHzのサブキヤリヤ信号の
正及び逆相信号とを乗算器１及び２にてそれぞれ
乗算し、各乗算出力を加算器３及び４にてそれぞ
れコンポジツト信号中のメイン信号と加算して、
左右チヤンネル信号を分離導出するステレオ復調
方式であり、セパレーシヨン調整のためにコンポ
ジツト信号中のメイン信号の加算時に、夫々可変
減衰器５及び６によりIFフイルタの特性切換に
応じてレベル制御をなすようにしている。 Therefore, in order to achieve the best separation, it is necessary to multiply the sub-signal LR by 1/k, but in practice a method as shown in FIG. 1 is adopted. In the figure, the composite signal that is the FM detection output shown by equation (2) is multiplied by the positive and negative phase signals of the 38KHz subcarrier signal in multipliers 1 and 2, respectively, and the outputs of each multiplication are multiplied by adder 3 and In step 4, add each signal to the main signal in the composite signal,
This is a stereo demodulation method that separates and derives the left and right channel signals, and when adding the main signal in the composite signal for separation adjustment, the level is controlled by variable attenuators 5 and 6, respectively, according to the switching of the characteristics of the IF filter. I have to.
かゝる構成において、(2)式で示されるコンポジ
ツト信号Ｖ（ｔ）とサブキヤリヤ信号の正逆信号
±cosωsｔとの乗算出力V1（ｔ），V2（ｔ）は、
メイン信号をＭ（ｔ）、サブ信号をＳ（ｔ）とし
て整理すれば、それぞれ次式となる。 In such a configuration, the multiplication outputs V 1 (t), V 2 (t) of the composite signal V (t) shown by equation ( 2 ) and the positive/inverse signal ±cosω s t of the subcarrier signal are as follows:
If the main signal is organized as M(t) and the sub signal as S(t), the following equations are obtained.
V1（ｔ）＝ｋ／２Ｓ（ｔ）＋Ｍ（ｔ）cosωsｔ
＋ｋ／２Ｓ（ｔ）cos2ωsｔ …(3)
V2（ｔ）＝−ｋ／２Ｓ（ｔ）−Ｍ（ｔ）cosωsｔ
−ｋ／２Ｓ（ｔ）cos2ωsｔ …(4)
従つて、減衰器５，６においてコンポジツト信
号Ｖ（ｔ）をk/2倍して加算器３，４にて(3)、(4)
式で示される乗算出力とそれぞれ加算すれば、各
加算器３，４の出力における低周波成分（オーデ
イオ成分）ＶA1（ｔ），ＶA2は、
ＶA1（ｔ）＝ｋ／２Ｓ（ｔ）＋ｋ／２Ｍ（ｔ）＝kL（
ｔ）…
(5)
ＶA2（ｔ）＝−ｋ／２Ｓ（ｔ）＋ｋ／２Ｍ（ｔ）＝kR
（ｔ）
…(6)
と表わされ、左右チヤンネル信号がセパレーシヨ
ン最良状態にて分離されることになる。 V 1 (t)=k/2S(t)+M(t) cosω s t +k/2S(t) cos2ω s t …(3) V 2 (t)=−k/2S(t)−M(t) cosω s t −k/2S(t) cos2ω s t (4) Therefore, the composite signal V(t) is multiplied by k/2 in the attenuators 5 and 6, and the adder 3 and 4 (3) (Four)
When added to the multiplication output shown by the formula, the low frequency components (audio components) V A1 (t) and V A2 at the output of each adder 3 and 4 are as follows: V A1 (t) = k/2S (t) +k/2M(t)=kL(
t)… (5) V A2 (t)=-k/2S(t)+k/2M(t)=kR
(t) ...(6) The left and right channel signals are separated in the best separation state.
しかしながら、かゝる方法ではメイン信号成分
Ｍ（ｔ）のレベルをｋの変化により減衰させてい
るために、IF帯域幅の切換制御によつてステレ
オ復調出力のレベルが変化するという欠点があ
る。 However, in this method, since the level of the main signal component M(t) is attenuated by changing k, there is a drawback that the level of the stereo demodulated output changes depending on the switching control of the IF bandwidth.
本発明の目的はセパレーシヨン調整を良好に維
持しつつステレオ復調出力レベルの変化を招来す
ることのないステレオ復調回路を提供することで
ある。 SUMMARY OF THE INVENTION An object of the present invention is to provide a stereo demodulation circuit that maintains separation adjustment well and does not cause changes in the stereo demodulation output level.
本発明によるステレオ復調回路は、FM検波信
号であるコンポジツト信号成分を有する信号とサ
ブキヤリヤ信号との乗算をなすことにより左右チ
ヤンネル信号をそれぞれ分離して復調するように
したステレオ復調回路を対象としており、その特
徴とするところは、サブキヤリヤ信号のレベルを
制御してコンポジツト信号成分を有する信号との
乗算をなすようにしたことにある。 The stereo demodulation circuit according to the present invention is intended for a stereo demodulation circuit that separates and demodulates left and right channel signals by multiplying a signal having a composite signal component, which is an FM detection signal, by a subcarrier signal. Its feature is that the level of the subcarrier signal is controlled to perform multiplication with a signal having a composite signal component.
以下に本発明を図面により説明する。 The present invention will be explained below with reference to the drawings.
第２図は本発明の実施例を示す回路ブロツク図
であり、第１図と同等部分は同一符号により示さ
れている。図において、38KHzのサブキヤリヤ
信号cosωsｔはVCA（電圧制御可変利得アン
プ）等のレベル制御器７に入力されてレベル制御
され、正逆相信号である±Acosωsｔとなる。
こゝにＡは定数である。これら正逆信号と(2)式で
示されるコンポジツト信号Ｖ（ｔ）とが各乗算器
１，２において乗算される。従つて、各乗算出力
V1（ｔ），V2（ｔ）はそれぞれ次式で示される。 FIG. 2 is a circuit block diagram showing an embodiment of the present invention, and parts equivalent to those in FIG. 1 are designated by the same reference numerals. In the figure, a 38 KHz subcarrier signal cosω s t is input to a level controller 7 such as a VCA (voltage controlled variable gain amplifier), where the level is controlled, and becomes ±A cos ω s t, which is a positive and negative phase signal.
Here, A is a constant. These forward and reverse signals are multiplied by the composite signal V(t) shown by equation (2) in each multiplier 1, 2. Therefore, each multiplication output
V 1 (t) and V 2 (t) are each expressed by the following equations.
V1（ｔ）＝Ａｋ／２Ｓ（ｔ）＋AM（ｔ）cosωsｔ
＋Ａｋ／２Ｓ（ｔ）cos2ωsｔ…(7)
V2（ｔ）＝−Ａｋ／２Ｓ（ｔ）−AM（ｔ）cosωsｔ
−Ａｋ／２Ｓ（ｔ）cos2ωsｔ…(8)
そして、各加算器３，４の他の加算入力である減
衰器５，６を固定減衰器としてその減衰量をＢな
る一定値とすれば、各加算器３，４の出力におけ
る低周波成分（オーデイオ成分）ＶA1（ｔ），ＶA
２（ｔ）は、
ＶA1（ｔ）＝Ａｋ／２Ｓ（ｔ）＋BM（ｔ）
＝（Ａｋ／２＋Ｂ）Ｌ（ｔ）＋（Ｂ−Ａｋ／２）Ｒ
（ｔ）
…(9)
ＶA2（ｔ）＝−Ａｋ／２Ｓ（ｔ）＋BM（ｔ）
＝（Ａｋ／２＋Ｂ）Ｒ（ｔ）＋（Ｂ−Ａｋ／２）Ｌ
（ｔ）
…(10)
となる。従つて、Ａ＝2B／ｋなる値に設定すれ
ば、
ＶA1（ｔ）＝2BL（ｔ） …(11)
ＶA2（ｔ）＝2BR（ｔ） …(12)
となる。こゝで、減衰器５，６の減衰量Ｂは一定
値であるから左右チヤンネル出力は何等レベル変
化を生じないことになつて好都合である。 V 1 (t)=Ak/2S(t)+AM(t) cosω s t +Ak/2S(t) cos2ω s t...(7) V 2 (t)=-Ak/2S(t)-AM(t) cosω s t −Ak/2S(t) cos2ω s t…(8) Then, the attenuators 5 and 6, which are the other addition inputs of each adder 3 and 4, are fixed attenuators, and the amount of attenuation is set to a constant value B. Then, the low frequency components (audio components) V A1 (t), V A at the output of each adder 3 and 4 are
2 (t) is V A1 (t)=Ak/2S(t)+BM(t) =(Ak/2+B)L(t)+(B-Ak/2)R
(t) ...(9) V A2 (t)=-Ak/2S(t)+BM(t) =(Ak/2+B)R(t)+(B-Ak/2)L
(t) ...(10) becomes. Therefore, if A=2B/k is set, V A1 (t)=2BL(t)...(11) V A2 (t)=2BR(t)...(12). Here, since the amount of attenuation B of the attenuators 5 and 6 is a constant value, there is no change in the level of the left and right channel outputs, which is advantageous.
従つて、IF段の周波数帯域の切換に応じてレ
ベル制御器７の利得（Ａ＝２Ｂ／ｋ）を制御すれば、
復調出力レベルを一定に維持しつつセパレーシヨ
ンが最良に調整されるものである。 Therefore, if the gain (A=2B/k) of the level controller 7 is controlled according to the switching of the frequency band of the IF stage, the separation can be adjusted optimally while maintaining the demodulated output level constant. be.
第３図は第２図の回路の具体例を示す図であ
り、第２図と同等部分は同一符号により示されて
いる。乗算器１はアナログスイツチ３４，３５及
び抵抗２５〜２８より成り、乗算器２はアナログ
スイツチ３６，３７及び抵抗２９〜３２より成つ
ている。そしてアナログスイツチ３４〜３７のオ
ンオフ制御のために、FM検波信号であるコンポ
ジツト信号成分を有するパルス列信号の正逆相出
力が用いられており、このパルス列信号として
は、例えば周知のパルスカウント方式による検波
出力を用いることができる。この方式による検波
出力には(2)式で示されるコンポジツト信号成分の
周波数スペクトラムを有している。 FIG. 3 is a diagram showing a specific example of the circuit shown in FIG. 2, and parts equivalent to those in FIG. 2 are designated by the same reference numerals. Multiplier 1 consists of analog switches 34, 35 and resistors 25-28, and multiplier 2 consists of analog switches 36, 37 and resistors 29-32. For on/off control of the analog switches 34 to 37, positive and negative phase outputs of a pulse train signal having a composite signal component, which is an FM detection signal, are used. The output can be used. The detected output from this method has a frequency spectrum of a composite signal component shown by equation (2).
一方、アナログスイツチ３４〜３７によりオン
オフされるべき信号としては38KHzのサブキヤ
リヤ信号のレベル制御器７の正逆相出力が用いら
れる。レベル制御器７は、オペアンプ１５と、帰
還抵抗１６，１７及び１９を有しており、スイツ
チ１８により帰還抵抗１７，１９の接続を制御し
て利得をコントロールするようになつている。こ
の出力がオペアンプ２２と抵抗２０，２１よりな
るインバータによつて逆相とされている。 On the other hand, the positive and negative phase outputs of the level controller 7 of the 38 KHz subcarrier signal are used as signals to be turned on and off by the analog switches 34 to 37. The level controller 7 has an operational amplifier 15 and feedback resistors 16, 17, and 19, and a switch 18 controls the connection of the feedback resistors 17, 19 to control the gain. This output is made to have an opposite phase by an inverter made up of an operational amplifier 22 and resistors 20 and 21.
かゝる構成において、正相サブキヤリヤ信号を
逆相コンポジツト信号をもつてスイツチ３４によ
りスイツチングし、また逆相サブキヤリヤ信号を
正相コンポジツト信号をもつてスイツチ３５によ
りスイツチング制御することにより、乗算器１の
出力にサブ信号Ｓ（ｔ）＝Ｌ（ｔ）−Ｒ（ｔ）の信
号成分が出力される。 In such a configuration, by switching the positive phase subcarrier signal with the negative phase composite signal by the switch 34, and by controlling the switching of the negative phase subcarrier signal with the positive phase composite signal by the switch 35, the multiplier 1 is switched. The signal component of the sub-signal S(t)=L(t)-R(t) is output as the output.
一方、逆相サブキヤリヤ信号を逆相コンポジツ
ト信号をもつてスイツチ３６によりスイツチング
し、また正相サブキヤリヤ信号を正相コンポジツ
ト信号をもつてスイツチ３７によりスイツチング
して、乗算器２の出力に逆相サブ信号−Ｓ（ｔ）
＝Ｒ（ｔ）−Ｌ（ｔ）の信号成分が得られる。こ
れら各乗算出力をオペアンプ３９，４１及び抵抗
３８，４０より成る乗算器３，４に入力し、減衰
抵抗５，６を介したコンポジツト信号のメイン信
号成分と夫々加算するものである。こうすること
により、第２図を用いて説明した原理と同一作用
をもつてセパレーシヨンが最良となりかつ出力レ
ベルが一定の復調回路として動作するものであ
る。尚、スイツチ１８がIF段の帯域幅切換と同
時にオンオフ制御される。 On the other hand, the negative phase subcarrier signal is switched with the negative phase composite signal by the switch 36, and the positive phase subcarrier signal is switched with the positive phase composite signal by the switch 37, so that the negative phase subcarrier signal is sent to the output of the multiplier 2. −S(t)
A signal component of =R(t)-L(t) is obtained. These multiplication outputs are input to multipliers 3 and 4 comprising operational amplifiers 39 and 41 and resistors 38 and 40, and are added to the main signal components of the composite signal via attenuation resistors 5 and 6, respectively. By doing so, the demodulation circuit operates as a demodulation circuit that has the same effect as the principle explained using FIG. 2, provides the best separation, and has a constant output level. Note that the switch 18 is controlled on and off simultaneously with switching the bandwidth of the IF stage.
叙上の如く、本発明によればセパレーシヨン調
整によつて何等復調出力レベルが変化しないので
特性の良好なステレオ復調回路が得られる。 As described above, according to the present invention, a stereo demodulation circuit with good characteristics can be obtained because the demodulation output level does not change in any way due to separation adjustment.
第１図は従来のステレオ復調回路のブロツク
図、第２図は本発明の実施例の回路ブロツク図、
第３図は第２図の回路の具体例を示す図。
主要部分の符号の説明 １，２…乗算器、３，
４…加算器、５，６…減衰器、７…レベル制御
器。
FIG. 1 is a block diagram of a conventional stereo demodulation circuit, FIG. 2 is a circuit block diagram of an embodiment of the present invention,
FIG. 3 is a diagram showing a specific example of the circuit shown in FIG. 2. Explanation of symbols of main parts 1, 2... Multiplier, 3,
4... Adder, 5, 6... Attenuator, 7... Level controller.
Claims (1)
有する信号とサブキヤリヤ信号との乗算をなすこ
とにより左右チヤンネル信号をそれぞれ分離して
復調するようにしたステレオ復調回路であつて、
前記サブキヤリヤ信号のレベルを制御するレベル
制御手段を有し、前記レベル制御手段の出力信号
と前記コンポジツト信号成分を有する信号との乗
算をなすようにしたことを特徴とするステレオ復
調回路。 ２ 前記コンポジツト信号成分を有する信号は、
該コンポジツト信号周波数スペクトラムを含むパ
ルス列信号であり、前記パルス列信号により前記
サブキヤリヤ信号をスイツチングして乗算をなす
ようにしたことを特徴とする特許請求の範囲第１
項記載のステレオ復調回路。 ３ 前記レベル制御手段は中間周波信号段の通過
帯域幅の変化に応じて前記レベルを変化させるよ
う構成されていることを特徴とする特許請求の範
囲第１項又は第２項記載のステレオ復調回路。[Scope of Claims] 1. A stereo demodulation circuit configured to separate and demodulate left and right channel signals by multiplying a signal having a composite signal component, which is an FM detection signal, by a subcarrier signal,
A stereo demodulation circuit comprising: level control means for controlling the level of the subcarrier signal; and multiplication of the output signal of the level control means and the signal having the composite signal component. 2. The signal having the composite signal component is
Claim 1: The composite signal is a pulse train signal including the frequency spectrum of the composite signal, and the subcarrier signal is multiplied by switching the subcarrier signal with the pulse train signal.
Stereo demodulation circuit described in section. 3. The stereo demodulation circuit according to claim 1 or 2, wherein the level control means is configured to change the level according to a change in the passband width of the intermediate frequency signal stage. .
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP56093615A JPS57208743A (en) | 1981-06-17 | 1981-06-17 | Stereo demodulation circuit |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP56093615A JPS57208743A (en) | 1981-06-17 | 1981-06-17 | Stereo demodulation circuit |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| JPS57208743A JPS57208743A (en) | 1982-12-21 |
| JPS6243624B2 true JPS6243624B2 (en) | 1987-09-16 |
Family
ID=14087225
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| JP56093615A Granted JPS57208743A (en) | 1981-06-17 | 1981-06-17 | Stereo demodulation circuit |
Country Status (1)
| Country | Link |
|---|---|
| JP (1) | JPS57208743A (en) |
Families Citing this family (1)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JPS604054U (en) * | 1983-06-20 | 1985-01-12 | 株式会社ケンウッド | stereo demodulator |
Family Cites Families (2)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JPS5586235A (en) * | 1978-12-25 | 1980-06-28 | Toshiba Corp | Fm stereo signal demodulator |
| JPS5853531B2 (en) * | 1979-02-13 | 1983-11-30 | 三洋電機株式会社 | stereo demodulation circuit |
-
1981
- 1981-06-17 JP JP56093615A patent/JPS57208743A/en active Granted
Also Published As
| Publication number | Publication date |
|---|---|
| JPS57208743A (en) | 1982-12-21 |
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