JPH0158754B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION [Technical Field of the Invention] The present invention relates to a power supply circuit used in a switching type power amplifier or the like in which a voltage waveform applied to a switching element during interruption becomes an arc of resonance.

[Technical background of the invention and its problems]

Conventional switching type power amplifiers switch the input power supply voltage directly into a rectangular waveform, as seen in the chopper type and full bridge type.
One is to change the conduction time width to change the power transmitted to the load, and the other is to configure a switching circuit that does not directly switch the input power supply voltage by adding an external circuit, as seen in class D power amplifiers. It is known that the voltage or flowing current is made to be an arc of resonance, and the amount of power supplied to the load is controlled using a separate converter, modulator, or the like.

As for the switching loss in the chopper method, the transition loss of the switch element increases as the switching frequency increases, and the limit of the switching frequency is determined by the switching speed of the switch element. On the other hand, in a class D power amplifier, even if the switching frequency increases, the transition loss of the switching element does not increase much, and highly efficient switching can be performed. However, on the other hand, since the resonance conditions of the voltage waveform applied to the switch element are determined by the external circuit, the power supplied to the load does not change over a wide range even if the conduction width or switching period of the switch element is changed. Therefore, to control the power in this class D power amplifier, it is necessary to install another variable voltage device before and after it.
It had the disadvantage of complicating the configuration.

[Purpose of the invention]

An object of the present invention is to provide a power supply circuit that can control the power supplied to a load over a wide range while maintaining a resonant waveform in a switching power amplifier or the like that utilizes a voltage resonant waveform.

[Summary of the invention]

The invention is for energy storage of flyback pulses by resonance, separate from the main switch element for voltage resonant switching. By connecting an auxiliary switch element in series with either or both of the first inductance and the second inductance for current smoothing of forward pulses, and controlling the conduction width of this auxiliary switch, the load can be controlled. It is designed to control the power supply.

〔Effect of the invention〕

According to this invention, the power supplied to the load can be controlled by controlling the auxiliary switch element, and the conduction width and switching period of the main switch element can be basically constant, so that the resonance condition can be maintained correctly.
Therefore, it is possible to control the power supplied to the load over a wide range while taking advantage of the high efficiency of the power supply circuit using voltage resonance type switching.

[Embodiments of the invention]

FIG. 1 is a circuit diagram showing an embodiment of the present invention. In the figure, 1 is an input DC power supply, one end of a parallel circuit consisting of a main switch element 2 and a resonant capacitor 3 is connected to one end of this power supply 1, and the other end of this parallel circuit and the other end of the power supply 1 are connected. A first inductor 5 is connected to the inductor 5 through a first auxiliary switch 4, and a second auxiliary switch 6 and a second
A rectifying and smoothing circuit consisting of a rectifying diode bridge 8 and a smoothing capacitor 9 is connected through an inductor 7 in series. The output of this rectifying and smoothing circuit is then supplied to the load 10. Note that damper diodes 11, 12, and 13 are connected in antiparallel to each of the switch elements 2, 4, and 6, respectively.

Here, some assumptions will be made to explain the operation of FIG. It is assumed that the main switch element 2 opens and closes with a constant conduction width and period, and the circuit shown in FIG. 1 has reached a steady state. In other words, the terminal voltage of capacitor 3 υ _C
As shown in FIG. 3b, the potential is zero at t=0 just before the main switch element 2 becomes conductive, and the terminal voltage of the smoothing capacitor 9, that is, the output voltage E out becomes constant, and equivalently, as shown in FIG. Battery 14 as shown
(hereinafter referred to as an output equivalent battery).

First, for the sake of simplicity, the operation will be explained under the condition that the auxiliary switch elements 4 and 6 are always in a conductive state.

The equivalent circuit during the period when the main switch element 2 is closed is as shown in FIG. 2a. In this case, the first inductor 5 has a proportional coefficient E in /L ₁ , which is the ratio of the voltage E in of the power supply 1 to the inductance value L ₁ of the inductor 5, as shown in the curve 21 of FIG. A proportionally increasing current i ₁ flows.
On the other hand, the second inductor 7 has a curve 2 in FIG. 3c.
As shown in Figure 3, the voltage E in of power supply 1 and the output voltage E
difference from out and inductance value of inductor 7
A current i ₂ flows which increases proportionally with time t using the ratio (E in - E out)/L ₂ as _a proportional coefficient. The sum of the currents flowing through the inductors 5 and 7 flows through the main switch element 2, and continues for the conduction period T on of the main switch element 2.

Next, when the main switch element 2 opens, the inductor 7
An inertial current flows during the period t ₁ , and this period t ₁
The equivalent circuit of is shown in FIG. 2b. In other words, the current flowing through the inductor 5 first flows through the power source 1.
The resonance capacitor 3 is charged via. On the other hand, the inertial current flowing through the inductor 7 also charges the capacitor 3 via the output equivalent battery 14 and the power source 1. Therefore, the terminal voltage of capacitor 3 υ _C
As shown in the τ ₁ period of the curve 22 in FIG. 3B, the capacitance value of the capacitor 3 and the parallel combined inductance value of the inductors 5 and 7 resonate and rise.

Next, when the inertial current of the inductor 7 stops flowing, the terminal voltage of the inductor 7 becomes zero, so a voltage with the opposite polarity is applied to the diode bridge 8, and the diode voltage with respect to the output equivalent battery 14 is applied. The combination changes as shown in Figure 2c.

Period t=(T on+τ ₁ ) of the state shown in Fig. 2 c
~(T−τ ₃ ) (where T is the switching period of the main switch element 2) is a period when the output equivalent battery 14 is
As shown in FIG. d, power is supplied and at the same time the capacitor 3 is charged. and capacitor 3
After charging is finished and the terminal voltage ν _C reaches the maximum voltage, the charge stored in the capacitor 3 becomes the current 1
furthermore, an inductor 5 and an output equivalent battery 1
4 and flows into the inductor 7. As this resonance process progresses further, eventually the capacitor 3
The terminal voltage υ _C has a polarity that makes the damper diode 11 connected antiparallel to the main switch element 2 conductive. As a result, the equivalent circuit is again shown in Figure 2 a.
Return to During the period τ ₃ during which the diode 11 is conductive, the inertial current remaining in the inductors 5 and 7 returns to the power supply 1.

The above is the basic operation of the circuit shown in Fig. 1 when power control is not performed.
We will explain what kind of problems occur in the operation of the circuit when the conduction width of the main switch element 2 is simply changed in order to control the power supplied to the circuit.

First, when the load 10 becomes lighter, the conduction width T on of the main switch element 2 is narrowed to T on' to reduce the output power, and the current i flowing through the inductor 5 becomes
As shown by curve 31 in Figure a, it increases linearly up to T on ', and then oscillates under a resonance condition determined by the actual circuit numbers. On the other hand, the terminal voltage of capacitor 3 υ _C
As shown by the curve 32 in FIG. 3b, after the main switch element 2 is disconnected, it oscillates around the voltage E in of the power source 1 under a resonance condition determined by the circuit constant.
Since the resonance energy at the time t=T on' when vibration starts is small, its amplitude is small, and therefore υ _C
cannot return to zero potential. When the next switching period of the main switch element 2 starts with the terminal voltage υ _C of the capacitor 3 remaining at a finite value in this way, the energy of (1/2Cυ _C ² )×T is instantly transferred to the main switch element 2. The main switch element 2
The temperature rise during switching becomes large. If this exceeds the allowable value, the main switch element 2 will be damaged. This inconvenience occurs when the main switch element 2 is conductive.
This occurs because there is little magnetic energy stored in the inductor 5 during on'. So inductor 5
must have sufficient energy stored in it so that the resonant arc of the terminal voltage υ _C of the capacitor 3 can return to zero potential before the start of the next switching cycle. However, for the main switching element 2, the conduction width T is always on, the switching operation is performed in a constant state where the period T ratio is around 50%, and the power supplied to the load 10 is controlled by changing the conduction width of the second auxiliary switch element 7, while the capacitor 3 If the first auxiliary switch element 4 is opened and closed with a conduction width T on'' that is sufficient to maintain the resonance condition of the terminal voltage υ _C of the capacitor 4, the terminal voltage υ _C of the capacitor 4 will always maintain the voltage resonance condition. The power supplied to 10 can be changed freely.

Further, according to this embodiment, by providing the auxiliary switch elements 4 and 6, the following secondary effects can be obtained. That is, first, if the circuit of FIG. 1 is temporarily stopped by opening the auxiliary switch elements 4 and 6, and then restarted, the capacitor 3 is connected to the power supply 1 when the main switch element 2 starts conducting. Since it is not charged to the voltage E in, it is possible to prevent excessive current from flowing into the main switch element 2 at the time of starting the circuit.

Furthermore, since the terminal voltage of the smoothing capacitor 9 is zero when the second normal circuit as shown in FIG. 1 is turned on, the slope of the curve 24 in FIG. An excessive current flows, and if a switch element that can withstand this excessive current is to be prepared, it would need to have a withstand current nearly ten times that of a steady load. However, in the circuit shown in FIG. 1, power energy can be transmitted to the load 10 with just the flyback pulse, so when this circuit is turned on, only the first auxiliary switch 4 is opened and closed, and the magnetic field stored in the first inductor 5 is If energy is supplied to the load 10 and the second auxiliary switch 7 is opened and closed after the terminal voltage of the smoothing capacitor 9 has risen sufficiently, the main switch element 2 will be lower than the normal state. Since excessive current will not flow, a device with a low withstand current will suffice.

In Fig. 3, assuming that there are no auxiliary switch elements 4 and 6 in Fig. 1 and this part is always connected, the main switch element 2 opens and closes at a certain period after opening and closing for a certain period of time. When the inductors 5 and 7 are at rest, the residual energy of the inductors 5 and 7 oscillates through the power supply 1 and is attenuated by the load 10 and the loss of the circuit. It is possible to eliminate such damped vibrations. That is, by opening and closing the main switch element 2 and the auxiliary switch elements 4 and 6 in synchronization, the residual energy accumulated in the inductors 5 and 7 is transferred to the diode bridge 8.
Since the power is returned to the power supply 1 via the , it is possible to improve efficiency.

In addition, in Fig. 1, especially when dealing with high power, if a gate transistor or a gate turn-off thyristor that can turn off the current by itself is used as the main switch element 2, the auxiliary switches 4 and 6 can be commutated even if a thyristor or the like is simply used. No circuit required.

This invention is not limited to the above-mentioned embodiments,
For example, even if the first auxiliary switch 4, the first inductor 5, and the diode 12 are omitted, the power supplied to the load can be controlled by the second auxiliary switch 6 while maintaining the voltage resonance condition. Conversely, when the first inductor 5 is present, it is also possible to omit the second auxiliary switch 6 and control the power supplied to the load using only the first auxiliary switch 4. Further, when a transformer is inserted before the load, the second inductor 7 may be replaced by its leakage inductance. Note that the leakage inductance is the inductance seen from the primary side of the modulator by shooting the secondary side of the modulator.

Furthermore, the circuit according to the present invention can be applied to both switching type power amplifiers and switching type power supplies; in the case of the former, for example, the auxiliary switching element is switched with a pulse width modulated wave modulated by the input signal, and in the case of the latter In this case, switching may be performed in response to fluctuations in the voltage applied to the load.

[Brief explanation of drawings]

FIG. 1 is a circuit configuration diagram showing an embodiment of the present invention, FIG. 2 is an equivalent circuit diagram in each operation mode, and FIG. 3 is an operation waveform diagram of each part of FIG. 1. 1...Input DC power supply, 2...Main switch element, 3...
Resonance capacitor, 4... First auxiliary switch element, 5... First inductor, 6... Second auxiliary switch element, 7... Second inductor, 8... Diode bridge for light flow, 9... Smoothing capacitor, 10
...Load, 11-13...damper diode.

Claims (1)

[Claims] 1. One end of a parallel circuit consisting of a main switch element that opens and closes with a predetermined conduction width and cycle and a resonant capacitor is connected to one end of an input DC power source, and the other end of this parallel circuit is connected to the other end of the input DC power source. A rectifying and smoothing circuit is connected to the other end via an inductor, and an auxiliary switch element is connected in series with the inductor in a power supply circuit that supplies power to a load via this rectifying and smoothing circuit. A power supply circuit characterized in that power supplied to a load is controlled by controlling the conduction width of an element. 2. The power supply circuit according to claim 1, wherein the inductor is constituted by a leakage inductance of a transformer. 3 Connect one end of a parallel circuit consisting of a main switch element and a resonant capacitor that open and close with a predetermined conduction width and cycle to one end of the input DC power supply, and connect the other end of the parallel circuit and the other end of the input DC power supply. In a power supply circuit which connects a first inductor and also connects a rectifying and smoothing circuit via a second inductor, and supplies power to a load via this rectifying and smoothing circuit, the first and second inductors are connected to each other. A power supply circuit characterized in that at least one auxiliary switch element is connected in series with at least one of the auxiliary switch elements, and the power supplied to the load is controlled by controlling the conduction width of the auxiliary switch element. 4. The power supply circuit according to claim 3, wherein the second inductor is constituted by a leakage inductance of a transformer.