JPH0325041B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION Field of Industrial Application The present invention relates to a filter that can be used in transmitters and receivers for radios, televisions, personal radios, and other communication devices in general.

Conventional configurations and their problems In recent years, the number of broadcast waves from radio and television and communication waves from communication devices has increased, and high stability and reliability are required in the performance of filters that select desired radio waves. . On the other hand, reducing the manufacturing costs of these receivers, transmitters, and communication devices is also a major issue.
In particular, drastic technological development is required for filter circuit components in the high frequency section, which is difficult to rationalize.

A conventional multi-stage double-tuned filter circuit component will be described below with reference to the drawings. Figure 1 shows a basic multi-stage tuned filter circuit, where 1 to 3 are tuning inductors, 4 to 6 are tuning capacitors, and 7
and 8 are coupling capacitors, 9 is an input terminal, and 10 is an output terminal. There is also a conventional example (not shown) in which the coupling capacitors 7 and 8 are not used, and the coupling is made using electromagnetic induction between tuning inductors. Conventionally, the components constituting this filter circuit were inductor components 11 to 13 and capacitor components 14 to 18 connected by conductors 61 and 62 as shown in FIG.

However, in the above configuration, the inductor parts and capacitor parts are larger in size than other high-frequency parts, and the height dimension in particular hinders miniaturization and thinning of the device.

The inductance of an inductor component tends to shift due to mechanical vibration, and since the ferrite core has a large temperature dependence, the inductance is unstable and the tuning frequency fluctuates greatly.

The inductor component and capacitor component exist as separate components and are connected by a conductor routing circuit, resulting in a large amount of lead inductance and stray capacitance, making the circuit operation unstable.

Since it is a collective circuit of individual parts with independent minimum unit functions, there are limits to reducing the number of parts and rationalizing manufacturing.

It had the following problems.

Purpose of the Invention The purpose of the present invention is to realize a multi-stage double-tuned filter circuit block in which an inductor part and a capacitor part are integrated, thereby making the form of the filter circuit block ultra-thin and compact, and further improving mechanical efficiency. It is stable against vibrations, has little temperature dependence of the tuning frequency, is stable at high frequencies by eliminating the negative effects of connection leads, and can streamline the manufacturing process by reducing the number of parts.

Structure of the Invention The filter of the present invention has at least three or more electrodes disposed opposite to each other via at least two or more dielectrics having the same thickness and the same dielectric constant, and the ground terminals of the respective electrodes are electrodes facing each other. By setting the opposite directions between the electrodes and setting the area where the above-mentioned respective electrodes face each other or the equivalent length of the electrodes to a required value, a plurality of different frequency tuning parts can be formed and an arbitrary frequency tuning part can be formed. It is configured to exhibit double tuning frequency selection characteristics,
As a result, one electrode acts as a distributed inductor between opposing electrodes, and this electrode and the other electrode face each other to form a distributed constant circuit with an open tip, and its equivalent length is operated. The basic idea is to realize a distributed capacitor due to the negative reactance generated by setting the length to be less than λ/4 of the frequency wavelength, and to make it act in parallel with the above-mentioned distributed inductor, and the value of this distributed capacitor is In order to arbitrarily set the value of the above-mentioned distributed inductor, the electrode pattern is formed as desired, and the opposing area between the electrodes and the equivalent length of the electrodes are simultaneously set, and a distributed capacitor having an arbitrary capacitance and the required Distributed inductors having arbitrary inductances are alternately stacked to form a plurality of tuning sections having different tuning frequencies, which act as a multi-stage double tuning filter.

DESCRIPTION OF EMBODIMENTS Hereinafter, embodiments of the present invention will be described with reference to the drawings.

FIG. 3 shows a configuration diagram of a filter in an embodiment of the present invention. In Fig. 3, a indicates a front view, b indicates a side view, and c indicates a back view (hereinafter the same applies to Figs. 4 to 8).
a, 19b are dielectric substrates, 20 to 22
are electrodes forming a distributed constant circuit to form a distributed inductor and a distributed capacitor. The ground terminals of the electrodes 20 to 22 are set so that the opposing electrodes are on opposite sides, as shown in FIG. (The same applies to Figures 4 to 8 below)
A○ side and B○ side shown in Fig. 3a and A○ shown in Fig. 3C
The electrodes 20 to 22 face each other in different patterns (the same applies to FIGS. 4 to 8 below). Here derivative 19a
and 19b have the same thickness and the same dielectric constant (the same applies in FIGS. 4 to 8 below). With the above configuration, the electrodes 20 and 21 and the dielectric 19a form one single tuning section, and the electrodes 21 and 22 and the dielectric 19b form another single tuning section. The tuning frequencies of the respective single tuning sections are different, and they are combined to form a double tuning filter with sharp selectivity. (The same applies to FIGS. 4 to 8 below. An explanation of the operation will be described later.) FIG. 4 shows a configuration diagram of a filter in another embodiment of the present invention. An electrode 24 having one bent part is connected to the dielectric substrates 23a and 23b.
26 are arranged facing each other.

FIGS. 5 and 6 show configuration diagrams of filters in other embodiments of the present invention. In FIG. 5, electrodes 28 to 30 each have a shape having a plurality of bent portions via dielectric substrates 27a and 27b.
are placed facing each other. In FIG. 6, electrodes 32 to 34 having a plurality of bent portions and forming a folded shape are arranged to face each other with dielectric substrates 31a and 31b interposed therebetween.

FIG. 7 shows a configuration diagram of a filter in an embodiment of the present invention. Dielectric substrates 35a and 35
spiral-shaped electrodes 36 to 38 via b
are placed opposite each other.

FIG. 8 shows a configuration diagram of a filter in another embodiment of the present invention. Inside the dielectric substrate 39, electrodes 40 to 42 are arranged facing each other with equal spacing between them. In addition to the shape of the electrode in the embodiment shown in FIG. 8, an electrode having a bent portion as in the embodiments shown in FIGS. 4 to 7 may be used.

The operation of the filter of this embodiment constructed as described above will be explained below, using the embodiment shown in FIG. 6 as a representative example. First, the inductor is formed by folded electrodes 32 and 34 as shown in FIG. 6a. Next, the capacitors are folded electrodes 32 and 33.
One is generated by the dielectric substrate 31a existing between the folded electrodes 33 and 34, and the other is generated by the dielectric substrate 31b existing between the folded electrodes 33 and 34. Here, the folded electrode 33 forming the capacitor is common to the above-mentioned capacitors.

Next, the operating principle of the tuner used in the filter of the present invention will be explained.

9a to 9g are equivalent circuits for explaining the operation of the tuner of the present invention. In Fig. 9a, for each transmission line having an electrical length l and with the ground terminals set in opposite directions, a voltage e
It is assumed that a signal source 272 that generates a signal is connected to the transmission line electrode 270 and supplies the signal. As a result, a traveling wave voltage e _A is excited at the open terminal at the tip of the transmission line electrode 270. On the other hand, since the transmission line electrode 271 is disposed close to the transmission line electrode 270, facing each other or in parallel, a voltage is induced by mutual induction.
Let e _B be the traveling wave voltage induced in the open terminal at the tip of the transmission line electrode 271.

Here, since the ground terminals of the transmission line electrodes 270 and 271 are set in opposite directions, the induced traveling wave voltage e _B has an opposite phase to the excited traveling wave voltage e _A. Since the tips of the transmission paths of the traveling wave voltages e _A and e _B are open, voltage standing waves are formed in the transmission path formed by the transmission path electrodes 270 and 271. Here, the voltage distribution system representing the distribution mode of voltage standing waves in the transmission line electrode 270 is K
The voltage distribution coefficient at the transmission line electrode 271 can be expressed as (1-K).

Therefore, when the potential difference V generated at any opposing portion of the transmission line electrodes 270 and 271 is determined, it can be expressed as V=Ke _A -(1-K)e _B (1). Here, if the respective transmission line electrodes 270 and 271 have the same electric peak length l, then e _B = -e _A ...(2), so the potential difference V in the first equation is V = Ke _A + (1-K) e _A = e _A ...(3). That is, a potential difference V can be generated in all parts where the transmission line electrodes 270 and 271 face each other.

Here, it is assumed that the transmission line electrodes 270 and 271 have an electrode width W (the thickness of the electrodes is thin), and are opposed to each other at a distance d with a dielectric material having a dielectric constant ε _S interposed therebetween. do. In this case, the capacitance C _O formed per unit length of the transmission path is C _O =Q/V=Q/e _A ...(4) Q=ε _O ε _S W・V/d=ε _O ε _S We _A /d...(5) Therefore, C _O =ε _O ε _S W/d...(6).

Therefore, the transmission line shown in Figure 9a can be found using the formula 6 for each unit length as shown in Figure 9b.
This becomes a transmission path including a C _O distributed capacitor 273. Furthermore, since the voltage standing wave distribution (or current standing wave distribution) in each transmission line electrode 270 and transmission line electrode 271 are in an antiphase relationship with each other as described above, this transmission line is equivalently It will operate as a balanced mode transmission line. This is equivalent to a balanced mode transmission line formed by transmission line electrodes 275 and 276 excited in a balanced mode by a balanced signal source 27 having a balanced voltage e', as shown in FIG. 9c. become.
Needless to say, its electrical length is the same as the original electrical length l shown in FIG. 9a. Furthermore, as shown in FIG. 9(d), this balanced mode transmission line has integrated distributed inductors 277 and 278 and a distributed capacitor 273 due to the distributed inductor component of the transmission line and the lumped inductor component generated by the bent shape of the transmission line, respectively. It can be equivalently expressed as a distributed constant circuit consisting of:

Next, the relationship between the formation of the distributed capacitor 273 and the electrical length l of the transmission path will be explained.
The characteristic impedance Z _O per unit length in a balanced mode transmission line as shown in Figure 10a is the 10th
It can be expressed by the equivalent circuit shown in Figure b. Its characteristic impedance Z _O is generally becomes. If the transmission path is lossless, then becomes. This assumption can be applied to many of the embodiments of the tuner of the present invention, and to simplify the explanation, the characteristic impedance Z _O
Use. The capacitance C _O in the eighth equation is the same as the capacitance C _O per unit in the transmission line found in the sixth equation. In other words, the characteristic impedance per unit length in the transmission line
Z _O is a function of the capacitance C _O , which is also a function of the permittivity ε _S of the dielectric material involved in the capacitor C _O , the width W of the transmission line electrodes and the spacing d between the respective transmission line electrodes.

As mentioned above, the characteristic impedance per unit length in the transmission line is Z _O , the electrical length is l, and the equivalent reactance X generated at the terminal of the transmission line with the end open is as follows: _O cotθ can be expressed as (9). Here, θ=2πl/λ...(10), and especially In this case, the equivalent reactance X is X≦0 (12). In other words, the equivalent reactance at the terminal of the transmission line can be the capacitive reactance. Therefore, depending on the electrical length l of the transmission path, θ becomes the
When formula 11 is satisfied, for example, a capacitor can be formed by setting the electrical length l to λ/4 or less. And the capacitance C of the capacitor that can be formed is As expressed by , any capacitance C can be realized by changing θ, that is, by setting the electrical length l of the transmission path.

FIG. 11 is a diagram illustrating the operation mode of the transmission line explained in Equations 9 to 13 above. FIG. 11 shows how the equivalent reactance X generated at the terminal changes in accordance with the change in the electrical length l of a transmission line with its tip in an open state. As is clear from Fig. 11, the electrical length l of the transmission line is λ/4 or less or λ/2 ~
It is possible to form a negative terminal reactance in cases such as at 4λ/3. In other words, a capacitor can be equivalently formed.
Further, by arbitrarily setting the electrical length l of the transmission path under conditions that generate negative terminal reactance, it is possible to realize the capacitance C to an arbitrary value.

The capacitor C thus formed can be equivalently replaced as a lumped capacitor 279 shown in FIG. 9e. The inductor formed by combining the distributed inductor component existing in the transmission path and the lumped inductor component generated by bending the transmission path can be equivalently replaced as the lumped constant inductor 280. Then, if the virtual balanced signal source 274 and the ground in each transmission line are replaced so that they are equivalent to the state shown in the original FIG. 9a, the state shown in FIG. 9f is obtained. Become. If the ground terminal is shown in common in FIG. 9f, it is clear that the lumped constant capacitor 27 will eventually be connected to the lumped constant capacitor 27 as shown in FIG.
9 and a lumped constant inductor 280, and can realize a tuner.

Although the above-described operation realizes the tuner of the present invention, the configuration of the tuner of the present invention and its operating principle are completely different from those of conventional tuners. Therefore, in order to prove that the tuner according to the present invention is completely different from conventional tuners or those having other configurations even if they use the same transmission line as the tuner of the present invention. The structure and operation of a conventional tuner or a tuner with other transmission line configurations will now be described and compared.
This clarifies the difference from the tuner according to the present invention and also clarifies the novelty of the tuner according to the present invention.

FIG. 12 shows the operation when the transmission line electrode is formed of the same material as that used in the tuner of the present invention, but the difference is that the ground terminals are set in the same direction. be. 1st
In FIG. 2a, it is assumed that a transmission line with an open end consisting of transmission line electrodes 281 and 282 is driven by a signal source 283 that generates a voltage E. As a result, a standing wave voltage e _A is excited at the open terminal at the tip of the transmission line electrode 281, and a standing wave voltage e _B is excited at the open terminal at the tip of the transmission line electrode 282, which is installed opposite or in parallel. shall be induced. Here, since the ground terminals of the transmission line electrodes 281 and 282 are set in the same direction, the standing wave voltages e _A and e _B are in phase with each other. Therefore, the respective voltage distribution coefficients at transmission line electrodes 281 and 282 have the same K. As a result, the potential difference V at any part where the transmission line electrodes face each other becomes V=Ke _A -Ke _B (14). Here, if the electrical lengths of the respective transmission line electrodes 281 and 282 are the same length, e _A = e _B ... (15), so the potential difference V in Equation 14 is V = Ke _A - Ke _A =0...(16) In other words, no potential difference occurs in any part of the transmission path. FIG. 12b shows a configuration in which the signal source 283 in FIG. 12a is replaced at the end of the transmission line, and is equivalent to installing an unbalanced signal source 284 that generates voltage e'. In this equivalent circuit, there are only parallel transmission lines with no potential difference between them. In other words, it is clear that this is equivalent to the case where only one transmission line electrode 285 exists, as shown in FIG. 12c. Then, by equivalently replacing the signal source 283 and the ground terminal with the original circuit shown in FIG. 12a, the circuit shown in FIG. 12d is obtained. In other words, only an equivalent lumped constant inductor 286 is formed, which is composed of a distributed inductor component of the transmission line and a lumped inductor component generated by the curved shape of the transmission line. As is clear from the above, since a capacitor cannot be formed in parallel with an inductor, the intended parallel resonant circuit tuner cannot be realized.
Figure 13 shows a general microstrip line formed with the same transmission line electrode as in the tuner of the present invention as one side of the transmission line electrode, but the electrode facing the transmission line electrode has a sufficiently wide ground. This shows the operation when the curved points are different. In FIG. 13a, a transmission line electrode 287 faces a sufficiently wide ground electrode 288, is driven by a signal source 289 that generates a voltage e, and a standing wave voltage e _A is excited at the open terminal at the tip of the transmission line. Let K be the voltage distribution coefficient. On the other hand, assuming that a standing wave voltage e _B having a voltage distribution coefficient W is virtually generated at the ground electrode 288, the transmission line electrode 287 and the ground electrode 28
The potential difference V at any part where 8 faces each other is expressed as: V=Ke _A −Ke _B (17). However, the standing wave electric e _B at the ground electrode 288 is uniformly at the ground potential (zero potential), and e _B =0 (18). Therefore, there is no voltage distribution coefficient at ground electrode 288 either. As a result, the potential difference V becomes V=Ke _A (19). This makes it possible to form a distributed capacitor between the transmission line electrode 287 and the ground electrode 288. However, the transmission line electrode 28
7 closely faces the ground electrode 288, the two ends of the transmission line electrode 287 are almost in a shot state due to mutual induction. Therefore, the Q performance of the inductor component in the transmission line electrode 287 is significantly degraded. That is, this microstrip line has an equivalent loss resistance 29 as shown in FIG. 13b.
A parallel resonant circuit is formed of a lumped constant inductor 291 and a lumped constant capacitor 292, each including zero. Here, the equivalent loss resistance 290 actually has a considerably large resistance value, so
Losses in the resonant circuit become very large. Therefore, as a tuner, it is only possible to realize a tuner with clearly very low Q performance. Actually, it is not suitable for practical use.

FIG. 14 shows the circuit configuration of a λ/4 resonator, which is the most commonly used circuit in the past. This shows that the tuner is completely different from the tuner of Second
In FIG. 0, the balanced mode transmission line electrodes 293 and 294 have an electrical length l set equal to λ/4 at the resonance frequency, and have shortened tips. and a balanced signal source 2 that generates a voltage e
95, each transmission line electrode is driven in a balanced mode. The ground terminal is set at the neutral point of the balanced signal source 295, and does not specifically ground any terminal in the transmission line electrode. In this case, the equivalent terminal reactance X generated at the terminal of the transmission line is as follows, where Z _O is the characteristic impedance of the transmission line: X=Z _O tanθ (20). Here, the characteristic impedance Z _O is the same as that shown in the eighth equation. In this resonator, the electrical length l of the transmission path is l=λ/4...(21), so θ=π/2...(22). Therefore, _the terminal reactance X in Equation 20 is as follows: However, when comparing the configuration of this λ/4 resonator with the configuration of the tuner of the present invention, first of all, regarding the terminal conditions of the transmission line, the tuner of the present invention is in an open state, whereas the conventional λ/4 resonator has an open state. It is clear that the /4 resonator is in a short state and therefore has a completely different configuration in terms of terminal conditions. Furthermore, regarding the setting of the electrical length l of the transmission line, in the tuner of the present invention, it is set to less than λ/4 of the tuning frequency, and in practice it is set to a very short value of about λ/16. However, in a conventional λ/4 resonator, the resonant frequency is strictly set to λ/, so the configuration may be fundamentally different in setting the electrical length l of the transmission path. it is obvious. Furthermore, due to the difference in the electrical length l of the transmission path in the configuration, the tuner of the present invention can be made smaller even if both are designed to have the same tuning frequency or resonant frequency; In the case of four resonators, it is necessary to provide a very long transmission path, which has the disadvantage of increasing the size. In order to downsize conventional λ/4 resonators, some have shortened the length of the transmission path by interposing a dielectric material with a very high permittivity, but the dielectric material with a high permittivity used for this is generally a dielectric material. The body loss tan δ is very large, and therefore the Q performance as a resonator is significantly degraded. Furthermore, the temperature dependence of the dielectric constant of a dielectric material having a high dielectric constant is generally large, and therefore, there is also the disadvantage that it is difficult to ensure the stability of the resonant frequency.

Next, in order to clarify the superiority of the performance of the tuner of the present invention, the results of an experiment comparing the performance of a conventional tuner will be shown and explained. FIG. 15 is a graph showing the experimental results of measuring the temperature dependence of the tuning frequency. FIG. 16 is a graph showing the experimental results of measuring the temperature dependence characteristics of resonance Q. In FIGS. 15 and 16, characteristic (A) is the temperature dependence of the tuner according to the present invention, and is an experimental result when an alumina ceramic material or a resin printed circuit board is used as the dielectric material. On the other hand, characteristic (B) is as shown in Figure 2.
This is the temperature-dependent characteristic of the tuner most commonly used in the past. From these experimental results, it is clear that in the tuner of the present invention, even if it is constructed using a general dielectric material, its tuning frequency is extremely stable, and furthermore, the resonance Q is high and it is stable. On the other hand, in conventional tuners, the tuning frequency is affected by the fundamental instability of the magnetic permeability μ and Q in the ferrite core that constitutes the inductor, and by changes in the inductance due to expansion and contraction of the coil portion, respectively. It was difficult to ensure the stability of the resonance Q. Thereby, other temperature compensation components or other self-stabilizing compensation circuits were added to compensate for the instability.

FIG. 17 shows an operational equivalent circuit of a single tuned circuit having two different tuning frequencies forming this filter. In FIG. 17a, 43 and 44 are transmission circuits equivalent to folded electrodes forming an inductor, and 45 are transmission circuits equivalent to folded electrodes acting together with transmission line electrodes 43 and 44 to form distributed capacitors 46 and 47. It is a transmission circuit. Here, the earth point of the transmission circuit 45 is connected to the transmission circuits 43 and 44 forming the inductor.
Since the transmission circuits 43, 44, and 45 each share a magnetic field having the same magnetic field polarity, the polarities of the currents flowing through the transmission circuits 43 and 44 are the same, On the other hand, a current having a polarity opposite to that of the current flowing through the transmission circuits 43 and 44 flows through the transmission circuit 45. Therefore, the transmission circuits 43 and 44 are inductively coupled, but the transmission circuit 44 is
and does not inductively bind to 45. Therefore, as shown in FIG. 17b, the transmission circuit 44
This is equivalent to a ground circuit 48 in which a current with a polarity opposite to that of the current flowing through the inductor circuits 3 and 45 flows, and forms distributed capacitors 51 and 52 facing the inductor transmission circuits 49 and 50. FIG. 17c shows this as a distributed constant circuit, which is formed by distributed inductors 53 and 54 and distributed capacitors 55 and 56. Figure 17d shows this as a lumped constant equivalent circuit, which forms a parallel resonant circuit of inductor 57 and capacitor 58 and a parallel resonant circuit of inductor 59 and capacitor 60, each with a different tuning frequency. The parallel resonant circuits are coupled by mutual induction to form a double-tuned multistage filter. The inductance of the inductor of this filter can be arbitrarily designed depending on the number of times the folded electrode is folded or the equivalent length of the electrode. On the other hand, the capacitance of the distributed capacitor can be arbitrarily designed by selecting the opposing areas of the opposing folded electrodes and the thickness and dielectric constant of the dielectric substrate.

Next, we will further discuss the formation of distributed capacitors in the 18th section.
This will be explained with figures. Let the transmission circuit equivalent length of the folded electrode be l, and this transmission circuit equivalent length l is λ/4 at the operating frequency wavelength considering the wavelength shortening rate 1/√ determined by the dielectric constant ε of the dielectric substrate used. Set to something shorter than the length. By arbitrarily designing the ratio of the transmission circuit equivalent length l to the λ/4 length at the operating frequency wavelength, it is possible to arbitrarily set the value of the capacitance reactance _XC . The capacitance C=1/2πf ₀ X _C is obtained by this capacitance reactance X _C and the operating frequency f ₀ . A capacitor having this capacitance C is equivalent to the capacitor 58 or 60 shown in FIG. 17d. By setting the transmission circuit equivalent length l to the required length, the opposing area between the electrodes is determined and the distributed capacitance C can be set, and the distributed inductance can also be set at the same time depending on the transmission circuit equivalent length l. can.

FIG. 19 shows an example of the frequency selection characteristics of the multi-stage double-tuned filter in each of the above embodiments, and the selection characteristics of the tuning section composed of electrodes having a relatively long transmission circuit equivalent length are relatively low. On the other hand, the selection characteristic of a tuning section composed of electrodes having a relatively short transmission circuit equivalent length has a relatively high tuning frequency. A double tuning selection characteristic can be obtained by combining the respective tuning sections and synthesizing the selection characteristics.

In the structure of each of the above embodiments, the number of electrode layers is three and the number of dielectric layers is two, but the number of layers can be set arbitrarily in alternately disposing the electrode layers and dielectric layers. It is also optional to provide a tap as an input or output terminal of the filter circuit block at a portion of the electrode placed on the outside that exhibits the required impedance. Note that the electrodes in each of the above embodiments include metal conductors,
Printed conductors or thin film conductors can be used, and alumina ceramic, plastic, Teflon®, glass, mica, etc. can be used as the dielectric substrate.

Effects of the Invention As is clear from the above explanation, the present invention is configured such that electrodes of the same shape are placed opposite each other with a thin dielectric layer interposed therebetween, so that a plurality of inductor parts and a plurality of capacitor parts can be easily connected to each other with a simple configuration. It is possible to realize an ultra-thin and compact multi-stage double-tuned filter block whose parts can be integrated.

Since the multi-stage multi-tuned filter can be made into a module, its tuning frequency is extremely stable, and in particular, deviations in the tuning frequency due to mechanical vibration can be completely eliminated.

Since each inductor and capacitor are connected in a leadless manner, there is no influence of lead inductors or stray capacitors, and therefore the filter circuit operation becomes extremely stable.

It is possible to reduce the number of parts, streamlining manufacturing and reducing costs.

Since the electrode layer and the dielectric layer can be formed by a printing process or a bonding method, it is possible to manufacture a filter having stable double-tuned frequency performance in large quantities at low cost.

It is possible to realize a filter block that is ultra-thin and compact, yet has sharp double-tuned frequency selection characteristics. This excellent effect can be obtained.

[Brief explanation of drawings]

Figure 1 is a basic multi-stage double-tuned filter circuit diagram.
Figure 2 is a perspective view of a conventional multi-stage tuned filter configuration.
3 to 8 are configuration diagrams of a filter in an embodiment of the present invention, in which a is a front view, b is a side view, c is a back view, and FIGS. 8a to 8g,
Figures 9a, b and 10 are explanatory diagrams showing the operating principle of the tuner used in the filter of the present invention, and Figure 11a
~d, Figures 12a, b, and 13 are explanatory diagrams showing the operating principle of a conventional tuner, Figures 14 and 1
Figure 5 is a characteristic diagram of the tuning frequency and resonance Q with respect to temperature changes for the present invention and the conventional tuner, and Figures 16 and 17.
18 is a diagram illustrating the operating principle of a filter in an embodiment of the present invention, and FIG. 18 is a diagram illustrating an example of frequency selection characteristics of a filter in an embodiment of the present invention. 19a, 19b, 23a, 23b, 27a, 2
7b, 31a, 31b, 35a, 35b, 39...
...Dielectric substrate, 20, 21, 22, 24, 25,
26, 28, 29, 30, 32, 33, 34, 3
6, 37, 38, 40, 42, 42... Electrode.

Claims (1)

[Claims] 1. Three electrodes are placed opposite each other through two dielectrics having the same thickness and the same dielectric constant,
In addition to setting the terminals connected to the ground of each electrode to be on opposite sides between the opposing electrodes, the area where the above-mentioned respective electrodes face or the equivalent length of the electrodes is set to the required value. Therefore, a filter is characterized in that it forms a plurality of different frequency tuning sections and exhibits arbitrary dual tuning frequency selection characteristics. 2. Claim 1 in which an electrode having at least one bending portion exhibiting an arbitrary bending angle or bending rate and an arbitrary bending direction is used.
The filter described in section. 3. The filter according to claim 1, wherein the electrode has a spiral shape. 4. According to any one of claims 1 to 3, the length of one electrode is arbitrarily set shorter than the length of the other electrode, and the electrodes are arranged facing each other or in parallel at an arbitrary part. filter.