JPH067745B2 - Current resonance type converter - Google Patents

Description

Description: <Industrial application field> The present invention relates to improvement of output control characteristics of a current resonance converter.

<Prior Art> A conventional current resonance type converter utilizes LC resonance to
It is a system that smoothes the change in the current of the switching element, and is characterized by small noise and switching loss during switching.

The output voltage control in this current resonance type converter is performed by the ratio (fs / f) of the switching frequency fs and the resonance frequency fn.
This is done by changing n). The relationship is approximately represented by the following equation.

V ₀ / Vi = fs / fn ... However, Vi is an input voltage and V ₀ is an output voltage.

Normally, the method of controlling the output voltage by changing the switching frequency is used, but when the input fluctuation is large, a large frequency change occurs, so it is difficult to design the output filter, and multiple converters are connected in parallel. When driving, there are problems such as interference. On the other hand, there is a prior art related to the same applicant who controls the output voltage by changing the resonance frequency by using a variable inductor for resonance, and has an advantage that the switching frequency can be made constant.

<Problems to be Solved by the Invention> However, in the method of controlling the output by changing the value of the resonance inductance as shown in the above-mentioned prior art, while the switching frequency can be made constant, the peak of the resonance current is generated when the input range is wide. The value becomes large. For example,
FIG. 8 shows the inductance L with respect to the input voltage fluctuation of k times.
FIG. 6 is a characteristic diagram when the output voltage is controlled by changing r. When the input voltage Vi = Vi1, Lr and Cr are determined so that the resonance frequency fn can output the required power voltage V ₀ and the characteristic impedance Zn can minimize the resonance current peak value Ip.

When the output voltage control is performed, the inductance Lr decreases (in order to increase the resonance frequency fn) as the input voltage Vi increases, and the characteristic impedance Zn.
Is decreased, the resonance current peak value Ip (= Vi1 / Z
n) is increasing. At the input voltage Vi2 (= kVi1),
The resonance frequency fn needs to be changed by k times (fn1 → kfn1), the characteristic impedance Zn needs to be changed by 1 / k times (Zn → Zn / k), and the resonance current peak value Ip needs to be changed by k ² times (Ip1 → k ² Ip1). . Therefore, when the input range k is large, a large current flows through the switching element, the stress applied to the switching element becomes large, and there is a problem that it cannot cope with a wide input range.

The present invention has been made in view of the above-described problems of the prior art, and is a current resonance type converter of a method of controlling output by changing a value of a resonance inductance, and a switching element even if an input voltage changes. It is an object of the present invention to provide a current resonance type converter which can cope with a wide input voltage range without increasing the peak value of the current flowing in the converter.

<Means for Solving the Problems> The configuration of the present invention for solving the above problems is an LC resonance circuit to which a signal related to an input voltage is applied, and a switch means, a variable inductor and a capacitor are connected in series, A switch control circuit that controls the switch means by comparing an input voltage with a first reference voltage, and a control current of the variable inductor is increased or decreased based on a difference between an output voltage based on a voltage across the capacitor and a second reference voltage. An output voltage control circuit for controlling the output voltage to change to a constant value corresponding to the second reference voltage by changing the resonance frequency of the LC resonance circuit with a control current, and the switch control circuit controls the output voltage. When the input voltage is lower than the first reference voltage, the resonance is 2
It is characterized in that the resonance is performed once or more, and the resonance is performed once when the input voltage is higher than the first reference voltage.

<Operation> According to the present invention, in the current resonance type converter of the method of controlling the output voltage in consideration of the resonance inductance, the resonance is performed twice or more when the input voltage is low, and the resonance is performed once when the input voltage is high. By doing so, it is possible to reduce the change in the peak value of the resonance current due to the input voltage fluctuation, so that the stress of the switching element is reduced and it is possible to cope with a wide input voltage range.

<Examples> Hereinafter, the present invention will be described with reference to the drawings.

FIG. 1 is a block diagram showing an embodiment of a current resonance type converter according to the present invention, which is of a half bridge circuit type. The present invention relates to a full-waveform current resonance type converter configured such that a reverse current flows through a diode or the like connected in parallel to the switching element. There is a half-wave current resonant converter configured to block with a diode or the like connected to.

In FIG. 1, Vi is an input voltage, 1 and 2 are input voltages V
A dividing capacitor for dividing i into two, 3 and 4 are input voltages Vi
A switching element such as MOSFET connected in series at both ends of
5 and 6 are parasitic diodes of MOSFETs 3 and 4, 7 and 8 are MOSF
MOSFET3, connected between the gate and source terminals of ET3,4
4 is a gate drive circuit for driving 4; 9 is a resonance variable inductor in which one end of the inductor winding is connected to the connection point of the divided capacitors 1 and 2; 10 is a connection point of the MOSFETs 3 and 4 and the inductor winding of the variable inductor 9; A transformer in which both ends of the primary winding are connected to the other end of the wire, 11 is a resonance capacitor connected to both ends of the primary winding of the transformer 10, 12, 1
3 is a rectifying diode whose anode side is connected to both ends of the secondary winding of the transformer 10, 14 is a choke coil whose one end is connected to the cathode side of the rectifying diodes 12 and 13, and 15 and 16 are choke coils 14 respectively. A filter capacitor and a load resistor connected between the other end and the middle point of the transformer 10, and 14 and 15 form an output filter. Reference numerals 17 and 18 denote voltage dividing resistors for dividing the voltage across the load resistor 16, 19 denotes a second reference voltage, and 20
Is an error amplifier for amplifying the difference between the divided output by the voltage dividing resistors 17 and 18 and the second reference voltage 19, 21 is a base driving circuit driven by the voltage across the load resistor 16, 22
Is a transistor for current control that amplifies the output of the base drive circuit 21 and drives the control winding of the variable inductor 9, and 17 to 22 form an output voltage control circuit. Further, 23 and 24 are voltage dividing resistors for dividing the input voltage Vi, 25 is a first reference voltage, 26 is a voltage dividing resistor 23, 2.
And the first reference voltage 25
An error amplifier for amplifying the difference between
6 is a gate drive control circuit that determines the number of resonances according to the magnitude of the input voltage Vi from the output of 6, and controls the switching timing of the MOSFETs 3 and 4 based on the number of resonances. ing.

The operation of the current resonance converter having such a configuration will be described below. First, the basic operation will be described with reference to the operation waveform diagram of FIG.

(I) T0 ≦ t ≦ T1 period Immediately before time T0, the two MOSFETs 3 and 4 are both off,
On the secondary side, the load current I0 flows back through the rectifying diodes 12 and 13. Gate-source voltage V at time T0
When the MOSFET 3 is turned on by gs1, the drain-source current Ids1 linearly increases through the primary winding of the transformer 10 and the variable inductor 9, which causes the rectification diode 13
Continues until turns off.

(II) Period of T1 ≦ t ≦ T2 At time T1, when the rectifying diode 13 is reverse-biased and turned off, resonance by the resonance capacitor 11 and the variable inductor 9 starts.

(III) T2 ≦ t ≦ T3 period The resonance current IL becomes negative and the parasitic diode 5 of the MOSFET 3
A current flows in the direction of regenerating electric power to the input side through.
During this period, the MOSFET 3 is turned off by the gate-source voltage Vgs1 (t = Ta), so that the MOSFET 3 can be turned off when no current is flowing (zero current switching) and no switching loss occurs. At this time, the peak value of the resonance current must satisfy the following conditions. (Conditions for zero current switching) Vi / Zn = I0 / N N: Turn ratio of the transformer 10 (IV) Period of T3 ≦ t ≦ T4 At time T3, the resonance current IL stops flowing, so the resonance capacitor 11 continues to discharge linearly until the rectifying diode 13 is turned on, and discharges at time T4. Ends.

During the period of T1 ≦ t ≦ T3, only the rectifying diode 12 is forward biased and turned on on the secondary side.

After that, when the MOSFET 4 is turned on, the same operation as above is performed, but the resonance current IL flowing through the variable inductor 9
Direction is opposite to that of MOSFET 3. In this way MOSFETs 3 and 4
While and are alternately turned on, the input energy is sent to the secondary side.

Next, output voltage control by the variable inductor will be described with reference to FIGS. 1 and 3 which are comparative diagrams of resonance waveforms corresponding to input voltage fluctuations.

As shown in FIG. 3, when operating at the input voltage Vi2 (FIG. 3 (a)), the input voltage is Vi1 (Vi2 = kVi1, k>
1) (FIG. 3 (b)), the relationship between the input / output voltage ratio and fs / fn is expressed by the following equation: V0 = (fs / fn) Vi Therefore, immediately after that, the output voltage V0 also decreases. When the output voltage V0 decreases, the negative input of the error amplifier 20 becomes the second
Becomes smaller than the reference voltage 19 and the output of the error amplifier 20 becomes positive. Since the base drive circuit 21 and the current control transistor 22 are configured so that the control current Ict becomes small when the output of the error amplifier 20 is positive, the inductance Lr of the variable inductor 9 increases. That is, the input voltage V
When the inductance of the variable inductor 9 at i1 is Lr1 and the inductance at the input voltage Vi2 is Lr2, the characteristic impedance Zn, the resonance frequency fn, and the peak value Ip of the resonance current IL are respectively expressed by the following equations.

(When the input voltage is Vi1) Ip1 = Vi1 / Zn1 ...- 1 (when the input voltage is Vi2) ^{Ip2 = Vi2 / Zn2 = k 2} · Ip1 ... -2 ^{However, Lr2 = (1 / k 2} ) Lr1, Vi2 = kVi1, k>
It is O.

Further, the output voltage before and after the fluctuation of the input voltage is given by the following equation.

(Before change) Vo2 = (fs / fn2) Vi2 ...- 1 (After change) Vo1 = (fs / fn1) Vi1 ...- 2 However, Vi2 = kVi1, fs is constant Therefore, when the input voltage decreases from Vi2 to Vi1 Since the inductance of the variable inductor is increased to Lr1, the resonance frequency drops to fn1 and becomes stable when the output voltage becomes Vo1 = 02. In this way, by controlling the variable inductor, the output voltage can be kept constant against input fluctuations.

Next, a switch control circuit (23 to 27 in FIG. 1) that reduces the resonance current peak value when the input range is doubled.
The operation will be described with reference to FIG. The input voltage Vi
The range is Vi1 ≦ Vi ≦ Vi2, Vi2 = 2Vi1. The input voltage Vi is divided by the voltage dividing resistors 23 and 24, and the error amplifier 26 amplifies the difference between the divided output and the first reference voltage 25. The output is Range of It is determined whether it is within the range.

If the range is, as shown in FIGS. 4 (a) and (b),
The gate drive control circuit 27 issues a command to the gate drive circuits 7 and 8 of the MOSFETs 3 and 4 so that resonance is performed twice in one switching cycle. on the other hand, If the range is, as shown in FIGS. 4 (c) and 4 (d),
It issues a command to resonate once.

If it is attempted to cover the entire input range with a single resonance as in the conventional case, the resonance current I becomes as shown by a (dotted line) in FIG.
p requires a 4-fold variation range. Therefore, a current of at least IoMAX + 4Ip1 flows through the MOSFET. On the other hand, according to the device of FIG. 1, the input range can be divided into two ranges, and the characteristic impedance Zn that can reduce the peak current can be designed in both ranges. ) And c, the current peak value may be reduced. That is, when the switching frequency fs, the output voltage V _0, and the input range k are determined from the above equation,
The required range of the resonance frequency fn is obtained as shown in Fig. 6a (dotted line). In FIG. 6A, when the entire input range is covered by one resonance, the input voltage Vi1 to 2Vi1 (input / output voltage ratio V ₀
/ Vi1~V ₀ / 2Vi1) resonance frequency against fn1~2f
Indicates that n1 is required. On the other hand, the above formula is expressed as follows: V ₀ / Vi = fs / fn = Tn / Ts where Tn = 1 / fn, Ts = 1 / fs The input / output ratio is the switching cycle Ts and the resonance current It is determined by the duty ratio of the period Tn. Therefore, as shown in Fig. 7, when the resonance of fn1 is performed once during the Ts period (Fig. A)
Is equivalent to the case where resonance of 2fn1 is performed twice (b), and V ₀ / Vi becomes equal (fs / fn1 = f
s / 2 / 2fn1). Servant By setting the desired input / output voltage ratio V ₀ / 2Vi 1 to V
_It is understood that ₀ / Vi1 can be obtained.

Input voltage Vi1 when resonating twice and Input voltage when resonating once And a calculation formula of the characteristic impedance Zn at 2Vi1, the resonance peak current Ip, and the peak value of the current IL flowing through the MOSFET. However, the input voltage In consideration, the characteristic impedance is designed so that the regenerative current is minimized in consideration of the on / off time of the MOSFET, the delay time of the gate drive circuit and the control circuit, and the value is set to Za.

Therefore, the value of the resonance peak current Ip becomes Ip1 to 2Ip1 in both input ranges, and can be kept in the double change range. Actually, since the second resonance is slightly attenuated compared to the first resonance, the resonance peak current Ip becomes a little larger by supplementing the input energy for the attenuation, but it is far more than the one resonance covers all the inputs. A small resonance peak current Ip will suffice.

In the above embodiment, the case where the input range is doubled has been described, but even in an arbitrary input range, the resonance peak current can be reduced by dividing the input range into two by using resonance twice.

In the above embodiment, the input range 1 (resonance 1
However, an example in which the resonance current is reduced in two ranges, that is, the input range 2 (resonance 2 times) is shown, but in general, any n number of ranges (resonance n times) should be set according to the input voltage range. Can be done.

Further, the switching element illustrated in the above embodiment is not limited to the MOSFET.

<Effects of the Invention> As described above in detail with reference to the embodiments, according to the present invention, the output voltage can be controlled to a constant value while the switching frequency remains constant, and the resonance due to the input voltage fluctuation can be achieved. Since the change in the peak value of the current can be reduced, the stress of the switching element is reduced, and a current resonance type converter that can handle a wide input voltage range can be realized.

[Brief description of drawings]

FIG. 1 is a block diagram showing an embodiment of a current resonance type converter according to the present invention, FIG. 2 is an operation waveform diagram for explaining the operation of the apparatus of FIG. 1, and FIG. 3 corresponds to input voltage fluctuation. FIG. 4 is a resonance waveform diagram showing how the resonance current peak value is reduced by the device of FIG. 1, and FIG. 5 is a relationship between the input voltage, the resonance frequency and the resonance current peak value of the device of FIG. FIG. 6 is a characteristic diagram showing the relationship between the input / output voltage ratio and the resonance frequency when the input range is double, and FIG.
The figure shows the relationship between the switching frequency and the resonance frequency in the case of double resonance, and FIG. 8 shows the conventional current resonance type converter in which the output voltage is controlled by changing the inductance Lr with respect to the input voltage fluctuation of k times. It is a characteristic diagram in a case. 9 ... Resonant variable inductor, 11 ... Resonant capacitor,
17, 18, 23, 24 ... Voltage dividing resistors, 19 ... Second reference voltage, 20, 26 ... Error amplifier, 21 ... Base drive circuit, 22 ... Current control transistor, 25 ... First reference voltage, 27 ... Gate drive control circuit, Vi ... Input voltage, V ₀ ... Output voltage.

Claims (1)

[Claims] 1. A discontinuous mode current resonance type converter in which an input DC power supply is provided with a switching element, an LC resonance circuit including a variable inductor and a capacitor, a rectification circuit and a smoothing circuit, and an output voltage is obtained from the smoothing circuit. A circuit for detecting the input DC voltage during operation and determining which range of the input voltage range the voltage is divided into, and switching the ON time of the switch element according to the determination result. Control means for changing the number of resonances in one cycle of switching according to the value of the input DC voltage, and increasing or decreasing the control current of the variable inductor from the difference between the output voltage and the reference voltage to change the resonance frequency of the LC resonance circuit. And an output voltage control circuit for controlling the output voltage to a constant value corresponding to the reference voltage by changing the output voltage. Characteristic current resonance type converter.