JPH0783610B2 - Power converter - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION Object of the Invention (Field of Industrial Application) The present invention relates to a power converter for supplying AC power of variable voltage and variable frequency to an AC motor or the like.

(Prior Art) A self-extinguishing element such as a GTO (gate turn-off thyristor) is used as a power converter that supplies variable-voltage variable-frequency AC power to a large-capacity AC motor (induction motor, synchronous motor, or linear motor). The voltage type pulse width modulation control inverter (called PWM inverter) used has come to practical use.

FIG. 9 shows a multiple pulse width conversion control inverter with an output transformer (hereinafter referred to as multiple PWM inverter) and a pulse width modulation control inverter without an output transformer (hereinafter directly connected PWM).
A power conversion device in which a power conversion device is combined with an inverter).

In the figure, C _d is a DC smoothing capacitor that serves as a DC voltage source, INV-
1 is the first inverter, TR is the output transformer, INV-2 is the second
Inverter, LOAD is a load device, CT _L is a current detector, C _L
Is a comparator, G (S) is a current control compensation circuit, R _L is a proportional amplifier, SW ₁ and SW ₂ are switch circuits, PWM ₁ and PWM ₂ are pulse width modulation control circuits, and SWC is a switch control circuit.

The first and second inverters INV-1 and INV-2 are pulse width modulation control inverters that convert a DC voltage into an AC having a variable voltage and a variable frequency, and show a case of single-phase output. First inverter
INV-1 generates the output voltage V _L1 via the output transformer TR. Further, the second inverter INV-2 directly generates the output voltage V _L2 . For load device LOAD, V _L = V _L1 + V _L2
Voltage is applied, and by adjusting the voltage V _L ,
It controls the load current I _L.

When the output frequency ₀ is zero, the voltage V _L1 is output from the transformer TR.
Can not be generated, the second inverter IN
The V-2, and controls the load current I _{L, 0} is upon reaching certain extent, to control the load current I _L by the first inverter INV-1.

The control operation will be briefly described below.

Let the minimum frequency at which the output transformer can operate be _min ,
When the output frequency ₀ is lower than _min , the switch control circuit SWC turns on the switches SW ₁ and SW ₂ to the side a.

Then, the pulse width modulation control circuit P of the first inverter INV-1
The input signal ▲ V ^* ₁ ▼ of WM ₁ is set to zero, and the input signal ▲ V ^* of the pulse width modulation control circuit PWM ₂ of the second inverter INV-2 is set ^.
₂ ▼ is given from the current control compensation circuit G (S).

First, the current detector CT _L detects the load current I _L and inputs it to the comparator C _L. With the comparator C _L , the above current detection value I _L
And its current command value ▲ I ^* _L ▼ compare, the deviation epsilon _{L =} ▲
I ^* _L ▼ the -I _L is amplified by next current control compensation circuit G (S). The output signal of G (S) is input to the pulse width modulation control circuit PWM ₂ via the switch circuit SW ₂ . The second inverter INV-2 generates a voltage V _L2 proportional to the input signal ▲ V ^* ₂ ▼ of the PWM ₂ .

When ▲ I ^* _L ▼> I _L , the deviation ε _L becomes a positive value,
▲ V ^* ₂ ▼ That is, the output voltage V of the second inverter INV-2
_L2 is increased and the load current I _L is increased so that I _L ≅ ▲ I ^* _L ▼. On the contrary, when ▲ I ^* _L ▼ <I _L , the deviation ε _L becomes a negative value, and V _L2 is decreased to decrease I _L. Again, I _L ≈ ▲ I ^* _L ▼ Controlled.

During this time, the first output voltage V ₁ of the inverter INV-1 is, PWM ₁
Input signal ▲ V ^* ₁ ▼ = 0, that is, a voltage proportional to 0, that is, controlled to be zero voltage, and the secondary voltage of the output transformer TR.
V _{L1 is} also zero.

When the output frequency ₀ becomes higher than the minimum frequency _min , the switch control circuit SWC turns on the switches SW ₁ and SW ₂ to the side b.

This time, the output signal of the current control compensation circuit G (S) is input to the pulse width modulation control circuit PWM ₁ of the _first inverter, and ▲ V is input to the pulse width modulation control circuit PWM ₂ of the second inverter.
^* ₂ ▼ ＝ ▲ I ^* _L ▼ ・_RL signal is input. Where R
_L is equivalent to the resistance of the load device LOAD, ▲ I ^* _L
▼ ・_RL is the voltage drop due to resistance. That is, ₀ >
_The output frequency of the second inverter INV-2 is _{0 at min.}
Regardless of this, the voltage drop will be borne by the resistance of the load. In addition, the load current I _L is the first inverter INV-1
Will be controlled by.

FIG. 10 shows the load terminal voltage V _L for the output frequency ₀ of the device of FIG. 9 and the input signals ▲ V ^* ₁ ▼, ▲ of the pulse width modulation control circuits PWM ₁ and PWM ₂ of the first and second inverters. V ^* ₂ ▼
Shows the relationship.

When the load current I _L is constant, the load terminal voltage V _L is represented by the output frequency ₀ as follows.

V _L = V _C + I _L · R _L + jε ₀ L _L I _L (1) V _C : Load back electromotive force (for motor) R _L : Load resistance L _L : Load inductance ε ₀ : 2π ₀ The broken line in FIG. 10 (a) shows the voltage drop due to the resistance R _L of the load.

The broken line in FIG. 10 (b) represents the input signal ▲ V ^* ₁ ▼ of the pulse modulation control circuit PWM ₁ of the first inverter INV-1. ₀ ≤
Until _min , ▲ V ^* ₁ ▼ = 0, and in the region of ₀ > _min , ▲ V ^* ₁ ▼ becomes a voltage proportional to ₀ . Also,
The solid line in the figure (b) represents the input signal ▲ V ^* ₂ ▼ of the pulse width modulation control circuit PWM ₂ of the second inverter INV-2. ₀ ≤
Up to _min, it is controlled to output the voltage _VL of the formula (1), and in the region of ₀ > _min , ▲ V ^* ₂ ▼ = ▲ I ^* _L ▼ ・_RL
= Becomes constant. However, when ▲ I ^* _L ▼ = constant.

Thus, in the device shown in FIG. 2, the second inverter INV-
2 is only required to generate the required voltage V _L up to the minimum frequency _{min at} which the output transformer TR can operate.
Compared with the inverter INV-1, the capacity is smaller. Also, the first inverter INV-1 is an output transformer TR.
Since the voltage V _L1 is generated via the inverter, it is possible to easily perform series multiplex operation or parallel multiplex operation of the inverter, and it is easy to increase the capacity. Therefore, it is convenient as a power converter for driving a large capacity AC motor.

(Problems to be Solved by the Invention) The above-described conventional power conversion device has the following problems.

That is, when the output frequency ₀ becomes high and the load current I _{L is} to be controlled by the first inverter INV-1,
The following inconveniences occur.

Normally, the secondary voltage V _L1 of the output transformer TR becomes a voltage proportional to the output voltage of the first inverter INV-1. But,
For example, the load current I by a drift or the like of the current detector CT _L
When DC component is included in _L , or inverter INV-1
When the output voltage contains a small amount of direct current due to variations in the switching characteristics of the self-extinguishing element, the transformer TR is gradually demagnetized by direct current, and eventually the iron core moves in one direction. It saturates. As a result, the secondary voltage V _L1 of the transformer TR decreases, and the load current I _L becomes smaller than the command value ▲ I ^* _L ▼. Therefore, the inverter INV-1 operates to further increase the output voltage V ₁ and further saturate the iron core. Therefore, the primary current I ₁ of the transformer TR suddenly increases and becomes an overcurrent, which may destroy the element of the inverter INV-1. A method of increasing the iron core cross-sectional area and providing an air gap so that the output transformer is not magnetized may be considered, but in that case, the weight and size of the output transformer are increased, which results in an uneconomical system.

The present invention has been made in view of the above problems, and an object of the present invention is to provide a power conversion device that eliminates the magnetic bias of the output transformer and can accurately control the load current.

[Structure of Invention]

(Means for Solving the Problems) In order to achieve the above object, the present invention relates to an AC load, an output voltage of a multiple pulse width modulation control inverter (called a multiple PWM inverter) having an output transformer, and an output. In a power converter that supplies a sum of output voltages of a pulse-width modulation control inverter without transformer (referred to as a direct-coupling PWM inverter), the direct-coupling PWM inverter includes a means for giving a command value of a load current supplied to the AC load, and a load. A load current detection means for detecting a current, and a PWM control means for controlling the output voltage of the direct connection PWM inverter according to the deviation between the load current command value and the load current detection value are provided, and the multiple PWM inverter is the output. A means for obtaining a primary current detection value of the transformer, a means for giving an exciting current command value for the output transformer, a counter electromotive force of the AC load, and an input of the AC load. Means for giving a voltage command value corresponding to the sum of the voltage drop due to the inductance, the load current command value, and a means for obtaining a primary current command value of the output transformer from the exciting current command value, and the primary current command. A PWM control means for controlling the output voltage of the multiple PWM inverter by a signal of a sum of a voltage value and a deviation between the primary current detection value and the voltage command value, or the output transformer Means for detecting the exciting current of the output transformer from the detected primary current value and the detected load current value, means for giving an exciting current command value for the output transformer, the counter electromotive force of the AC load, and the AC A means for giving a voltage command value corresponding to the sum of the voltage drop due to the inductance of the load, a signal corresponding to the deviation between the excitation current command value and the excitation current detection value, and a signal of the sum of the voltage command value By said multiple PWM
It is equipped with PWM control means for controlling the output voltage of the inverter.

(Operation) The direct connection PWM inverter controls the current supplied to the load regardless of the output frequency ₀ .

The multiple PWM inverter is composed of one or a plurality of PWM inverters, and each inverter controls the primary current or exciting current of each output transformer. The command value of the primary current of the transformer is given by the sum of the command value of the load current control and the command value of the exciting current of the transformer.
When the command value of the exciting current is given so that its effective value is substantially constant, the output voltage of each output transformer increases or decreases in proportion to the output frequency. The sum of the output voltage of each transformer and the output voltage of the direct connection inverter is applied to the AC load, and the load current is controlled by adjusting the output voltage of the direct connection inverter. As a result, the multiple PWM inverter bears the voltage proportional to the output frequency, that is, the voltage drop due to the back electromotive force (speed electromotive force) and the inductance in the case of the motor load, and the direct connection inverter takes the remaining necessary voltage, that is, the load. The voltage drop due to the resistor and the transient voltage accompanying the load current control will be borne. In terms of capacity, a direct-coupled inverter is much smaller than a multiple PWM inverter.

When the exciting current of the transformer is controlled to be almost constant, the output voltage of the transformer increases or decreases in proportion to the output frequency, and the iron core does not saturate even if the frequency becomes low. Also, multiple PW
Even if the DC voltage is included in the output voltage due to variations in the switching characteristics of the self-extinguishing elements that compose the M inverter, and the cause of DC bias magnetism in the output transformer occurs, the exciting current of each transformer is controlled by the inverter. Therefore, the above DC voltage is automatically compensated, and each transformer is always operated in a state without magnetic bias. Therefore, one of the transformer
The next current does not become excessive, and the element is not destroyed by the demagnetization of the transformer.

(Embodiment) FIG. 1 is a configuration diagram showing an embodiment of a power conversion device of the present invention, and FIG. 2 is a configuration diagram showing an embodiment of a control circuit of the device of FIG.

In the figure, V _d is the DC voltage source, an INV-1~INV-4 is multiplexed PWM inverter, INV-5 is directly connected to the PWM inverter, TR ₁ to Tr ₄ are output transformer, LOAD AC load (U-phase). Although the converter represents only one output phase, the V phase and W phase are similarly configured.

In FIG. 2, A _{1 to} A ₆ are adders, C _{1 to} C ₅ are comparators, G _{1 to} G ₄ and G _L are current control compensation circuits, K _N is an operational amplifier, and PWM _{1 to} PWM ₅ are Pulse width modulation control circuit, TRG represents a carrier wave (triangular wave) generator. In addition, in Fig. 1, CT ₁ ~ C
T ₅ represents a current detector.

FIG. 3 shows a concrete structure of the PWM inverter INV-1 shown in FIG. 1. The INV-1 is composed of self-extinguishing elements S _{11 to} S ₁₄ and free wheeling diodes D _{11 to} D _14. There is.

FIG. 4 is a time chart diagram for explaining the PWM control operation of the inverter of FIG.

X ₂ and Y ₁ are PWM-controlled carrier signals, and Y ₁ is the inverted value of X ₁ (or a signal with a 180 ° phase shift). Control input signals e ₁ and X ₁ are compared, and elements S ₁₁ and S of inverter INV-1 are compared.
_{Producing 12} gate signals g ₁ . That is, when the e ₁ X ₁ g ₁ = ₁ at S _11: ON, S _12: Off e ₁ <X ₁ 〃 g ₁ = 0 in S _11: Off, S _12: turned on. Also, e ₁ and Y ₁ are compared to _generate the gate signal g ₁ ′ of the elements S ₁₃ and S ₁₄ . That, e ₁ Y ₁ when g ₁ '= 1 at S _13: Off, S _14: On e ₁ <g ₁ when Y' in = 0 S _13: ON, S _14: turned off.

If the output voltage V ₁ of the inverter INV-1 is taken as 1 the primary / secondary turns ratio of the transformer TR _1, when S ₁₁ and S ₁₄ are V ₁ = + V _d S ₁₂ and S ₁₃ when on is on V ₁ = −V _{d In} other modes, V ₁ = 0 and the waveform at the bottom of the figure is obtained. The average value
₁ (indicated by a broken line) has a value proportional to the above-mentioned control input signal e ₁ .

In this way, the output voltage V ₁ of the inverter INV-1 is controlled at a frequency twice the carrier frequency of PWM control.

The other three PWM inverters INV-2 to INV-4 are controlled in the same manner, or the carrier signals X _{2 to} X ₄ of which the phases are shifted by 45 ° in terms of electrical angle are used (Y ₂ ~ Y ₄ is X ₂ ~ X ₄ respectively
Will be the reverse value of). As a result, the output transformer TR ₁ voltage sum _{V 1 + V 2 + V 3} + V 4 that occur through to Tr ₄ becomes a voltage that is multiplexed with respect to the carrier frequency _C of the PWM control, 8-
_The voltage waveform is controlled by _C. Therefore, the output voltage V ₁
In + V ₂ + V ₃ + V ₄ , the low-order harmonic components that accompany PWM control are canceled out, and only the high-order harmonics appear. This higher order harmonic component can be easily removed by a filter such as a reactor.

On the other hand, the direct-coupled inverter INV-5 has a configuration as shown in Fig. 5 and shows the U-phase component of the three-phase GREATS connection, and includes self-extinguishing elements S ₅₁ and S ₅₂ and a freewheeling diode D.
It is composed of ₅₁ and D ₅₂ . The neutral line of the load is DC voltage V
Connected to the middle line of _d .

FIG. 6 is a time chart diagram for explaining the PWM control operation of the inverter INV-5. The carrier signal X ₅ and the control input signal e ₅ are compared to _generate the gate signal g ₅ of the elements S ₅₁ and S ₅₂ . That e ₅ S in g ₅ = 1 when X _{5 51:} ON, S _52: Off e ₅ <S in g ₅ = -1 when X _{5 51:} OFF, S _52: turned on.

Inverter INV-5 of the output voltage V ₅ is S ₅₁ is the on _{V 5 = + (V d /} 2) S 52 〃 V ₅ = - a (V _d / 2). The average value ₅ is a value proportional to the control input signal e ₅ . In this way, the output voltage V ₅
Will be controlled by the carrier frequency of PWM control.

Hereinafter, the control operation of the device of the present invention will be described with reference to FIGS. 1 to 6.

The load current I _U is controlled by the direct connection inverter INV-5.

That is, the input to the comparator C ₅ detects the load current I _U by the current detector CT _5. With the comparator C ₅ , the load current command value ▲ I
^* _U ▼ is compared with the above detection value I _U , and the deviation ε _U = ▲ I ^* _U ▼
− Calculate I _U. The deviation ε _U is the next current control compensation circuit G _L
Amplified by and connected directly to the inverter INV- via adder A _6.
The input signal e ₅ is given to the pulse width modulation control circuit PWM ₅ of ₅ .
As described above, the direct connection inverter INV-5 generates the voltage ₅ proportional to the input signal e ₅ .

For ^{_{▲ I * U ▼> I U}} , the deviation epsilon _U a positive value, increasing the output voltage V ₅ of the direct inverter INV-5, the I _U ≒ ▲ I ^* _U ▼ increasing the load current I _U Controlled as. Conversely, ▲ I
^{When *} _U ▼ <I _U , the deviation ε _U becomes a negative value, the output voltage V ₅ is reduced, and the load current I _U is reduced. Therefore, after all, I _U ≒ ▲ I ^* _U ▼ and calms down. Current command value ▲ I ^* _U
If ▼ is changed to a sine wave, the actual current I _U is also controlled to follow the sine wave.

The direct-coupled inverter INV-5 basically generates a voltage R _L · I _{U corresponding} to the voltage drop due to the resistance R _L of the load. Therefore, a signal obtained by multiplying the current command value ▲ I ^* _U ▼ by the load resistance _RL is added to the adder A ₆ . The input signal e ₅ of PWM ₅ is expressed by the following equation.

e ₅ = _RL · ▲ I ^* _U ▼ ε _U · _GL (1) That is, the direct-coupled inverter INV-5 always generates a voltage drop due to the load resistance, and in addition to this, it depends on the current deviation ε _U. Voltage is generated to control the load current. Since the voltage drop due to the resistance does not depend on the output frequency ₀ ,
_{When 0} = 0, multiple inverter INV- with output transformer
No voltage can be generated from 1 to INV-4, and it is necessary to supply it from the directly connected inverter.

The multiple PWM inverters INV-1 to INV-4 generate a voltage proportional to the output frequency ₀ .

That is, in the case of a motor load, the sum of the voltage of the counter electromotive force V _{CU and} the voltage drop jωL _L · I _U due to the load-side inductance L _L is supplied.

In FIG. 2, the compensation signal ▲ V ^* _CU ▼ + jωL _L obtained by the counter electromotive force compensation value ▲ V ^* _CU ▼ and the load current command value ▲ I ^* _U ▼
The input signal e ₁ of the PWM control circuit PWM ₁ of the inverter INV- ₁ is given to ▲ I ^* _U ▼ via the operational amplifier K _N and the adder A ₂ . Similarly, the input signals e _{2 to} e ₄ of PWM ₂ to PWM ₄ are also given. For convenience of explanation, assuming that other signals are zero, e ₁ = e ₂ = e ₃ = ₄ = (▲ V ^* _CU ▼ + jωL _L・ ▲ I ^* _U ▼) / n (2) ω = 2π・₀ , n = 4. n is the number of stages of the inverters forming the multiple inverter, and in this case, there are four stages.

As a result, the output voltage V ₁ + V ₂ + V ₃ + V ₄ of the multiple inverter becomes a voltage proportional to the sum of the above input signals e _{1 to} e ₄ , and ▲ V ^* _CU
A voltage proportional to ▼ + jωL _L ▲ I ^* _U ▼ is generated. The back electromotive force ▲ V ^* _CU ▼ is a value proportional to the output frequency ₀ , and the second term jωL _L ▲ I ^* _U ▼ is also proportional to ₀ . Therefore, the output voltage V ₁ + V ₂ + V ₃ + V ₄ of the multiple inverter becomes a value proportional to the output frequency ₀ , and the output transformer TR ₁
The effective values of the exciting currents I _{OU1 to} I _OU4 of TR ₄ are almost constant.

However, in actuality, a slight DC bias or the like may be applied to the output transformer due to the drift of the control circuit or the imbalance of the switching characteristics of the elements. When a bias of DC voltage is applied to the transformer, the transformer is gradually demagnetized, and eventually the iron core is saturated and an excessive exciting current flows to the transformer, which not only burns the transformer but also damages the inverter. The constituent element may be destroyed by an overcurrent.

Therefore, in the device of the present invention, the four inverters INV-1 to INV-4 forming the multiple PWM inverter control the primary currents I _{U1 to} I _U4 of the respective output transformers. Inverter INV-1
The control operation of the primary current I _U1 will be described below.

First, the primary current I _U1 of the transformer TR ₁ is detected by the current detector CT _1.
Is detected and input to the comparator C ₁ . In the comparator C _1, it compares the primary current command value ▲ I ^* _U1 ▼ and the current detection value I _U1, deviation epsilon _{1 =} ▲ Request I ^* _U1 ▼ -I _U1. The deviation epsilon ₁ is amplified by the next current control compensation circuit G _1, through the adder A _2, the PWM control circuit PWM ₁ gives the input signal e _1. Therefore, e _{1 in} equation (2) can be rewritten as the following equation.

e ₁ = (▲ V ^* _CU ▼ + jωL _L・ ▲ I ^* _U ▼) / 4 + ε ₁・ G ₁ (3) The above primary current command value ▲ I ^* _U1 ▼ is the load current command value ▲ I
^* _U ▼ and transformer TR ₁ excitation current I _OU1 command value ▲ I ^* _OU1 ▼ given by the sum. _{Assuming that the} four transformers have the same capacity and rating, ▲ I ^* _OU1 ▼ = ▲ I ^* _OU2 ▼ = ▲ I ^* _OU3 ▼ = ▲ I ^*
_OU4 ▼ = ▲ I ^* _OU ▼. Therefore, ▲ I ^* _U1 ▼ = ▲ I ^* _U ▼ + ▲ I ^* _OU ▼ = ▲ I ^* _U2 ▼ = ▲ I ^* _U4 ▼… (4)

When ▲ I ^* _U1 ▼> I _U1 , the deviation ε ₁ has a positive value, the input signal e _{1 of the} equation (3) is increased, the primary current I _U1 is increased, and I _U1 ≈ ▲ I ^* _U1 Control so that it becomes ▼. At this time, the load current I _U is I _U ≈ ▲ I by the direct connection inverter INV-5.
Since it is controlled to ^* _U ▼, as a result, the exciting current I _OU1 of the transformer TR ₁ is controlled so as to match the command value ▲ I ^* _OU ▼.

On the contrary, when ▲ I ^* _U1 ▼ <I _U1 , the deviation ε ₁ becomes a negative value and the input signal e ₁ , that is, the output voltage of the inverter INV-1 is reduced, and I _U1 ≈ ▲ I ^* _U1 Control so that it becomes ▼.

In this way, the primary current I _U1 of the transformer TR ₁ is
Controlled to match I ^* _U1 ▼, resulting in transformer TR
The exciting current I _{OU1 of 1} also matches the command value ▲ I ^* _OU ▼. Excitation current I _OU1 ≈ ▲ I ^* even when DC bias is applied to transformer TR ₁ due to element imbalance
_OU ▼ is reached, and the control system operates so as to finally cancel the DC bias.

The other inverters INV-2 to INV-4 are controlled similarly.

As described above, the multiple PWM inverter is basically a voltage drop j due to the back electromotive force V _CU and the load side inductance L _L.
It is necessary to generate the sum voltage of ω _{L L} I _U. Therefore, it is necessary to adjust the exciting currents I _{OU1 to} I _{OU4 of} the transformers TR _{1 to} TR ₄ to a value corresponding to it. When the mutual inductance at the rated output of each transformer is M, the command value ▲ I ^* _OU ▼ of the exciting current input to the adder A ₁ is given by the following equation.

In this way, the multiple PWM inverter generates a voltage commensurate with the back electromotive force V _{CU of the} load and the voltage drop jωL _L I _U due to the inductance, and the direct connection inverter has the voltage drop due to the resistance of the load and the load. A transient voltage necessary for current control is generated so that the load current I _U is controlled to match its command value ▲ I ^* _U ▼. At this time, even if a DC bias voltage is applied to the output transformer due to variations in the characteristics of the elements, the primary current control of the transformer (in other words, the excitation current control) is performed, so that DC bias magnetization does not occur. Therefore, the overcurrent due to the demagnetization of the transformer is eliminated, and the elements constituting the inverter are not destroyed.

FIG. 7 is a block diagram showing another embodiment of the control circuit of the device of the present invention.

In the figure, SW ₁ and SW ₂ are changeover switch circuits, C _{1 to} C ₆ are comparators, A _{1 to} A ₈ are adders, SH is a Schmitt circuit, G _{1 to} G ₄ and
G _L is a current control compensation circuit, K _N1 , K _N2 and K _N are operational amplifiers, PW
M _{1 to} PWM ₅ are pulse width modulation control circuits, and TRG ₁ and TRG ₂ are carrier wave generators.

The embodiment of FIG. 7 differs from the control circuit of FIG. 2 in the following points.

First, in the control circuit of FIG. 2, the four PWM inverters INV-1 to INV-4 forming the multiple inverter control the primary currents I _{U1 to} I _U4 of the output transformers. The embodiment is different in that the exciting currents I _{OU1 to} I _OU4 of the respective transformers are controlled. When the four transformers have the same capacity and the same rating, the command value ▲ I ^* _OU ▼ of the exciting current is given as shown in the equation (5). Excitation current of each transformer I _OU1 ~
I _OU4 is _calculated as follows from the detected values of the primary currents I _{U1 to} I _U4 of the respective transformers and the detected values of the load current I _U.

A comparator C ₁ and the exciting current detection value I _OU1 its command value ▲
I ^* _OU ▼ is compared, deviation ε ₁ = ▲ I ^* _OU ▼ −I _OU1 is amplified by the control compensation circuit G ₁ , and the inverter INV-1 is controlled by pulse width modulation control, I _OU1 ≈ ▲ I ^* It is controlled to be _OU ▼. The exciting currents I _{OU2 to} I _OU4 of the other transformers are also controlled so as to match the command value ▲ I ^* _OU ▼. As a result, like the control circuit shown in FIG. 2, the output transformer is not demagnetized, so that element destruction can be prevented.

Next difference is that multiple PW is lower than the minimum frequency _min with the output frequency ₀ of the inverter (for example, choose to _min = 1 Hz output transformer as the minimum frequency operable)
The fundamental wave (output frequency component) voltage generated from the M inverter is to be zero.

That is, the comparator C ₆ compares the output frequency ₀ with the set value _{min of the} minimum frequency, and the Schmitt circuit switches the contacts of the switch circuits SW ₁ and SW ₂ . ₀ <
_{When min} , SW ₁ and SW ₂ are connected to the contact a. Therefore, the transformer excitation current command value ▲ I ^* _OU ▼ becomes zero, and the multiple PWM
The basic voltage generated from the inverter is also zero. _{0
When it reaches min} , the switches SW ₁ and SW ₂ are switched to the contact b, and a voltage drop jωL _L I _U due to the back electromotive force V _{CU of the} load and the inductance is generated from the multiple PWM inverter. On the other hand, the direct connection PWM inverter INV-5 is ₀ < _min
At this time, the back electromotive force V _{CU of the} load, the voltage drop jωL _L I _U due to the inductance, and the voltage drop R _L · I _U due to the resistance are all borne to control the load current I _U. In this way
_In the region of ₀ < _min , useless exciting current is prevented from flowing through the output transformers TR _{1 to} TR ₄ .

Further, in the embodiment of FIG. 7, compensation control is performed so that the harmonic voltage generated by the direct-coupled inverter INV-5 is canceled by the multiple inverters INV-1 to INV-4. That is, the PWM control input signal e _{5 of} the direct connection inverter INV- ₅ and the PWM control circuit PWM
The ripple component [Delta] V ₅ of the voltage V ₅ to the inverter INV-5 is generated from the output signal g _{5 5} predicting operation, multiple PWM the inverted value - [Delta] V ₅
The inverters INV-1 to INV-4 are controlled to generate.

As described in the PWM control operation explanatory diagram of the direct connection inverter in FIG. 6, the carrier signal X ₅ and the control input signal e ₅ are compared with each other to generate the gate signal g ₅ .

e when ₅ X ₅ g ₅ = + 1 in S _51: ON, S _52: Off e ₅ <X ₅ 〃 g ₅ = -1 at S _51: Off, S _52: ON, and the output voltage V ₅ is determined It This output voltage V ₅ is separated into a fundamental wave component ₅ and a harmonic wave component ΔV ₅ . That, [Delta] V ₅ = V ₅ - a _5. This is the PWM control input signal e
Substituting ₅ levels gives:

Let C be the _peak value of carrier signals X _{1 to} X ₅ , and set the above gate signal.
Multiply g ₅ by E _max with the operational amplifier K _H. K _H output g ₅・ E
_max is a voltage proportional to the output voltage V ₅ , and from that value
Subtracting input signal e ₅ in proportion to _{5, Δe 5 = g 5 ·} E max -e 5 αΔV 5 ...... (7) is obtained. The output of the adder A ₇ becomes an inverted value of the expression (7) and is multiplied by (1 / 2n) by the operational amplifier K _N2 to obtain the compensation control signal ▲ H ^* _OU ▼.

▲ H ^* _OU ▼ = − (g ₅ · E _max −e ₅ ) / (2n) (8) n is the number of serially connected multiple inverters, where n = 4. Adder A ₅ and via the adder A ₁ to A _4, the compensation signal ▲ H ^* _OU ▼ is input to the PWM ₁ ~PWM _4. Hereinafter, for convenience of explanation, other signals will be described as zero.

In Fig. 8, multiple PW when e ₁ = e ₂ = e ₃ = e ₄ = ▲ H ^* _OU ▼
The operation explanatory view of the M inverter is shown.

X _{1 to} X ₄ and Y _{1 to} Y ₄ are PWM-controlled carrier signals, and Y ₁ to Y ₄ are the inverted values of X _{1 to} X ₄ , respectively. Also, X ₁ , X ₂ , X ₃ , and X ₄ are each 45 degrees out of phase in electrical angle.

The main circuit configuration of the inverter INV-1 is as shown in FIG. 3 as described above.

Compare the input control signal ▲ H ^* _OU ▼ with the carrier wave (triangular wave) X ₁ ,
Generate the gate signal g ₁ and switch the elements S ₁₁ and S _{12 of the} inverter IMV-1.
Control the ignition. That is, when ▲ H ^* _OU ▼ X ₁ , g ₁ = “1” and S ₁₁ : on, S ₁₂ : off When ▲ H ^* _OU ▼ <X ₁ , g ₁ = “0” and S ₁₁ : off, S ₁₂ : on. Further, ▲ H ^* _OU ▼ is compared with Y ₁ , and a gate signal g ₁ ′ is produced to control elements S ₁₃ and S ₁₄ to roll call. That is, when ▲ H ^* _OU ▼ Y ₁ , g ₁ ′ ＝ “1”, S ₁₃ : off, S ₁₄ : on ▲ H ^* _OU ▼ <Y ₁ 〃 g ₁ ′ ＝ “1”, S ₁₃ : on, S ₁₄ : off. As a result, the output voltage V ₁ of the inverter INV- ₁ is S
V ₁ = + V _d when ₁₁ and S ₁₄ are on, V when S ₁₂ and S ₁₃ are on
₁ = -V _d, the waveform shown in FIG. 8 becomes V ₁ = 0 in other modes.

Similarly, the input control signal ▲ H ^* _OU ▼ is compared with the triangular waves X ₂ and Y _2, and the gate signals g ₂ and g ₂ ′ of the inverter INV-2 are generated, and ▲ H
^* _OU ▼ is compared with X ₃ and Y ₃ and gate signal of inverter INV-3
Create g ₃ and g ₃ ′, compare ▲ H ^* _OU ▼ with X ₄ and Y ₄ , and create gate signals g ₄ and g ₄ ′ of the inverter PNV-4.

As a result, the output voltages V _{2 to} V ₄ of the inverters INV-2 to INV- ₄ are as shown in the figure.

The sum of the output voltages V _{1 to} V ₄ of the inverters INV-1 to INV- ₄ is the 8th
The waveform is as shown at the bottom of the figure, and the input signal ▲ H ^* _OU ▼
The value is proportional to. This sum voltage V ₁ + V ₂ + V ₃ + V ₄ = -Δ
V ₅ is a voltage that cancels the harmonic component ΔV ₅ of the voltage V ₅ generated by the directly connected inverter INV-5.

That is, the compensation control signal ▲ H ^* _OU ▼ represented by the equation (8)
Is the PWM control input signal e ₁ = e ₂ of the inverters INV-1 to INV-4.
= E ₃ = e ₄ = ▲ H ^* _OU ▼, the average value of the output voltage V _{1 to} V ₄ of each inverter becomes a value proportional to the input signal e _{1 to} e ₄ and the sum voltage of them. V ₁ + V ₂ + V ₃ + V ₄ is 4 ・
The value is proportional to ▲ H ^* _OU ▼.

Considering equations (7) and (8) in equation (9) with n = 4 Becomes

In this way, compensation control can be performed so that the harmonic voltage ΔV ₅ generated by the direct-coupled inverter INV-5 is canceled by the multiple inverters INV-1 to INV-4. As a result, the carrier frequency of the direct-coupled inverter can be lowered, and switching loss and snubber circuit loss can be reduced.

〔The invention's effect〕

As described above, according to the power converter of the present invention, the multiple PWM
It is possible to prevent DC bias magnetism of the output transformer of the inverter, and it is possible to prevent destruction of the elements constituting the inverter due to overcurrent. At the same time, the load current is accurately controlled by the direct-coupled inverter, and the sinusoidal current with less waveform distortion can be supplied to the AC load.

[Brief description of drawings]

FIG. 1 is a main circuit configuration diagram showing an embodiment of a power conversion device of the present invention, FIG. 2 is an embodiment of a control circuit configuration diagram of the device of FIG. 1, and FIGS. 3 and 5 are shown in FIG. Partial circuit diagram of device, 4th
6 and 6 are time charts for explaining the operation of the device of the present invention, FIG. 7 is a block diagram showing another embodiment of the control circuit of the device of the present invention, and FIG. 8 is an explanation of the operation of FIG. FIG. 9 is a configuration diagram of a conventional power conversion device, and FIG. 10 is a characteristic diagram for explaining the device of FIG. V _d ...... DC voltage source INV-1 to INV-4 ...... Multiple PWM inverter INV-5 ...... Direct connection PWM inverter TR _{1 to} TR ₄ ...... Output transformer LOAD ...... AC load CT _{1 to} CT ₅ ...... Current detection , A _{1 to} A ₈ …… Adder C _{1 to} C ₆ …… Comparator G _{1 to} G ₄ , G _L …… Current control compensation circuit K _N , K _N1 , K _N2 , K _H …… Operational amplifier PWM _{1 to} PWM ₅ ...... Pulse width modulation control circuit TRG, TRG ₁ , TRG ₂ ...... Carrier generator SH …… Schmidt circuit SW ₁ , SW ₂ …… Switch circuit

Claims (5)

[Claims] 1. A power converter for supplying an AC load with a sum of the output voltage of a multiple pulse width modulation control inverter having an output transformer and the output voltage of a pulse width modulation control inverter having no output transformer, A means for giving a command value of a load current supplied to a load, a load current detecting means for obtaining a load current detection value, and a pulse width modulation without the output transformer according to a deviation between the load current command value and the load current detection value. PWM control means for controlling the output voltage of the control inverter, means for obtaining the primary current detection value of the output transformer and means for giving an exciting current command value for the output transformer, counter electromotive force of the AC load, and the AC Means for giving a voltage command value corresponding to the sum of the voltage drop due to the inductance of the load, the load flow command value, and the excitation current command value from the output A means for obtaining a primary current command value of the lance; and a signal corresponding to a deviation between the primary current command value and the primary current detection value, and a signal of a sum of the voltage command value, thereby having the output transformer. A power converter comprising PWM control means for controlling the output voltage of a multiple pulse width modulation control inverter. 2. A power converter for supplying a sum of the output voltage of a multiple pulse width modulation control inverter having an output transformer and the output voltage of a pulse width modulation control inverter having no output transformer to an AC load. A means for giving a command value of a load current supplied to a load, a load current detecting means for obtaining a load current detection value, and a pulse width modulation without the output transformer according to a deviation between the load current command value and the load current detection value. PWM control means for controlling the output voltage of the control inverter, means for detecting the exciting current of the output transformer from the primary current detection value of the output transformer and the load current detection value, and an exciting current command value of the output transformer And a means for giving a voltage command value corresponding to the sum of the back electromotive force of the AC load and the voltage drop due to the inductance of the AC load. Controlling the output voltage of the multiple pulse width modulation control inverter having the output transformer by a signal of the sum of the voltage command value and a signal according to the deviation between the excitation current command value and the excitation current detection value. An electric power converter comprising a PWM control means. 3. The pulse width modulation control inverter without the output transformer comprises means for compensating and controlling so as to generate a voltage corresponding to a voltage drop due to a resistance component of the AC load. The power converter according to item 1 or 2. 4. A multiple pulse width modulation control inverter having an output transformer is provided with means for controlling the exciting current of the output transformer to be zero when the output frequency of the inverter is below a certain minimum frequency. The power conversion device according to claim 1 or 2. 5. A multiple pulse width modulation control inverter having an output transformer comprises means for zeroing the back electromotive force compensation of the load and the voltage drop compensation of the inductance when the output frequency of the inverter is below a certain minimum frequency. The power conversion device according to claim 1 or 2, characterized in that.