JP3994165B2 - Tuning circuit design apparatus and method using tunnel junction elements - Google Patents

Description

  The present invention relates to a tuning circuit design apparatus and method using a tunnel junction element, and more particularly to a tuning circuit design apparatus using two half-wavelength tunnel junction elements optimized for low noise and wideband characteristics. And a method.

A tunnel junction (SIS junction) having a superconductor-insulator-superconductor sandwich structure (SIS structure) has a large capacitance in structure and short-circuits a high-frequency signal by itself. Therefore, in order to efficiently couple the input signal to the SIS junction, it is necessary to reduce the junction size and further integrate a tuning circuit for removing the junction capacitance. In such a tuning circuit, in principle, the tuning can be fractional bandwidth Delta] f / f ₀ is limited by 1 / ωC _J R _N of SIS junction. Here, omega is the angular frequency, the capacitance of C _J is SIS junction, R _N is normal resistance. Therefore, in order to ensure 20% as a ratio band, the value of .omega.C _J R _N product at the center frequency f ₀ is needed to be about 5. .omega.C _J R _N product is strongly dependent on the critical current density J _C of SIS junction, are related by _{J C = ωC S I C R} N / (ωC J R N). Here, C _S is a capacitance per unit area of the SIS junction, and I _C is a critical current. For example, assuming that the junction capacitance per unit area is 100 fF / μm ² , Band10 (789 to 950 GHz), which is the highest frequency band of ALMA (Atacama Large Millimeter-Wave Submillimeter Wave Interferometer), uses approximately 20 kA when using an Nb junction. / cm ^2, a high critical current density of NbN using bonding when about 40 kA / cm ² is required.

In the current SIS junction fabrication technology, the higher the critical current density of the junction, the more the electrical characteristics of the junction tend to deteriorate, which causes an increase in noise temperature due to subgap leakage current and the like. Furthermore, in the ultra-high frequency region where the wavelength is extremely short, such as Band 10, the tuning circuit is also shortened, it becomes difficult to scale down by the conventional design method, and the junction size needs to be about sub μm ^2. It has been difficult to realize a mixer using a SIS junction element having noise and broadband characteristics.

Therefore, the present inventor treats the SIS junction as a distributed constant circuit as a tuning circuit using a relatively large SIS junction in the terahertz band, and configures a resonator with an elongated junction, so that the junction capacitance can be easily increased by the junction itself. A SIS mixer that can be tuned is proposed. However, according to studies of the present inventors, in such a SIS mixer, operating band is limited by conventional manner approximately 1 / ωC _J R _N, is still necessary SIS junction of the high critical current density in a broadband operation Met. The present inventors have further focused on the nature of the distributed constant type tunnel junction, by using a plurality of resonant circuits, it proposes a SIS mixer that can obtain a wide band ratio than 1 / ωC _J R _N (For example, refer to Patent Document 1).
JP 2003-218415 A

As described above, the present inventors focused on the nature of the distributed constant type tunnel junction, and proposed SIS mixer capable of obtaining a wide band ratio than 1 / ωC _J R _N by using a plurality of resonant circuits ing. However, according to the study of the present inventor, the design of such a SIS mixer depends on the experience of the designer. Therefore, even a person who is proficient in design has only designed with the approximate numerical value that he expected to be close to optimum based on experience. For this reason, the design result is not always the optimum design value (optimum solution). Further, when various characteristics such as reflection loss are to be changed, the design burden is extremely large, and the desired characteristics cannot be realized as expected.

  An object of the present invention is to provide an apparatus for designing a tuning circuit using a tunnel junction element capable of designing a tuning circuit optimized for low noise and wide band characteristics.

  It is another object of the present invention to provide a tuning circuit design method using a tunnel junction element capable of designing a tuning circuit optimized for low noise and wideband characteristics.

A tuning circuit design apparatus using a tunnel junction element according to the present invention includes a circuit parameter table for storing circuit parameters determined according to a reflection loss and a ratio band in a signal source, and a table creation for creating the circuit parameter table. And a circuit parameter reading unit that obtains a corresponding circuit parameter by referring to the circuit parameter table using the input reflection loss and specific band in the signal source. The table creating unit is configured to connect the two tunnel junction elements and the microstrip line with pure resistance for a circuit to be processed including two half-wavelength tunnel junction elements and a half-wavelength microstrip line connecting between them. After replacing with a lossy transmission line, moving the pure resistance of the tunnel junction element close to the signal source to the signal source side, standardizing the circuit constant with the pure resistance of the two tunnel junction elements as 1, About a simple circuit model obtained by dividing an input unit connected to the signal source and having an impedance different from the pure resistance, a pure resistance moved to the signal source side, and a three-stage bandpass filter comprising a lossless transmission line, Based on the reflection coefficient in the signal source obtained based on the reflection loss in the signal source, the maximum value of the reflection coefficient of the three-stage bandpass filter is calculated. Because the the three-stage band-pass filter, on the basis maximum value to said input fractional bandwidth, as the reflection coefficient at the center frequency and the predetermined frequency before and after is equal to or less than the maximum value, the circuit parameters The circuit parameter table is created by storing the circuit parameters according to the reflection loss and the ratio band in the signal source.

A tuning circuit design method using a tunnel junction element according to the present invention is a tuning circuit design apparatus using a tunnel junction element, and a reflection coefficient calculation for calculating a maximum value of a reflection coefficient using a reflection loss in a signal source. And a tuning circuit design method using the tunnel junction element, which is executed in a design apparatus including a circuit parameter calculation unit that obtains a circuit parameter using the maximum value and the ratio band . The reflection coefficient calculation unit is configured to process the two tunnel junction elements and the microstrip line as a pure resistance with respect to a processing target circuit including two half-wavelength tunnel junction elements and a half-wavelength microstrip line connecting the two half-wavelength tunnel junction elements. After replacing with a lossless transmission line, the pure resistance of the tunnel junction element close to the signal source is moved to the signal source side, and the circuit resistance is normalized by setting the pure resistance of the two tunnel junction elements to 1 A simple circuit model obtained by dividing the input unit connected to the signal source and having an impedance different from that of the pure resistor, a pure resistor moved to the signal source side, and a three-stage bandpass filter comprising a lossless transmission line read out. Maximum the reflection coefficient calculation portion, front on the basis of the reflection coefficient at the signal source, which has been calculated based on the reflection loss in the relaxin signal source, the reflection coefficient of the three-stage band-pass filter in the read simplified circuit model Ru determine the value. The circuit parameter calculation unit, for the three-stage band-pass filter in the read simplified circuit model, on the basis of the maximum value and the previous SL ratio band, the reflection coefficient at the center frequency and the predetermined frequency before and after The circuit parameters are obtained so as to be equal to or less than the maximum value .

  According to the apparatus and method for designing a tuning circuit using a tunnel junction element according to the present invention, a designer simply inputs a reflection loss and a specific band at a signal source, and two half-wavelength tunnel junction elements between them. A tuning circuit composed of a half-wavelength microstrip line connecting the two is connected to a circuit having an optimum solution that achieves low noise and wideband characteristics, that is, an optimum circuit parameter that realizes the reflection loss and the ratio band. Can be designed. Therefore, it is possible to design without depending on the designer's experience, and even a person who is not proficient in designing can design a tuning circuit using a tunnel junction element having an optimum design value (optimal solution) as a design result. In addition, desired characteristics can be realized by easily changing characteristics such as reflection loss.

(Design principle)
First, the principle used in the tuning circuit design apparatus and design method using the tunnel junction element of the present invention will be described with reference to FIGS.

Among tunnel junction elements, a SIS junction having a particularly elongated structure can be expressed by using transmission line parameters as shown in FIG. This is because the SIS junction having an elongated structure constitutes a microstrip line and can be handled as a transmission line having a loss due to a quasiparticle tunnel current, as is well known. In FIG. 1A, the characteristic impedance of the SIS junction is Z _J , the length is d _J , and the propagation constant is γ _J = α _J + jβ _J. α _J is an attenuation constant due to the loss, and β _J is a phase constant.

When the transmission line is assumed to be an open end, the input impedance Z _in is
Z _in = Z _J coth (γ _J d _J ) (1) If this transmission line is
sin (α _J d _J ) ≈α _J d _J ... If the line satisfies the equation (2), the equation (1) is
_{Z in = Z J × (Z} J / α J d J · cos (β J d J) + jZ J · sin (β J d J)) / (Z J · cos (β J d J) + jZ J / α J d _J · sin (β _J d _J )) ··· (3) can be rewritten.

This equation (3) represents the input impedance of the circuit shown in FIG. Therefore, under the above-described conditions, the open-ended distributed constant type SIS junction shown in FIG. 1A is at the end of the lossless transmission line shown in FIG. 1B (that is, the line with α _J = 0). It can be said that it is equivalent to a circuit having a pure resistance load Z _J / α _J d _J. From the above, it can be considered that the reactance frequency fluctuation component of the distributed constant SIS junction is simply due to the lossless transmission line connected to the pure resistance.

As a method for efficiently tuning such reactance frequency fluctuation components, use of a half-wave bandpass filter structure can be considered. In general, the half-wave bandpass filter is composed of a low impedance line and a high impedance line. Therefore, a SIS junction is assigned to the low impedance line, and a microstrip line is assigned to the high impedance line. In designing such a filter, a well-known filter design theory such as Chebyshev theory can be employed. Since Chebyshev theory is similar to a broadband impedance matching circuit, as is well known, the impedance characteristics can be controlled in a wide band by the filter circuit structure, which is suitable for the filter structure of the present invention (RE Collin: Foundations for Microwave Engineering (McGrow-Hill, New York, 1992) 2nd ed., Chap. 5, p.303-p393 .).

(Design method)
In the present invention, as a processing target circuit (design target circuit), that is, as a filter circuit structure, as shown in FIG. 2A, two half-wavelength (length d _J ) SIS junction elements ( A tuning circuit having the simplest three-stage band-pass filter structure, which is composed of a tunnel junction element) and a microstrip line having a half wavelength (length d _m , d _J ≠ d _m ) connecting between them, is set. The reason for designing in this way will be described later. The design of such a tuning circuit differs from the design of a normal band-pass filter in that there is a loss in a part of the transmission line that constitutes the filter structure, and the input / output impedance is not made the same accordingly. It is. Therefore, in the design apparatus and method of the present invention, a signal source impedance Z _s different from the termination load resistance is assumed, and the maximum reflection loss and its ratio band between this and the tuning circuit are input as design conditions, and the SIS at that time Find the optimal solution that minimizes the critical current density at the junction. This optimum solution is the characteristic impedance of each filter in the three-stage bandpass filter structure, that is, the SIS junction and the microstrip line.

At this time, the signal source impedance can be freely controlled by adding a quarter wavelength impedance matching circuit to such a tuning circuit. Therefore, the value of the signal source impedance Z _s is not limited. Thereby, the optimal solution can be obtained by freely selecting the signal source impedance Z _s .

As described above, a part of such a tuning circuit includes a loss. Therefore, when the circuit is simplified by using an equivalent circuit, the distributed constant SIS junction in the tuning circuit is expressed as a pure resistance by the equation (2). And a lossless transmission line. Furthermore, the phase β _J d _J of the SIS junction is in the vicinity of π, that is, in the vicinity of the frequency (central frequency) at which the distributed constant SIS junction has a half wavelength.
α _J d _J tan (β _J d _J ) << 1 (1) The pure resistance part of the SIS junction close to the signal source may be moved to the signal source side (Z _s side) under the condition satisfying the equation (4). it can. That is, when the SIS junction in the element structure of FIG. 2A is replaced with the simplified equivalent circuit of FIG. 1B, the pure resistance portion of the SIS junction close to the signal source that should be connected between aa ′ is obtained. It can be moved to the signal source side (Z _s side). Therefore, the circuit shown in FIG. 2A can be rewritten to the circuit shown in FIG.

Next, the circuit constant is normalized by the termination load resistance R _L , and the circuit is divided into three as shown in FIG. The circuit of FIG. 2C is referred to as a simple circuit model. In the simple circuit model, the load side from aa ′ has a normal three-stage bandpass filter structure including a lossless transmission line. This filter part is designed by applying Chebyshev theory, but it is necessary to follow conditions that minimize the critical current density of the SIS junction.

That is, the standardized characteristic impedance of the half-wave line connected to the terminating load resistors R _L and aa ′ is α _J d _J. Since α _J is an attenuation constant, the lower the critical current density of the SIS junction, the lower the normalized characteristic impedance. When the line satisfies the equation (2), α _J d _J << 1, so that the transmission line having a lower critical current density of the SIS junction has a higher ratio with the normalized termination load resistance of 1Ω. According to the filter theory, as is well known, the larger the ratio, the larger the maximum reflection coefficient of the filter characteristics (the above-mentioned RE Collin paper). Accordingly, when the reflection coefficient of the circuit when the signal source impedance for the filter portion (a-a ′ to the load side (right side) circuit) is 1Ω is set to ρ, the maximum value is obtained under the condition where ρ is given. If this is taken, the condition for obtaining the minimum critical current density is obtained.

To find this solution, the impedance of the tuning circuit is first described using ρ. Assuming that the input impedance on the load side from aa ′ is Z _f , as is well known,
ρ = | (1−Z _f ) / (1 + Z _f ) | (5) From now on, Z _f can be expressed using ρ.
Z _f = (1 + ρ) / (1-ρ) or Z _f = (1-ρ) / (1 + ρ) (6) As shown in FIG. 2, the input impedance Z _in of the tuning circuit is connected to Z _f in parallel with 1Ω.
Z _in = (1 ± ρ) / 2 (7) Now, considering that the reflection coefficient between the signal source and the tuning circuit is less than a certain value of ρ _s , the relational expression is described using the signal source impedance Z _s ′ normalized by R _L :
ρ _s ≦ | (Z _s ′ −Z _in ) / (Z _s ′ + Z _in ) | (8) is obtained. Thus, [rho is 'a function of, Z _s that maximizes the [rho' Z _s so that is obtained.

(Design target)
In order to use the design principle described above, in the present invention, as a tuning circuit to be designed, two half-wavelength (d _J ) SIS junction elements J1 and J2 shown in FIG. A tuning circuit including a microstrip line M having a wavelength (d _m ) is set. 3A is a plan view of the tuning circuit, FIG. 3B is a cross-sectional view of the tuning circuit, and shows a cross section taken along the 3b-3b cutting line in FIG. FIG. 3A is a diagram corresponding to FIG.

The reason for using the two SIS junctions J1 and J2 is as follows. That can be obtained as described above, a wide enough bandwidth ratio than 1 / ωC _J R _N by using multiple SIS junction. However, on the other hand, when three or more SIS junction elements are used, when the Josephson current unnecessary for tuning is erased by applying a magnetic field, the SIS junction elements cannot cancel each other. If two are used, the Josephson currents can be canceled with each other by using SIS junctions having the same shape. Therefore, the two SIS junction elements J1 and J2 of the present invention have the same shape. As described above, the reason why the SIS junctions J1 and J2 and the microstrip line M have a half-wavelength is that, as described above, the pure resistance of the SIS junction J1 that is close to the signal source using the equation (4) in the vicinity of the half-wavelength frequency. This is because the approximation that moves the part to the signal source side (Z _s side) is used.

Two SIS junction elements J1, J2, to treat as a transmission line having a loss due to quasiparticle tunneling current as described above, and, in order to obtain a wide enough bandwidth ratio than 1 / ωC _J R _N, The shape is elongated. Here, elongate means that the shape can maintain a single propagation mode. Since propagation mode in this example is the TEM mode, orthogonal so as not to cause other propagation modes, while the length d _J of SIS junction elements J1, J2 are half wavelength, its width (d _J For example) is 1/6 wavelength or less.

  As shown in FIG. 3B, the SIS junction elements (tunnel junction elements) J1 and J2 include an upper electrode (S) and a lower electrode (S) made of a superconductor, and an insulating film ( I). The microstrip line M includes an upper line and a lower line (and its interlayer insulating film). The upper electrode and the lower electrode of the SIS junction element and the upper line and the lower line of the microstrip line are made of the same layer.

  In this example, the lower electrodes of the SIS junction elements J1 and J2 and the lower line of the microstrip line M are composed of a first NbN (niobium nitride) layer 102 which is a superconductor. The first NbN layer 102 is a ground electrode (ground plane) formed to a thickness of 200 nm by epitaxial growth on the single crystal MgO substrate 101 which is an insulator. As shown in FIG. 3A, the width of the ground electrode 102 is sufficiently wide (about 5 to 10 times) wider than the widths of the SIS junction elements J1 and J2 and the microstrip line M. Therefore, the microstrip line M becomes a coplanar waveguide. On the first NbN layer 102, an MgO film 103, which is an interlayer insulating film formed to a thickness of 200 nm by epitaxial growth, is formed. After the opening of the tunnel portion is formed in the MgO film 103, the tunnel insulating film 104 made of the MgO film is formed extremely thin (for example, 1 nm) on the exposed first NbN layer 102 by epitaxial growth. The upper electrodes of the SIS junction elements J1 and J2 and the upper line of the microstrip line M are composed of the second NbN layer 105 which is a superconductor. The second NbN layer 105 is a wiring and an electrode formed to a thickness of 400 nm by epitaxial growth on the MgO interlayer film 103. The SIS junctions (tunnel junctions) J1 and J2 are made of NbN / MgO / NbN. Therefore, it is an all NbN tuning circuit.

(Design equipment)
FIG. 4 is a configuration diagram of a tuning circuit design apparatus using a tunnel junction element, and shows a configuration of a tuning circuit design apparatus using a tunnel junction element of the present invention.

  The design apparatus 1 includes a table creation unit 2, a circuit parameter table 3, a circuit parameter read unit 4, and a device design unit 5. The table creation unit 2 includes a reflection coefficient computation unit 21 and a circuit parameter computation unit 22 and creates a circuit parameter table 3. The device design unit 5 uses the circuit parameters read by the circuit parameter reading unit 4 and the input design parameters to design a device that constitutes the design target circuit (tuning circuit) of the present invention. The design apparatus 1 comprises a computer, and the table creation unit 2, the circuit parameter reading unit 4 and the device design unit 5 execute the processing program existing on the main memory (not shown) on the CPU (not shown). It is realized by.

  The reflection coefficient calculation unit 21 calculates the maximum value of the reflection coefficient in a predetermined case using the reflection loss in the signal source, and sends it to the circuit parameter calculation unit 22. The circuit parameter calculation unit 22 calculates the circuit parameter (impedance ratio in the filter portion) for the processing target circuit assumed in the present invention using the calculated maximum value of the reflection coefficient and the ratio band, This is stored in the circuit parameter table 3. The circuit parameter table 3 stores circuit parameters determined according to the reflection loss and the ratio band in the signal source. The reflection loss in the signal source is set in increments of “1 dB” in a practical range, for example, a range of about 0 to −20 dB centering on −10 dB. The specific band is set in increments of “1%” in a practical range, for example, a range of about 10 to 30% centering on 20%.

The circuit parameter table 3 stores circuit parameters for a simple circuit model of a processing target circuit (tuning circuit) composed of two half-wavelength tunnel junction elements and a half-wavelength microstrip line connecting between them. The simple circuit model is a circuit composed of two half-wavelength tunnel junction elements and a half-wavelength microstrip line connecting between the two half-wavelength tunnel junction elements. And the pure resistance of the tunnel junction element close to the signal source is moved to the signal source side, and the circuit constant is normalized with the pure resistance of the two tunnel junction elements set to 1, and then the signal source And divided into a three-stage band-pass filter comprising an input section having an impedance different from that of the pure resistance, a pure resistance moved to the signal source side, and a lossless transmission line. A circuit parameter for the simple circuit model is obtained by obtaining a maximum value of a reflection coefficient of the three-stage bandpass filter based on a reflection coefficient in the signal source obtained based on a reflection loss in the signal source, and The filter is obtained based on the maximum value and the input ratio band so that the reflection coefficient is not more than the maximum value at the center frequency and a predetermined frequency before and after the center frequency. The obtained circuit parameters are stored in the circuit parameter table 3 in accordance with the reflection loss and the specific band at the signal source.

  The table creation unit 2 may be omitted. For example, the circuit parameter table 3 obtained by another design apparatus 1 (the table creation unit 2) may be used by reading it from a medium such as a CD-ROM storing the circuit parameter table 3. The circuit parameter table 3 does not have to be created by the table creation unit 2, and may be obtained by, for example, manually calculating circuit parameters according to the principle of the present invention. The circuit parameter table 3 may be omitted. For example, a process similar to the process executed by the table creating unit 2 may be executed each time to calculate circuit parameters corresponding to the input reflection loss and specific band.

  The circuit parameter reading unit 4 obtains (reads out) a corresponding circuit parameter by referring to the circuit parameter table 3 using the reflection loss and the ratio band in the input signal source, and sends them to the device design unit 5. The device design unit 5 designs the processing target circuit assumed in the present invention using the input design parameter and the circuit parameter, and outputs the result (design value).

  FIG. 5A is a design process flow of a tuning circuit using a tunnel junction element, and shows an example of a design method in a tuning circuit design apparatus using a tunnel junction element of the present invention.

  When the reflection loss and the specific band in the signal source are input (step S1), the circuit parameter reading unit 4 refers to the circuit parameter table 3 using the input reflection loss and the specific band, The corresponding circuit parameter (impedance ratio) is read and sent to the device design unit 5 (step S2). When the design parameters are input (step S3), the device design unit 5 uses this and the circuit parameters from the circuit parameter reading unit 4 to assume the processing target circuit, that is, two half-wavelengths. A device of a tuning circuit composed of a tunnel junction element and a half-wavelength microstrip line connecting the tunnel junction element is designed. (Step S4).

  FIG. 5B is a design processing flow of a tuning circuit using a tunnel junction element, and in particular, a design method in a tuning circuit design apparatus using a tunnel junction element of the present invention when the circuit parameter table 3 is omitted. An example is shown. That is, this is a processing flow in the case where the circuit parameter is calculated each time according to the input of the reflection loss and the specific band, and is executed instead of this in step S2 of FIG.

Since the reflection loss and the ratio band in the signal source are input in step S1, the reflection coefficient calculation unit (corresponding to the processing unit, hereinafter the same) 21 reads the simple circuit model (step S11) and inputs this. The maximum value of the reflection coefficient of the three-stage bandpass filter is obtained based on the reflection coefficient at the signal source obtained based on the reflection loss at the signal source (step S12), and the processing unit corresponding to the circuit parameter calculation unit ( The same shall apply hereinafter) The circuit parameter calculation unit 22 designs a three-stage bandpass filter (filter part) for the three-stage bandpass filter based on the maximum value and the input ratio band (step S13). That is, the circuit parameter calculation unit 22 obtains the circuit parameters so that the reflection coefficient is equal to or less than the maximum value at the center frequency and predetermined frequencies before and after the center frequency. In other words, the impedance ratio for the filter part is calculated, and this (design value) is output as a circuit parameter to the circuit parameter reading unit 4 or the device design unit 5 (in this case, the circuit parameter reading unit 4 is omitted). (Step S14). Thereafter, steps S3 and S4 are executed.

  FIG. 6 is a process flow for creating a circuit parameter table, and shows an example of a method for creating a circuit parameter table in the design apparatus of the present invention.

  The table creation unit 2 reads out the simple circuit model (step S21), and when the initial value, final value, and increment of the reflection loss in the signal source are input, and the initial value, final value, and increment of the ratio band are input ( In step S22), these are sent to the reflection coefficient calculation unit 21 and the circuit parameter calculation unit 22. The circuit parameter calculation unit 22 sets the input initial value of the specific band (for example, 10%) as the value of the specific band (step S23), and the reflection coefficient calculation unit 21 sets the initial value of the input reflection loss (for example, 0 dB) is set as a reflection loss value (step S24). Thereafter, the reflection coefficient calculation unit 21 calculates the maximum value of the simple circuit model from the reflection coefficient obtained based on the set reflection loss (step S25), and sends it to the circuit parameter calculation unit 22. The circuit parameter calculation unit 22 designs a three-stage bandpass filter (filter part) from this maximum value and the set ratio band, that is, calculates a circuit parameter (impedance ratio for the filter part) (step) S26), the design value is stored as a circuit parameter in the corresponding part of the circuit parameter table 3 (step S27).

  After that, the reflection coefficient calculation unit 21 updates the current reflection loss value by one increment (for example, -1 dB) (step S28), and the updated reflection loss value becomes the final value (for example, -20 dB). ) Is checked (step S29). If not, step S25 and subsequent steps are repeated. If it exceeds, the reflection coefficient calculation unit 21 notifies the circuit parameter calculation unit 22 to that effect. In response to this, the circuit parameter calculation unit 22 updates the current ratio band value by one increment (for example, 1%) (step S210), and the updated ratio band value becomes the final value (for example, for example, 30%) is checked (step S211). If not, step S24 and subsequent steps are repeated. If so, the process ends.

(Tuning circuit design)
Using the design apparatus 1 of the present invention, a tuning circuit was designed that satisfies the conditions that the reflection loss with respect to the signal source impedance to the tuning circuit is -10 dB or less and the ratio band is 20%. That is, “reflection loss−10 dB” and “ratio bandwidth 20%” are inputs to the circuit parameter reading unit 4 of the design apparatus 1.

Since ρ _s = 0.314 is obtained from the reflection loss of −10 dB, the equations (7) and (8) are graphed as shown in FIG. The shaded area satisfies the condition of the reflection coefficient ρ. The maximum value of ρ is when Z _s ′ = 0.409, and the maximum value is 0.575. The reflection coefficient calculation unit 21 obtains the maximum value “0.575” of the reflection coefficient.

Therefore, if the signal source impedance Z _s ′ = 0.409 of the tuning circuit with the circuit constant normalized by the termination load resistance R _L , while achieving a reflection coefficient ρ _s = 0.314 or less with respect to the tuning circuit, FIG. The reflection coefficient ρ in the filter circuit structure on the load side can be set to a maximum value of 0.575 from aa ′ of 2 (c). From this result, in applying the Chebyshev theory in the design of the filter portion, the optimum solution of the tuning circuit can be obtained if the filter design is performed under the conditions of the reflection coefficient ρ = 0.575 and the relative bandwidth Δf / f ₀ = 20%. That is, “reflection coefficient ρ = 0.575” and “ratio band Δf / f ₀ = 20%” are given to the circuit parameter calculation unit 22.

  The circuit parameter calculation unit 22 calculates circuit parameters by applying the design principle of a half-wave filter having a well-known broadband Chebyshev characteristic (the above-mentioned R. E. Collin paper). Usually, the filter characteristic is described by the relationship between the phase θ and the reflection coefficient ρ. The basic three-stage half-wave Chebyshev filter characteristics are designed so that the reflection coefficient becomes zero at a total of three points of the center frequency (π in phase) and the frequencies before and after that. FIG. 8A shows ρ-θ characteristics representing typical characteristics of a filter having a maximum reflection coefficient of 0.575. FIG. 8B shows circuit parameters of the filter. The filter design is based on the design of a broadband 1/4 wavelength impedance matching circuit, and the center phase is described as π / 2, so the specific band is 40%. Since the phase at the filter is 2θ, the specific band is 20%.

Therefore, as circuit parameters of the tuning circuit shown in FIG. 2A, as shown in FIG. 8B, R _L : Z _J : Z _m : Z _s = 1: 0.12: 2.69: 0. 41 is obtained. This is a circuit parameter (impedance ratio of the filter portion) calculated by the circuit parameter calculation unit 22 and is a value stored in the circuit parameter table 3. These values are stored at positions corresponding to “reflection loss−10 dB (or the maximum value of the reflection coefficient ρ 0.575)” and “specific band 20%”. Therefore, the circuit parameter reading unit 4 can read out the circuit parameters corresponding to the above-mentioned inputs “reflection loss−10 dB” and “ratio band 20%”.

When the frequency characteristic of the tuning circuit is calculated, the impedance characteristic shown in FIG. 9A and the reflection loss shown in FIG. 9B are obtained. The frequency is standardized at the center frequency. From FIG. 9, it can be seen that, as designed, 20% of the specific band is achieved with the signal source impedance and the reflection loss of the tuning circuit of -10 dB or less. As can be seen from the Smith chart of FIG. 9A, by selecting a lower signal source impedance, the normal Chebyshev characteristic (the intersection of the impedance trajectories passes through the center of the chart, that is, the reflection coefficient is 0 at the frequency of three points). As a result, it can be seen that the maximum value of ρ in FIG. From this, the minimum critical current density is obtained and α _J d _J = 0.12. However, since the wavelength in the line varies depending on the magnetic field penetration length depending on the SIS bonding material and the capacitance per unit area, the minimum critical current density was obtained as follows in the actual device design. .

(Mixer design)
As a mixer having a tuning circuit composed of two distributed constant SIS junctions, a quasi-optical mixer composed of a non-reflective layered MGO super hemisphere lens and a twin slot antenna is designed as an input optical system. Therefore, FIG. 3 shows a tuning circuit of the mixer chip. A feeding point P of a twin slot antenna (not shown) is arranged at the center using a coplanar waveguide (see FIG. 3). The center frequency is designed to be 870 GHz, and the antenna impedance in the vicinity is about 65Ω (for example, J. Zmuidzinas, NG Ugras, D. Miller, M. Gaidis, HG LeDuc, “Low-noise slot antenna SIS mixers,” IEEE Trans. Appl. Supercond., Vol. 5, pp. 3053-3056, 1995. / or M. Gaidis, HG LeDuc, M. Bin, D. Miller, JA Stern and J. Zmuidzinas, “Characterization of low- noise quasi-optical SIS mixers for the submillimeter band, "IEEE Trans. Microwave Theory Tech., vol. 44, pp. 1130-1139, 1996.). The tuning circuit is integrated with the central conductor of one coplanar waveguide as a ground plane. A ¼ wavelength impedance transformer T for matching with the antenna impedance is added to the tuning circuit. As described above, the mixer is manufactured by the epitaxial NbN / MgO / NbN technology using the single crystal MgO substrate, and the tuning circuit is composed of the NbN / MgO / NbN tunnel junction and the NbN / MgO / NbN microstrip line. (For example, A. Kawakami, Z. Wang, and S. Miki, “Low-loss epitaxial NbN / MgO / NbN trilayers for THz applications,” IEEE. Trans. Appl. Supercond., Vol. 11, pp. 80- 83, 2001. / or A. Kawakami, Z. Wang, and S. Miki, “Fabrication and characterization of epitaxial NbN / MgO / NbN Josephson tunnel junctions,” J. Appl. Phys., Vol. 90, pp. 4796 -4799, 2001.).

  The tuning circuit was designed using the analysis result of the simple circuit model of the present invention, with a relative bandwidth of 20% (174 GHz) and a reflection loss of −10 dB or less with respect to the center frequency. The design parameters used in the design are shown in FIG. Various design data including the design parameters are input to the device design unit 5 or given in advance. Although the values of these design parameters are mainly based on actually measured values, the capacitance per unit area of the epitaxial NbN / MgO / NbN junction is assumed to be the same as that of the epitaxially grown NbN / AlN / NbN junction ( Z. Wang, Y. Uzawa, and A. Kawakami, “High current density NbN / AlN / NbN tunnel junctions for submillimeter wave SIS mixers,” IEEE Trans. Appl. Supercond., Vol. 7, pp. 2797-2800, 1997 .) The calculation of the characteristic impedance and propagation constant of the superconducting microstrip line and SIS junction transmission line necessary for the design of the tuning circuit is described by the present inventors in the literature Y. Uzawa and Z. Wang, Studies of High Temperature Superconductor, ed. AV. Narliker (Nova Science, Hauppauge, NY, 2002) Vol. 43, Chap. 9, p. 255.

First, the value of the minimum critical current density of the SIS junction necessary to satisfy the design condition is obtained according to the present invention. As a calculation example, as shown in FIG. 11, the value of α _J d _J = 0.12 of a SIS transmission line having a width of 1 μm at a center frequency of 870 GHz was plotted against the critical current density. From this figure, a critical current density satisfying α _J d _J = 0.12 is obtained as about 16 kA / cm ² (C _S = 110 fF / μm ² ). Compared with the value of 40 kA / cm ² according to the conventional design method, it can be reduced to half or less.

  FIG. 12 shows an outline of a circuit designed to be close to the circuit parameters (characteristic impedance ratio) obtained according to the present invention, and reflection loss characteristics when the antenna impedance is a constant value of 65Ω. FIG. 12 also shows the impedance locus when the load side is viewed from each point of the circuit. These are standardized at 65Ω. It can be seen that, according to the design principle of the present invention, the reactance component having the frequency dependence of the half-wave junction placed at the end is well compensated by the tuning circuit. In addition, by adding a ¼ wavelength impedance transformer T (the same in FIG. 3) to the tuning circuit, a ratio band of 20% or more was obtained. In order to compare with the tuning circuit of the conventional design method, the reflection loss characteristic of the “full wavelength” junction with the 1/4 wavelength impedance transformer T having the same critical current density and the same total junction length as the two junction tuning circuit is shown. This is indicated by a broken line in FIG. From this, it can be seen that the ratio band indicated by the broken line is significantly narrow, and that the tuning circuit according to the present invention is effective in reducing the critical current density.

  As described above, according to the present invention, in a tuning circuit design apparatus and design method using a tunnel junction element, two half-wave tunnels can be obtained simply by inputting a reflection loss and a specific band in a signal source. A tuning circuit composed of a junction element and a half-wavelength microstrip line connecting between the junction elements can be designed so as to have an optimum solution that can obtain a low-noise and broadband characteristic. Therefore, it is possible to design without depending on the designer's experience, and even a person who is not proficient in designing can design a tuning circuit using a tunnel junction element having an optimum design value (optimal solution) as a design result. In addition, desired characteristics can be realized by easily changing characteristics such as reflection loss.

It is a design principle explanatory view of a tuning circuit using a tunnel junction element. It is a design principle explanatory view of a tuning circuit using a tunnel junction element. 2 shows a tuning circuit using a tunnel junction element. 1 shows an apparatus for designing a tuning circuit using a tunnel junction element. 6 shows a design processing flow of a tuning circuit using a tunnel junction element. A flow of creating a circuit parameter table is shown. It is a figure which shows the example of a design of the tuning circuit using a tunnel junction element. It is a figure which shows the example of a design of the tuning circuit using a tunnel junction element. It is a figure which shows the example of a design of the tuning circuit using a tunnel junction element. It is a figure which shows the design parameter of the tuning circuit using a tunnel junction element. It is a figure which shows the example of a design of the tuning circuit using a tunnel junction element. It is a figure which shows the example of a design of the tuning circuit using a tunnel junction element.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 Design apparatus 2 Table preparation part 3 Circuit parameter table 4 Circuit parameter reading part 5 Device design part 21 Reflection coefficient calculating part 22 Circuit parameter calculating part 101 Single crystal MgO substrate 102 1st NbN layer 103 MgO film 104 Tunnel insulating film 105 2nd NbN layer

Claims (4)

A circuit parameter table for storing circuit parameters determined according to the reflection loss and the ratio band in the signal source;
A table creation unit for creating the circuit parameter table;
A circuit parameter reading unit that obtains a corresponding circuit parameter by referring to the circuit parameter table using a reflection loss and a specific band in the input signal source;
The table creation unit
For a circuit to be processed consisting of two half-wavelength tunnel junction elements and a half-wavelength microstrip line connecting between them, the two tunnel junction elements and the microstrip line are replaced with pure resistors and lossless transmission lines. Then, the pure resistance of the tunnel junction element close to the signal source is moved to the signal source side, the circuit constants are normalized with the pure resistance of the two tunnel junction elements set to 1, and then connected to the signal source. About a simple circuit model obtained by dividing an input unit having an impedance different from the pure resistance, a pure resistance moved to the signal source side, and a three-stage bandpass filter composed of a lossless transmission line,
Based on the reflection coefficient in the signal source determined based on the reflection loss in the signal source, to determine the maximum value of the reflection coefficient of the three-stage bandpass filter,
A circuit parameter of the three-stage bandpass filter is determined based on the maximum value and the input ratio band so that a reflection coefficient is equal to or less than the maximum value at a center frequency and a predetermined frequency before and after the center frequency. ,
An apparatus for designing a tuning circuit using a tunnel junction element , wherein the circuit parameter table is created by storing the circuit parameter according to a reflection loss and a specific band in the signal source. The design apparatus further includes
A device design unit that designs a tuning circuit that is the processing target circuit using the circuit parameters obtained by the circuit parameter reading unit and the input design parameters,
The device design unit obtains the impedance of the signal source when the reflection coefficient of the three-stage bandpass filter takes a maximum value based on the reflection coefficient of the signal source obtained based on the input reflection loss. ,
The device design unit adjusts the impedance of the signal source by connecting the signal source and a tunnel junction element close to the signal source with a ¼ wavelength microstrip line serving as an impedance matching circuit. The tuning circuit design apparatus using the tunnel junction element according to claim 1. The tunnel junction element comprises an upper electrode, a lower electrode, and a tunnel junction formed between them.
The microstrip line consists of an upper line and a lower line,
The tuning circuit using a tunnel junction element according to claim 1, wherein the upper electrode and the lower electrode of the tunnel junction element and the upper line and the lower line of the microstrip line are each made of the same layer. Design equipment. A tuning circuit design apparatus using a tunnel junction element , wherein a reflection parameter calculation unit that calculates a maximum value of a reflection coefficient using reflection loss in a signal source, and circuit parameters are calculated using the maximum value and a ratio band In a design method of a tuning circuit using the tunnel junction element, which is executed in a design apparatus including a circuit parameter calculation unit to be obtained
The reflection coefficient calculation unit is configured to process the two tunnel junction elements and the microstrip line as a pure resistance with respect to a processing target circuit including two half-wavelength tunnel junction elements and a half-wavelength microstrip line connecting the two half-wavelength tunnel junction elements. After replacing with a lossless transmission line, the pure resistance of the tunnel junction element close to the signal source is moved to the signal source side, and the circuit resistance is normalized by setting the pure resistance of the two tunnel junction elements to 1 A simple circuit model obtained by dividing the input unit connected to the signal source and having an impedance different from that of the pure resistor, a pure resistor moved to the signal source side, and a three-stage bandpass filter comprising a lossless transmission line Read ,
Maximum the reflection coefficient calculation portion, front on the basis of the reflection coefficient at the signal source, which has been calculated based on the reflection loss in the relaxin signal source, the reflection coefficient of the three-stage band-pass filter in the read simplified circuit model Find the value
The circuit parameter calculation unit, for the three-stage band-pass filter in the read simplified circuit model, on the basis of the maximum value and the previous SL ratio band, the reflection coefficient at the center frequency and the predetermined frequency before and after A circuit parameter is determined so as to be equal to or less than the maximum value . A tuning circuit design method using a tunnel junction element.

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