JP7786149B2 - Power conversion device and control method thereof - Google Patents

Description

This disclosure relates to a power conversion device and a control method thereof.

The modular multilevel converter (MMC) is a next-generation transformerless power converter suitable for large-capacity, high-voltage applications. MMCs can be applied, for example, to static synchronous compensation (STATCOM) systems and high-voltage direct current (HVDC) systems. Delta-connected MMCs, in particular, have attracted attention because they can pass a circulating current through the delta connection and output a negative-phase reactive current. Delta-connected MMCs have an internal delta connection, and one or multiple bridge cells connected in series are provided on each phase of the delta connection (see, for example, Patent Document 1).

A bridge cell has multiple semiconductor switch groups, each containing multiple semiconductor switches connected in series, and a DC capacitor connected in parallel to the multiple semiconductor switch groups.

Patent No. 5800154

In an MMC, for example, if an unbalanced fault occurs in the power system, non-zero active power will steadily flow into each phase, which can cause fluctuations in the voltage of the DC capacitor. As a countermeasure, a circulating current (zero-phase current) can be passed through the delta connection to maintain the orthogonal relationship between the AC voltage and AC current of each phase, thereby maintaining the voltage balance of the DC capacitor.

Conventional methods, including those described in Patent Document 1, calculate circulating current commands based on the error between the average voltage of all three-phase DC capacitors and the average voltage of each phase's capacitor. However, because this calculation method is based on DC capacitor voltage error, if an unbalanced fault occurs in the power system, the circulating current may respond slowly to the fault, causing the voltage of each phase's DC capacitor to fluctuate significantly. If an overvoltage occurs due to voltage fluctuations in the DC capacitors, the protection circuit may shut down the operation of the power conversion device.

This disclosure provides a power conversion device and a control method for the same that can reduce capacitor voltage fluctuations.

In one aspect of the present disclosure,
a delta connection section in which one or a plurality of series-connected bridge cells are delta-connected;
a control device that controls a circulating current flowing in the delta connection by the plurality of bridge cells,
The bridge cell includes a plurality of semiconductor switch groups each including a plurality of semiconductor switches connected in series, and a capacitor connected in parallel to the plurality of semiconductor switch groups,
A power conversion device is provided in which the control device uses the positive-phase dq-axis components ( _vd ⁺ , _vq ⁺ ) of the voltage of the power system to which the delta connection section is connected, the negative-phase dq-axis components ( _vd- ^, _vq- ⁾ of the voltage of the power system, the positive-phase dq-axis components ( _id ⁺ , _iq ⁺ ) of the current flowing in each phase of the delta connection section, and the negative-phase dq-axis components ( _id- , _iq- ⁾ of the current flowing in each phase of the delta connection section to calculate the command value ^of the circulating current.

In another aspect of the present disclosure,
A control method for a power conversion device including a delta connection portion in which one or a plurality of series-connected bridge cells are delta-connected, the control method controlling a circulating current flowing through the delta connection portion by the plurality of bridge cells,
The bridge cell includes a plurality of semiconductor switch groups each including a plurality of semiconductor switches connected in series, and a capacitor connected in parallel to the plurality of semiconductor switch groups,
A control method for a power conversion device is provided in which the positive-phase dq-axis components ( _vd ⁺ , _vq ⁺ ) of the voltage of the power system to which the delta connection section is connected, the negative-phase dq-axis components ( _vd- ^, _vq- ) of the voltage of the power system, the positive-phase dq-axis components ( _id ⁺ , _iq ⁺ ) of the current flowing in each phase of the delta connection section, and the negative-phase dq ^- axis components ( _id- ^, _iq- ⁾ of the current flowing in each phase of the delta connection section are used to calculate the command value of the circulating current.

This disclosure makes it possible to reduce voltage fluctuations in capacitors.

1 is a diagram illustrating a configuration example of a power conversion device according to an embodiment; FIG. 2 is a diagram illustrating an example of the configuration of a control device. FIG. 2 is a diagram illustrating an example of the configuration of a circulating current command unit. FIG. 10 is a diagram showing an example of a simulation waveform at the time of a system unbalance fault when a conventional method is used for circulating current command calculation. FIG. 10 is a diagram showing an example of a simulation waveform at the time of a system unbalance fault when the method disclosed herein is used for circulating current command calculation.

Embodiments of the present disclosure will be described below with reference to the drawings.

Figure 1 shows an example of the configuration of a power conversion device in one embodiment, illustrating an example of the circuit configuration of a delta-connected MMC. The MMC can be applied to, for example, a static var compensator (STATCOM) or a high-voltage direct current (HVDC) transmission system. The power conversion device 400 shown in Figure 1 includes a delta-connected unit 402 and a control device 401.

The delta connection unit 402 is a circuit in which one or more bridge cells connected in series are delta-connected. The delta connection unit 402 includes multiple clusters 50 (50UV, 50VW, 50WU) and multiple reactors 51 (51UV, 51VW, 51WU).

The UV-phase cluster 50UV has a plurality of bridge cells _52UV1 , _52UV2 , ... _52UVn connected in series via a pair of AC output terminals a, b. Three bridge cells 52UV1, 52UV2, _52UV3 are shown in Figure 1 as an example. The VW-phase cluster 50VW has a plurality of bridge cells _52VW1 , _52VW2 , ... _52VWn connected in series via a pair of AC output terminals a, b. Three bridge cells _52VW1 , _52VW2 , _52VW3 are shown in Figure 1 as an example. The _WU -phase cluster 50WU has a plurality of bridge cells _52WU1 , _52WU2 , _{... 52WUn} connected in series via a pair of AC output terminals a, b. 1 illustrates three bridge cells _52WU1 , _52WU2 , and _52WU3 . n represents the number of bridge cells connected in series in each cluster and is an integer equal to or greater than 1. In other words, the number of bridge cells in one cluster may be as small as one. The UV phase, VW phase, and WU phase are also simply referred to as the U phase, V phase, and W phase, respectively.

Each of the bridge cells 52 (52UV ₁ to 52UV _n , 52VW ₁ to 52VW _n , 52WU ₁ to 52WU _n ) has a pair of AC output terminals a, b, and is connected in series via the pair of AC output terminals a, b. Each of the bridge cells 52 has its first AC output terminal a connected to the second AC output terminal b of one of the bridge cells adjacent to it, and its second AC output terminal b connected to the first AC output terminal a of the other bridge cell adjacent to it.

Cluster 50UV, cluster 50VW, and cluster 50WU are delta-connected via reactors 51UV, 51VW, and 51WU, and are connected to the power grid 300. Connection to the power grid 300 may also be via a transformer (not shown). A circulating current flows within the delta connection. The control device 401 can adjust the negative-phase reactive current by controlling the circulating current flowing in the delta connection section 402 through the switching of multiple bridge cells 52.

Each of the multiple bridge cells 52 has a power conversion circuit with multiple switching elements and a drive circuit that operates the power conversion circuit. The multiple bridge cells 52 have the same configuration. The switching elements are, for example, semiconductor switches that have a transistor and a diode connected in anti-parallel to the transistor. Specific examples of transistors include IGBTs (Insulated Gate Bipolar Transistors) and MOSFETs (Metal Oxide Semiconductor Field Effect Transistors).

Each bridge cell 52 has the function of converting DC power in the capacitor 54 into AC power and outputting it to a pair of AC output terminals a and b, and the function of converting AC power input from the pair of AC output terminals a and b into DC power and supplying it to the capacitor 54.

Each bridge cell 52 has a pair of AC output terminals a, b, a capacitor 54, a power conversion circuit 53, and a drive circuit unit (not shown) (e.g., a GDU (Gate Drive Unit) and a power supply circuit, etc.).

Capacitor 54 is a capacitive element connected to a pair of AC output terminals a and b via power conversion circuit 53.

The power conversion circuit 53 is an inverter circuit connected between the capacitor 54 and a pair of AC output terminals a and b, and converts power bidirectionally between DC and AC. The power conversion circuit 53 is connected in parallel to the capacitor 54. Figure 1 shows an example of a full-bridge circuit having multiple switching elements 56.

The power conversion circuit 53 has multiple semiconductor switch groups 55, each including multiple semiconductor switches connected in series. The multiple semiconductor switch groups 55 are connected in parallel to the capacitor 54. The power conversion circuit 53 shown in FIG. 1 has a full-bridge configuration in which a first semiconductor switch group including multiple switching elements 56 connected in series and a second semiconductor switch group including multiple switching elements 56 connected in series are connected in parallel. A first AC output terminal a is connected to the connection point between the switching element 56 of the first upper arm and the switching element 56 of the first lower arm. A second AC output terminal b is connected to the connection point between the switching element 56 of the second upper arm and the switching element 56 of the second lower arm.

The multiple switching elements 56 shown in Figure 1 are IGBTs with diodes connected in antiparallel, but they may also be switching elements with switching functions such as MOSFETs or thyristors.

At least one of the switching element and the anti-parallel diode is preferably an element including a wide bandgap semiconductor such as _SiC (silicon carbide), GaN (gallium nitride), _Ga2O3 (gallium oxide), or diamond. By applying a wide bandgap semiconductor to the switching element, the effect of reducing loss in the switching element is enhanced. The switching element may also be an element including a semiconductor such as Si (silicon). Similarly, by applying an element including a wide bandgap semiconductor to the diode, the effect of reducing loss in the diode is enhanced. The diode may also be an element including a semiconductor such as Si (silicon).

Each bridge cell 52 is equipped with a drive circuit unit (not shown), such as a GDU and a power supply circuit.

The GDU is a drive circuit that drives the power conversion circuit 53, and more specifically, a gate drive circuit that drives the gates of multiple switching elements 56 configured in the power conversion circuit 53. The GDU drives the multiple switching elements 56 configured in the power conversion circuit 53 based on power supplied from the capacitor 54 via the power supply circuit.

The GDU applies a voltage between the gate and emitter of a corresponding one of the multiple switching elements 56 in accordance with a control signal from the control device 401, thereby turning the corresponding switching element on or off. This operation generates a square-wave voltage between a pair of AC output terminals a and b of the bridge cell 52.

The control device 401 is a controller that generates control signals (e.g., PWM signals (pulse-width modulated signals)) that turn on or off multiple switching elements 56 in accordance with a carrier period Tc (the inverse of the carrier frequency) common to multiple bridge cells 52. The control device 401 has a memory and a processor (e.g., a CPU (Central Processing Unit)). Each function of the control device 401 is realized by the processor operating in accordance with a program stored in the memory. The functions of the control device 401 may also be realized by an FPGA (Field Programmable Gate Array) or an ASIC (Application Specific Integrated Circuit).

The power conversion device 400 can output a multi-level voltage waveform with reduced harmonics and a voltage equal to or greater than the withstand voltage of the switching elements by having the control device 401 cause each of the multiple bridge cells 52 to output voltage waveforms with different phases. Therefore, the power conversion device 400 can be applied to, for example, reactive power compensation devices and DC transmission systems directly connected to extra-high voltage grids.

Furthermore, in an MMC, if an unbalanced fault occurs in the power system 300, for example, non-zero active power may steadily flow into each phase, causing fluctuations in the voltage of capacitor 54. As a countermeasure, the control device 401 of the power conversion device 400 controls the circulating current (zero-phase current) flowing through the delta connection 402, thereby maintaining the orthogonal relationship between the AC voltage and AC current of each phase and maintaining the voltage balance of capacitor 54.

Figure 2 is a diagram showing an example configuration of a control device. The control device 401 shown in Figure 2 includes a circulating current command unit 21, a circulating current control unit 22, a U-phase balance control unit 23-u, a V-phase balance control unit 23-v, a W-phase balance control unit 23-w, a power control unit 24, and a voltage command value generation unit 25. Each unit of the control device 401 is implemented by a processing device such as a DSP or FPGA. In this example, the number of bridge cells is set to three, so n = 1, 2, and 3 in Figure 2.

The circulating current command unit 21 is an arithmetic circuit that calculates a circulating current command value i ₀ ^* as a command for the circulating current to be flowed through the delta connection unit 402. The circulating current control unit 22, the U-phase balance control unit 23-u, the V-phase balance control unit 23-v, the W-phase balance control unit 23-w, the power control unit 24, and the voltage command value generation unit 25 may have the same configuration as that disclosed in the above-mentioned Patent Document 1.

1 , the line voltage _vS of the power system 300 is ( _vSuv , _vSvw , _vSwu ). The current i flowing from the MMC to the power system 300 is ( _iu , _iv , _iw ). The currents (inverter currents) flowing through each phase of the delta connection 402 are ( _iuv , _ivw , _iwu ). The voltages of the capacitors 54 in each bridge cell 52 are ( _vCnu , _vCnv , _vCnw ). In this example, n is 1, 2, and 3.

The relationships shown in equations 2 to 4 hold between the currents (i _u , i _v , i _w ) flowing from the power conversion device 400 to the power system 300 and the currents (i _uv , i _vw , i _wu ) flowing into each phase of the delta connection 402.

At this time, the circulating current i ₀ flowing through the delta connection 402 is expressed by Equation 5.

The present disclosure provides a command calculation method for controlling this circulating current.

When the positive-phase dq-axis components of the voltage (system voltage) of the power system 300 to which the delta connection unit 402 is connected are denoted by v _d ⁺ and v _q ⁺ , the negative-phase dq-axis components of the system voltage are denoted by v _d ^- and v _q ^- , and the imaginary unit is j, the three-phase system voltages (v _Suv , v _Svw , v _Swu ) are expressed as follows:

It is expressed as follows.

The positive-phase dq-axis components of the inverter current are defined as i _d ⁺ and i _q ⁺ , and the negative-phase dq-axis components of the inverter current are defined as i _d ^- and i _q ^- . The circulating current to be passed through the delta connection 402 in order to suppress DC voltage fluctuations of the MMC (voltage fluctuations of the capacitor 54) is defined as follows:

Then, the three-phase inverter current (i _uv , i _vw , i _wu ) is

It is expressed as follows.

The orthogonal relationship between the vector of the system voltage and the vector of the inverter current is maintained by circulating the zero-phase current (circulating current i ₀ ) in the delta connection 402. The condition under which the vector of the system voltage and the vector of the inverter current are orthogonal is as follows:

It is expressed as follows.

From Equation 13,

holds true.

From Equation 14,

holds true.

From Equation 15,

holds true.

Equations 19 and 20 can be obtained from equations 16 to 18.

Therefore, the circulating current to be passed through the delta connection 402 is

The circulating current command unit 21 shown in FIG. 2 calculates a circulating current command value i ₀ ^* expressed by Equation 21 as a command for the circulating current to be passed through the delta connection unit 402 .

3 is a diagram showing an example of the configuration of a circulating current command unit. The circulating current command unit 21 performs three-phase/two-phase transformation of the system voltages ( _vSuv , _vSvw , _vSwu ) using a three-phase/two-phase transformation block. The circulating current command unit 21 performs a positive-phase dq transformation on the result of the three-phase/two-phase transformation of the system voltages ( _vSuv , _vSvw , _vSwu ) using a positive-phase dq transformation block and performs a moving average using a moving average block to generate positive-phase dq-axis voltages ( _vd ⁺ , _vq ⁺ ). Similarly, the circulating current command unit 21 performs a negative-phase dq transformation on the result of the three-phase/two-phase transformation of the system voltages ( _vSuv , _vSvw , _vSwu ) using a negative-phase dq transformation block and performs a moving average using a moving average block to generate negative-phase dq ^- axis voltages ⁽ _vd- , _vq- ). Meanwhile, the circulating current command unit 21 performs three-phase/two-phase transformation of the inverter currents (i _uv , i _vw , i _wu ) using a three-phase/two-phase transformation block. The circulating current command unit 21 performs positive-phase dq transformation on the result of the three-phase/two-phase transformation of the inverter currents (i _uv , i _vw , i _wu ) using a positive-phase dq transformation block and performs moving averaging using a moving average block to generate positive-phase dq-axis currents (i _d ⁺ , i _q ⁺ ). Similarly, the circulating current command unit 21 performs negative-phase dq transformation on the result of the three-phase/two-phase transformation of the inverter currents (i _uv , i _vw , i _wu ) using a negative-phase dq transformation block and performs moving averaging using a moving average block to generate negative-phase dq-axis currents (i _d ^- , i _q ^- ).

The circulating current command unit 21 uses the positive-phase dq-axis voltage ( _vd ⁺ , _vq ⁺ ), negative-phase dq-axis voltage ( _vd- , _vq- ), positive-phase dq-axis current ( _id ⁺ , _iq ⁺ ) and negative-phase dq-axis current ( _id- ^, _iq- ⁾ obtained in ^this manner to calculate ^the circulating current command value _i0 ^* based on equation 21.

The circulating current _i0 is calculated by Equation 5, and the circulating current control unit 22 shown in Fig. 2 forms a feedback loop that causes this circulating current _i0 to follow the circulating current command value _i0 ^* . That is, the circulating current control unit 22 generates a first command value _vA * common to each phase to control the switching operation of the switching element 56 so that the circulating current _i0 follows the circulating current command value _i0 ^* generated by the circulating current command unit ^21. The configuration of the circulating current control unit 22 may be the same as that shown in Fig. 4 of Patent Document 1 mentioned above.

The configurations of the U-phase balance control unit 23-u, V-phase balance control unit 23-v, and W-phase balance control unit 23-w shown in Figure 2 may be the same as those shown in Figure 5 of the above-mentioned Patent Document 1. Balancing control is performed for each phase and for each bridge cell within that phase. In this example, the number of bridge cells is set to three, so n = 1, 2, and 3 in Figure 2.

The U-phase balance control unit 23-u, the V-phase balance control unit 23-v, and the W-phase balance control unit 23-w achieve voltage balance by forming effective power between the output voltages (v _Cnu , v _Cnv , v _Cnw ) of each bridge cell 52 and the inverter currents (i _uv , i _vw , i _wu ) flowing into each phase.

That is, for each bridge cell 52, the balance control unit of each phase generates second command values v Bnu *, v Bnv *, v Bnw * for controlling the switching operation of the switching element 56 in the bridge cell using the value obtained by multiplying the deviation between the DC capacitor phase ^- specific average value of that phase and the voltage value of the DC capacitor in that _bridge cell by ^the value of the AC _current flowing into _that ^phase .

Specifically, the U-phase balance control unit 23-u generates the second command value v _{Bnu *} for the U-phase by multiplying the deviation between the U-phase capacitor voltage average value v _Cuave and the voltage v Cnu of the U-phase capacitor 54 by the value of the inverter current i _uv flowing into the U-phase, and then multiplying this result by a _gain ^K _4. The U-phase capacitor voltage average value v _Cuave is a value obtained by averaging the voltage values of the U-phase capacitor 54.

The V-phase balance control unit 23-v generates a second command value v _Bnv * for the V-phase by multiplying the deviation between the V-phase capacitor voltage average value v _Cvave and the voltage v _Cnv of the V-phase capacitor 54 by the value of the inverter current i _vw flowing into the V-phase, and then multiplying this result by a gain K _4. The ^V -phase capacitor voltage average value v _Cvave is a value obtained by averaging the voltage values of the V-phase capacitor 54.

The W-phase balance control unit 23-w generates a second command value v _Bnw * for the W phase by multiplying the deviation between the W-phase capacitor voltage average value v _Cwave and the voltage v _Cnw of the W-phase capacitor 54 by the value of the inverter current i _wu flowing into the W phase and then multiplying this result by a gain K _4. The ^W -phase capacitor voltage average value v _Cwave is a value obtained by averaging the voltage values of the W-phase capacitor 54.

The configuration of the power control unit 24 shown in FIG. 2 may be the same as that shown in FIG. 6 of the above-mentioned Patent Document 1. The power control unit 24 generates third command values v _uv ^* , v vw *, and v _wu ^{* ,} which are line voltage command values for performing at least one of positive-phase reactive power control, negative-phase reactive power control, _and active power control. The block diagram shown in FIG. 6 of the above-mentioned Patent Document 1 is a general control block for positive-phase reactive power control, negative-phase reactive power control, and active power control in a static synchronous compensation (STATCOM) system. p ^* represents the source-side instantaneous active power command value, and q ^* represents the source-side instantaneous reactive power command value. v _Cave represents a value obtained by averaging the voltage values of all three-phase capacitors 54, and v _C ^* represents the voltage command value of the capacitor 54. i represents the current flowing from the delta connection 402 to the power system 300, and v _S represents the line voltage of the power system 300.

The first command value v _A ^* generated by the circulating current command unit 21 and the circulating current control unit 22, the second command values v _Bnu ^* , v _Bnv ^* and v _Bnw ^* generated by the balance control units of each phase, and the third command values v _uv ^* , v _vw ^* and v _wu ^* generated by the power control unit 24 are input to the voltage command value generation unit 25.

The configuration of the voltage command value generation unit 25 shown in Fig. 2 may be the same as that shown in Fig. 7 of Patent Document 1. The voltage command value generation unit 25 functions as a switching command value generation unit for generating switching command values that control the switching operation of the switching elements 56 in the bridge cells 52 of each phase. The voltage command value generation unit 25 adds together the first command value _vA ^* , the second command value _vBnu ^* , and a value obtained by dividing the third command value _vuv ^* by the number of bridge cells ("3" in this example) to generate the voltage command value _vnu ^* for the bridge cell 52 of the U phase. The voltage command value generation unit 25 adds together the first command value _vA ^* , the second command value _vBnv ^* , and a value obtained by dividing the third command value _vvw ^* by the number of bridge cells ("3" in this example) to generate the voltage command value _vnv ^* for the bridge cell 52 of the V phase. The voltage command value generating unit 25 adds the first command value v _A ^* , the second command value v _Bnw ^* , and the value obtained by dividing the third command value v _wu ^* by the number of bridge cells (in this example, "3"), to generate the voltage command value v _nw ^* for the W-phase bridge cell 52.

The generated voltage command values v _nu ^* , v _nv ^* , and v _nw ^* are normalized by the voltage of the capacitor 54 of each phase and used as switching command values. These switching command values are compared by a switching control means (not shown) with a triangular wave carrier signal (maximum value: 1, minimum value: −1) of carrier frequency fc to generate PWM switching signals. The generation of these PWM switching signals is realized using an arithmetic processing device such as a DSP or FPGA. The generated PWM switching signals are used to control the switching of the switching elements 56 in the corresponding bridge cells.

FIG. 4 shows an example of a simulated waveform at the time of a grid unbalance fault when a conventional method (Patent Document 1) is used for calculating a circulating current command. The simulation conditions are as follows:
System unbalance fault: U-phase ground fault U-phase residual voltage: 50%
U phase ground fault period: 150ms
MMC operation mode: rated capacitor operation When a U-phase ground fault occurs at t=1 s, the voltage of capacitor 54 fluctuates up to 30% in the conventional method.

Figure 5 shows an example of a simulated waveform during a system unbalance fault when the disclosed method is used to calculate the circulating current command. The simulation conditions are the same as those for Figure 4. When a U-phase ground fault occurs at t = 1 s, the disclosed method suppresses the voltage of capacitor 54 by a maximum of 9%.

In this way, the positive-phase dq-axis component ( _vd ⁺ , _vq ⁺ ), the negative-phase dq ^- axis component ( _vd- , _vq- ⁾ , the positive-phase dq-axis component ( _id ⁺ , _iq ⁺ ), and the negative-phase dq ^- axis component ( _id- ^, _iq- ) are used to calculate the circulating current command value. Therefore, for example, if an unbalanced fault occurs in the power system, the circulating current responds to the fault more quickly. As a result, compared to conventional methods, the voltage fluctuation of the capacitor can be reduced. Furthermore, for example, the increase in capacitor capacity can be suppressed, allowing for the miniaturization and cost reduction of the power conversion device.

Although the embodiments have been described above, the technology of the present disclosure is not limited to the above embodiments. Various modifications and improvements are possible, such as combining or substituting part or all of the other embodiments.

50, 50UV, 50VW, 50WU clusters 51, 51UV, 51VW, 51WU reactors _52UV1 , _52UV2 , 52UV _n -bridge cells _52VW1 , _52VW2 , 52VW _n- bridge cells _52WU1 , _52WU2 , 52WU _n -bridge cells 53 power conversion circuit 54 capacitor 55 semiconductor switch group 56 switching element 300 power system 400 power conversion device 401 control device 402 delta connection section 410 converter control section 420 current command section a, b AC output terminals

Claims (2)

a delta connection section in which one or a plurality of series-connected bridge cells are delta-connected;
a control device that controls a circulating current flowing in the delta connection by the plurality of bridge cells,
The bridge cell includes a plurality of semiconductor switch groups each including a plurality of semiconductor switches connected in series, and a capacitor connected in parallel to the plurality of semiconductor switch groups,
the control device uses, for calculation of the command value of the circulating current, positive-phase dq-axis components (v _d ⁺ , v _q ⁺ ) of the voltage of the power system to which the delta connection is connected, negative-phase dq-axis components (v _d ^- , v _q ^- ) of the voltage of the power system, positive-phase dq-axis components (i _d ⁺ , i _q ⁺ ) of the current flowing in each phase of the delta connection, and negative-phase dq-axis components (i _d ^- , i _q ^- ) of the current flowing in each phase of the delta connection,
The command value i ₀ ^* of the circulating current is expressed as follows, where j is the imaginary unit.
A power conversion device represented by A control method for a power conversion device including a delta connection portion in which one or a plurality of series-connected bridge cells are delta-connected, the control method controlling a circulating current flowing through the delta connection portion by the plurality of bridge cells,
The bridge cell includes a plurality of semiconductor switch groups each including a plurality of semiconductor switches connected in series, and a capacitor connected in parallel to the plurality of semiconductor switch groups,
a calculation of the command value of the circulating current using positive-phase dq-axis components (v _d ⁺ , v _q ⁺ ) of the voltage of the power system to which the delta connection is connected, negative-phase dq-axis components (v _d ^- , v _q ^- ) of the voltage of the power system, positive-phase dq-axis components (i _d ⁺ , i _q ⁺ ) of the current flowing in each phase of the delta connection, and negative-phase dq-axis components (i _d ^- , i _q ^- ) of the current flowing in each phase of the delta connection ,
The command value i ₀ ^* of the circulating current is expressed as follows, where j is the imaginary unit.
A control method for a power conversion device represented by the formula :