JPS5914980B2 - Switching regulator - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to a switching regulator that can obtain a stable output voltage through pulse width modulation and has an overcurrent protection function. This is to correct.

First, Japanese Patent Application No. 53-20480 (Japanese Unexamined Patent Publication No. 54-113)
The switching regulator proposed in No. 821) is shown in FIG. 1, and the functions of the overcurrent protection device 10 will be explained below.

In the figure, 1 is a blocking oscillation circuit, 2 is an output rectifier circuit,
3 is an error amplification circuit consisting of a voltage comparator Al, a transistor Q3, and a resistor; 4 is a control circuit; and 5 is a current detection circuit.

The switching regulator 15 configured in this manner generates a stable desired DC output voltage V between output terminals a and b when an unstable DC voltage V7 is supplied from the input power source E1. is obtained. FIG. 2 shows the voltage and current waveforms of each part of the circuit shown in FIG. 1, and the circuit operation will be described below. In the figure, A is the base input current ib waveform of the transistor Q1, the top is the collector current IC waveform of the transistor Q1, C is the output voltage Vic waveform of the current detection circuit 5, and 2 is the output current iFB waveform of the control circuit 4. .

First, we will discuss the case in which the circuit shown in Figure 1 is in steady operation. Now, when the transistor Q1 of the blocking oscillation circuit 1 starts conducting, a current tc flows into the l-order winding element Li of the transformer T(1). At this time, transformer T
An induced voltage is generated across the secondary winding L2 of the transistor Q1, but since the polarity of the diode D1 of the output rectifying circuit 2 is opposite to the induced voltage, the collector current IC of the transistor Q1 is almost equal to the excitation current ' ! -A- becomes. In other words, the time shown in Figure 2. At the same time as the transistor Q1 starts conducting, the collector current lc increases linearly.

When i5, the head knocking oscillation circuit 1 generates the minimum base current necessary for conduction of the transistor Q1: 9 - Ic, as shown in the base input current ib waveform shown in Fig. 2A.
If a current n times as high as p/Hfg (however, Bi, 1.TON) is supplied to the base, the transistor Q becomes conductive.

Also, during the conduction period of transistor Q1, transistor Q1
The collector current 16 supplied with sufficient base current NiO, /Hfe is not saturated, and the collector saturation voltage can be made sufficiently small. Furthermore, the output voltage V. and an error amplification circuit 3 corresponding to the deviation value between the reference voltage of the reference voltage element Er and the reference voltage of the reference voltage element Er.
The output from the voltage comparator A2, that is, the voltage V given to the inverting input terminal of the voltage comparator A2, is the output voltage V. The voltage υ is controlled to fall as the voltage υ rises. Collector current 16 output by current detection circuit 5 (however, ≧?−[MeO: egg?=
≠It has been improved.

In the control circuit 4, within the voltage applied to the non-inverting input terminal of the voltage comparator A2. The voltage applied to the inverting input terminal is compared with ψ. 〉U, the transistor Q2 is conductive and H, . When the relationship is <V, the transistor Q2 is cut off. Therefore, as mentioned above, the collector current of transistor Q1 is 1. Since the voltage increases linearly, the voltage I is proportional to the collector current 16. also increases linearly. At time t1 shown in FIG. 2, the relationship between IO and V becomes ViO>v, and as a result, transistor Q2 becomes conductive, and a collector current 1Fe of transistor Q2 flows as shown in FIG. The collector current 1FS acts to completely cancel out the base current 1b flowing into the transistor Q1, and can quickly eliminate even the accumulated carriers remaining in the transistor Q1. That is, when the output voltage '7J16 of the current detection circuit 5 exceeds the output voltage ν of the error amplifier circuit 3, the transistor Q2 becomes conductive and the transistor Q1
shuts off rapidly. The circuit of FIG. 1 has a base current Nl sufficient for the base current value required when the transistor Q1 is conductive, as shown in the base current 1b waveform shown in
By supplying ep/Hfe, collector saturation voltage is suppressed and collector loss is reduced.

Further, at the time when the collector current IO of the transistor Q1 reaches a peak value corresponding to the stabilization of the output voltage, that is, at time T2 shown in FIG. By operating the control circuit 4 to forcibly cut off the transistor Q1, it is possible to further reduce the collector loss when the transistor Q1 is cut off, thereby increasing the power conversion efficiency of the switching regulator. be able to. Furthermore, since the carrier accumulation time when the transistor Q1 is turned off can be significantly shortened, the maximum oscillation frequency can be increased, and thereby the pulse width modulation range of the switching regulator can be widened.
Therefore, Figure 1 shows that the load range can be expanded to the light load side, and the output voltage due to the frequency limit at light loads. Preventing abnormal rise in output voltage. Improves stability. The conduction time TON of the transistor Q1 in the oscillation state of the switching regulator shown in FIG.
) is expressed by the formula.

However, L1 is the inductance of the primary winding, R5 is the current detection resistor, V1 is the input voltage, and P is the output of the error amplification circuit.

As is clear from equation (1), the conduction time TON of the transistor Q1 is a function of the output voltage of the error amplifier circuit 3, and the blocking oscillation circuit 1 performs pulse width modulated oscillation due to factors such as load fluctuations and input fluctuations. By controlling the excitation energy accumulated in the transformer T, the output voltage o is stabilized. Furthermore, this switching regulator performs the overcurrent protection function as described below without incorporating any special parts for a current protection circuit.

As mentioned above, the circuit in FIG. 1 has an output current of 1.

With respect to the output voltage V. In order to stabilize the reference voltage of the reference voltage element Er and the output voltage V based on the above equation (1). The conduction time TON of the blocking oscillation circuit 1 is controlled by the error amplification circuit output P obtained in accordance with the deviation value of . However, the power supply to the error amplifier circuit 3 is the output voltage. Therefore, the operating range of the error amplifier circuit output v is the output voltage V. limited by. That is,
Error amplifier circuit output 0 and output voltage. Since the relationship is ≦o, the output current is 1. When the conduction time TON, which increases as the on-off time increases, reaches the relationship expressed by equation (2), the increase is limited. In the circuit shown in FIG. 1, after the above equation (2) is satisfied, the output current is 1.

output voltage when increasing, that is, during overcurrent.
and output current 1. I will explain the relationship between Here, when the transistor Q1 is cut off, the excitation energy stored in the transformer T causes the output voltage V from the output rectifier circuit 2 to be expressed by equation (3). is extracted as. However, it is assumed that there is no forward voltage drop in the rectifier tie D1.

Therefore, input power Pi is expressed as equation (4), output power P. is expressed by equation (5). Generally, power conversion efficiency is 100%
Assuming that, input power Pl and output power P. are equal, so from equations (4) and (5), the ratio of the conduction time TON and cut-off time TOFF of transistor Q1 is further expressed as (2) and (6)
From the formula, the cut-off time TOFF of the transistor Q1 at the time of overcurrent detection=... (7).

That is, from equations (2), (3), and (7), TON, tOF
If F is erased, the output voltage V when overcurrent is detected.

The relationship between and load resistance RL is 1... (8)
becomes.

Also the output voltage. and output current 1. The relationship is V. =IORL
As a result, the output current is 1. is 1. ... (9).

Therefore, if the load resistance RL is eliminated from equations (8) and (9), the output voltage V.

and output current 1. The relationship is as shown in the following equation (UO).・
・・・・・・・・・(substitute) Furthermore, this relationship is expressed by the output voltage VO-I as shown in Figure 3.
，／←? Output current.

=? It becomes a hyperbolic function with an asymptote of 5. Equation (substitute) holds true when the relationship between the output V of the error amplifier circuit 3 and the output voltage is υ=VO, and when P=o, that is, the output current is 1. When is not the solution current, the output voltage V from the above equation (3). is controlled and stabilized by the reference voltage of reference voltage element E〒. That is, in FIG. 3, the output voltage is V. >O, if the output current is IO>O, then the output voltage V. and output power section 1. The relationship is shown in (3) above.
) formula and (self) formula are combined, resulting in the fourth
It is possible to obtain the overcurrent protection function, so-called foldback characteristic, as shown in the figure. Next, the output voltage V shown in FIG.

and output current 1. Let's briefly explain the relationship. Output current 1.

is within the rated value, and the output υ of the error amplifier circuit 3 is
is within the operating range proportional to the deviation value between the DC output voltage obtained from the output rectifier circuit 2 and the reference voltage of the reference voltage element Er, the relationship of VO-10 becomes the P-Q line, and the above (3)
The output voltage V based on the formula. is stabilized. On the other hand, the output current is 1. is outside the rated value, that is, in an overcurrent state, and the voltage supplied to the error amplifier circuit 3, that is, the DC output voltage obtained between output terminals A and b. If the operating range is limited by
The output current will be a curve based on the above equation (UO). is controlled. Therefore, the circuit of FIG.
It is possible to obtain an overcurrent protection function such as a Q-0 curve, a so-called foldback characteristic. However, as can be understood from the above equation 00), since it is a function of the voltage Vi, the load current value changes as the input voltage fluctuates, as shown in Figure 4, when the input voltage Vi becomes low, the overcurrent protection When the input voltage Vi increases, it changes to the starting point q, and vice versa, it changes to the overcurrent protection starting point Q'5, which makes it difficult to design a high-precision switching regulator, and it is difficult to design a high-precision switching regulator. It could not be said that the function was necessarily sufficient.

SUMMARY OF THE INVENTION An object of the present invention is to provide a high-precision switching regulator which eliminates the above-mentioned drawbacks and corrects variations due to input fluctuations in the load current value at which overcurrent protection starts.

That is, the present invention connects the primary winding of the transformer, the collector-emitter of the oscillation transistor, and the first series circuit with the current detection circuit to the DC input power supply, and connects the parallel circuit of the capacitor and resistor and the feedback winding of the transformer. A second series circuit with the wire is connected between the base and emitter of the oscillation transistor, and the current detection circuit detects the induced voltage generated in the first circuit portion and the feedback winding, which detects a current proportional to the current in the primary winding. The second circuit part includes a second circuit part that adds the current voltage obtained by rectifying and smoothing the current to the first circuit part as the output of the current detection circuit, and includes a primary winding, an oscillation transistor, and a second series circuit. A relaxation oscillator circuit configured, a rectifier circuit that rectifies the AC output of the secondary winding of the transformer, and a DC output voltage obtained from the rectifier circuit and a reference voltage are compared to obtain an output corresponding to the deviation value, and the The output of the current detection circuit includes an error amplification circuit in which the upper limit of the output voltage is limited by a voltage proportional to the DC output voltage of the rectifier circuit, and a voltage comparison circuit that compares each output of the current detection circuit and the error amplification circuit. and a control circuit that applies a trigger signal to the base of an oscillation transistor of the relaxation oscillator circuit to change the intermittent state of the relaxation oscillator circuit to an off state only when j exceeds the output of the error amplifier circuit. shall be.

Next, one embodiment of the present invention is shown in FIG. 5, and will be described below.

In the figure, 11 is a blocking oscillation circuit which obtains a trigger signal to turn off from the outside, and is composed of a primary winding L1 of a transformer T, a feedback winding L3, a base resistor R1, a base capacitor C1, a starting resistor R2, and a transistor Q1. .

Reference numeral 12 denotes an output rectifier circuit, which includes a secondary winding L2 of a transformer T, a diode D, and a smoothing capacitor C2.

Reference numeral 13 denotes an error amplification circuit, which includes a reference voltage element Er and a voltage comparator A1.

14 is a control circuit, which includes a voltage comparator A2 and a transistor Q2.
Consisting of

15 is a current detection circuit, and the feedback winding L3 diode D
2, consisting of a capacitor C3 and resistors R3 and R4.

In the above configuration, since the operation of the main circuit is the same as that of the circuit shown in FIG. 1, the explanation will be omitted, and the operation of the current detection circuit 15 will be explained below.

When a current flows into the primary winding L1 of the transformer T from the input power source Ei, an induced voltage is generated in the feedback winding L3 of the transformer T in proportion to the current.

As a result, the conduction of the transistor Q1 is further deepened, and the induced voltage generated in the feedback winding L3 is transferred to the diode D2.
and rectify and smooth with capacitor C3 to obtain DC voltage,
The direct voltage is divided by resistors R2 and R4. If the voltage generated across the resistor R4 is VR, and the voltage across the resistor R5 that detects the collector current 10 of the transistor Q is Vl6, then the output V'a of the current detection circuit 15 is equal to the voltage V2 across the resistor R4 and the resistor current. The voltage Va=ViO+VR is obtained by superimposing the torsion pressure νIO across the resistor R4. Here, the voltage 1fR across the resistor R4 is expressed by the equation 00, and the voltage v″10 across the resistor R5 is expressed by the equation (self).

However, Vi is the input voltage, L1 is the inductance of the primary winding, N1 is the number of turns of the primary winding, N3 is the number of turns of the feedback winding, R3, R4, R, are each resistance value, It is assumed that there is no forward voltage drop in the rectifier diode D2.

Therefore, the present invention corrects the variation in the output Vi of the current detection circuit 5 in the circuit shown in FIG.

Next, FIG. 6 shows the operating voltage and current waveforms of each part of the circuit shown in FIG. 5, and the overcurrent protection operation will be specifically explained.

In the circuit of FIG. 5, the output signal "a" of the current detection circuit 15 is
If the operating conditions are set so that the overcurrent protection operation starts when the output voltage of the error amplifier circuit 13 becomes equal to the output voltage of the error amplifier circuit 13, as described above, when the relationship between each output becomes 1t'a>v, The control circuit 14 operates and the overbridge protection function works.

In FIG. 6, the time T, O-Tl2 is the transistor Q
1 is the conduction time TON, and time T,2 to T,3 is the cutoff time TOFF of the transistor Q.

Now, at time TlO, the transistor Q1 becomes conductive and the collector current is 1. increases linearly, so the collector current is 1. A voltage proportional to /L/. IO also increases linearly. The relationship between the output υa of the current detection circuit 15, which is the voltage obtained by superimposing the voltage V-10 on the voltage Ra given according to the input voltage i, and the output ν of the error amplification circuit 13 is ν・c> at time Tll. As a result, the collector current 1FR of the transistor Q2 of the control circuit 14 flows, and the transistor Q1 is strongly cut off. At time Tl2, transistor Q1 is cut off and has a collector current of 1. will stop flowing. However, current detection circuit 1
The output Va of the transistor Q5 has the voltage waveform shown in C of FIG. , as the input voltage Vl increases,
Voltage VR increases. Therefore, as the input voltage Vi changes, the voltage 'V'R changes, but the output t/a of the current detection circuit 15 (however, '7/'a-'L/'R
The voltage 17'IO changes as the voltage increases. As a result, in this embodiment, the output v of the error amplification circuit 13 can be kept approximately constant,
As a result, variations in the overcurrent protection starting point Q are corrected. Also, I in FIG.

-o characteristic, when the input voltage i increases, the solution current protection starting point Q can be set to point ♂, and conversely, the input voltage l
, the starting point Q can be set to point Q''.Thereby, the load current value at which the overcurrent protection operation starts can be kept approximately constant, and the power at the overcurrent protection operation starting point Q can also be set to the point Q''. In other words, the voltage division ratio between resistors R3 and R4 in the current detection circuit 15 shown in FIG.
By compensating the voltage νIO with the voltage sensor R so that the output 6 of the current detection circuit 15 is constant, it is possible to correct variations in the overcurrent protection operation starting point that occur in conventional circuits.

Although the above-described embodiments have been described with respect to a self-excited switching regulator using a blocking oscillation circuit, the same effects as described above can be expected even when the present invention is applied to a separately excited switching regulator.

According to the present invention, there is provided a switching regulator that has excellent power conversion efficiency regardless of the oscillation method, and that can reliably operate the overcurrent protection function without incorporating special overcurrent protection circuit components. Can be provided.

[Brief explanation of the drawing]

Figure 1: Circuit diagram showing a conventional example Figure 2: Diagram showing the operating voltage and current waveforms of each part of the circuit in Figure 1 Figure 3: Diagram used to explain Figure 1 Figure 4: Output in Figure 1 Voltage. and output current 1. FIG. 5: A circuit diagram showing an embodiment of the present invention. FIG. 6: A diagram showing the operating voltage and current waveform of each part of the circuit in FIG. 5. FIG. 7: Output voltage in FIG. 5. FIG. 3 is a diagram showing the relationship between output current 10 and output current 10. FIG. 11... Blocking oscillation circuit, 12... Output rectification circuit,
13...Error amplification circuit, 14...Control circuit, 15...Current detection circuit.

Claims (1)

[Claims] 1 Connect the primary winding of the transformer, the collector-emitter of the oscillation transistor, and the first series circuit with the current detection circuit to a DC input power supply, and connect the parallel circuit of the capacitor and resistor with the feedback winding of the transformer. 2 series circuits are connected between the base and emitter of the oscillation transistor, and the current detection circuit is 1
A first circuit section that detects a current proportional to the current in the next winding and a DC voltage obtained by rectifying and smoothing the induced voltage generated in the feedback winding are added to the first circuit section to output the current detection circuit. a relaxation oscillation circuit consisting of a second circuit portion having a primary winding oscillation transistor and a second series circuit; a rectifier circuit for rectifying the AC output of the secondary winding of the transformer; an error amplifier circuit which compares a DC output voltage obtained from a rectifier circuit with a reference voltage, obtains an output corresponding to the deviation value thereof, and whose upper limit of the output voltage is limited by a voltage proportional to the DC output voltage of the rectifier circuit; , includes a voltage comparison circuit that compares each output of the current detection circuit and the error amplification circuit, and disconnects the intermittent state of the relaxation oscillation circuit only when the output of the current detection circuit exceeds the output of the error amplification circuit. 1. A switching regulator comprising: a control circuit that applies a trigger signal that changes the state to the base of an oscillation transistor of a relaxation oscillation circuit.